OTFS modulation method suitable for underwater high-speed mobile communication

Abstract

The invention discloses an OTFS modulation method suitable for underwater high-speed mobile communication, and the method comprises the steps: generating a real number random number sequence and an imaginary number random number sequence, dividing a time delay grid and a Doppler grid, and generating binary random data; performing digital modulation on the binary random data to generate a coded signal in a time delay-Doppler domain; constructing a synchronization head signal, and carrying out frequency spectrum shifting on the coded signal and the synchronization head signal; the transmitting signal reaches a receiving end through an underwater acoustic channel to obtain a receiving signal; and finding the specific position of the required signal in the received signal, calculating the impulse response and Doppler parameter of the channel, and constructing a channel model to realize high-speed communication of the underwater acoustic channel. According to the method, the information is directly modulated to the time delay-Doppler stretching domain, the method accords with the actual physical process, the coupling effect of the Doppler domain and the time delay domain is considered, the effect is better, the communication rate is improved, and the error rate can be reduced.

Description

Technical Field

[0001] The invention belongs to the field of underwater acoustic communication, relates to an OTFS modulation technology, and specifically relates to an OTFS modulation method suitable for underwater high-speed mobile communication. Background Art

[0002] Orthogonal Time-Frequency Space (OTFS) is an emerging two-dimensional modulation technology that aims to provide reliable communication over time-varying channels. Its main feature is to convert the time-varying channel in the time domain into a channel that is independent of the Delay-Doppler (DD) domain, and perform modulation and demodulation in this domain. Therefore, OTFS modulation is more suitable for underwater acoustic communications with obvious time-varying characteristics, but at present it is mostly used in the field of wireless communications and rarely used in the field of underwater acoustic communications.

[0003] The existing OTFS modulation cannot achieve a large Doppler expansion in the underwater acoustic channel because the traditional OTFS modulation is based on the orthogonal frequency division multiplexing (OFDM) structure, which usually produces a high peak-to-average ratio and easily introduces nonlinear distortion, especially when transmitted in the underwater acoustic channel. The signal with a high peak-to-average ratio is easily affected by nonlinear effects during transmission, resulting in signal distortion and increased bit error rate.

[0004] In underwater acoustic channels, the Doppler effect is very significant and complex. Traditional OTFS modulation uses fast Fourier transform (FFT), which assumes that the frequency offset is linear. However, the Doppler effect in underwater acoustic channels is often nonlinear, which makes it difficult for traditional OTFS modulation to accurately capture and process these nonlinear Doppler effects. Summary of the invention

[0005] Purpose of the invention: In order to overcome the deficiencies in the prior art, an OTFS modulation method suitable for underwater high-speed mobile communications is provided, which directly modulates information into the delay-Doppler stretch domain, which is consistent with the actual physical process and takes into account the coupling effect between the Doppler domain and the delay domain, so as to achieve better results, increase the communication rate, and reduce the bit error rate.

[0006] Technical solution: To achieve the above object, the present invention provides an OTFS modulation method suitable for underwater high-speed mobile communication, comprising the following steps:

[0007] S1: Generate a real random number sequence and an imaginary random number sequence, add these two parts to get a complex random number sequence, divide the delay grid and Doppler grid, and generate binary random data;

[0008] S2: Digitally modulate the binary random data and generate a coded signal in the delay-Doppler domain by accumulating the modulated signal of each grid point;

[0009] S3: Time-reverse the generated random number sequence to construct a synchronization header signal, perform spectrum shifting on the coded signal and the synchronization header signal, and arrange them front and back in the time domain to form a complete signal to be transmitted;

[0010] S4: The transmitted signal reaches the receiving end through the underwater acoustic channel, and the received signal is obtained;

[0011] S5: The receiving end first finds the specific position of the required signal in the received signal through the synchronization header signal, then performs channel estimation, and further solves the impulse response and Doppler parameters of the channel, so as to accurately construct the channel model, and finally uses the least squares decoding to achieve high-speed communication in the underwater acoustic channel.

[0012] Furthermore, in step S1, the delay grid is divided linearly at equal intervals, starting from 0, and the delay value of each grid increases by a fixed step of every n points, where n is greater than the delay resolution of the signal itself × the sampling rate, and M signals are obtained, which are suitable for capturing the delay expansion characteristics in the channel; N Doppler grids stretch the signal in equal proportions, so that the Doppler frequency shift increases exponentially between different grids, and finally M×N signals are obtained.

[0013] Furthermore, in step S2, the digital modulation is QPSK modulation, the generated binary data is mapped to the corresponding QPSK constellation points, every two bits are mapped to one constellation point, and the modulated data is stored in an M*N matrix.

[0014] Furthermore, the spectrum shifting in step S3 is specifically as follows: calculating the required interpolation factor, that is, dividing the sampling frequency of the signal by the bandwidth of the signal (the bandwidth of the signal is determined according to the bandwidth range of the system operation); then interpolating the coded signal according to the interpolation factor to achieve bandwidth compression of the signal, and then multiplying it by ejωt in the time domain, where ω=2*pi*fc, fc is the center frequency, which is determined by the center frequency of the system operation, and shifting the coded signal to the position of the center frequency fc; finally, performing the same processing on the constructed synchronization header signal as the above-mentioned coded signal, the coded signal and the synchronization header signal can be spectrally shifted.

[0015] Furthermore, the starting part of the signal to be transmitted in step S3 is the synchronization header signal after spectrum shifting, the middle part has a zero interval and a length that is m-1 times the length of the synchronization header signal, and the last part is the coded signal after spectrum shifting.

[0016] Furthermore, the method for determining the specific position of the required signal in the received signal in step S5 is:

[0017] Different interpolation factors ka are used to interpolate the synchronization header signal, and then cross-correlate it with the received signal; ka = 1 + σ, where (v is the moving speed of the target, c is the speed of sound, which is 1500m / s) calculated; the interpolated signal is cross-correlated with the received signal, and the interpolation factor corresponding to the maximum moment of the cross-correlation peak is extracted to determine it as the Doppler frequency shift deviation dka ′ ; Then, dka ′ As a new interpolation factor, the synchronization head signal is Doppler compensated by interpolation, and the compensated signal is cross-correlated with the received signal. The position of the cross-correlation peak is the starting position of the signal, so as to find the real position of the signal.

[0018] Furthermore, the method for calculating the impulse response and Doppler parameters of the channel in step S5 is:

[0019] First, a rough Doppler estimation is performed, and the interpolation factor corresponding to the maximum time of the cross-correlation peak is extracted as a rough estimate of the Doppler frequency shift dka ′ , thereby determining a rough frequency shift range; then, based on the rough estimate, further narrow the search range and use a smaller step size to accurately estimate the Doppler frequency shift; in the same way as the rough Doppler estimate, interpolate the synchronization head signal according to different interpolation factors ki, make a more precise adjustment to the Doppler frequency shift, calculate the correlation again, and find a more accurate Doppler frequency shift amount dki ′ ; Finally, according to the accurate estimation of Doppler frequency shift dki ′ The synchronization header signal is interpolated so that the interpolated synchronization header signal syncd _ki Cross-correlate with the intercepted received signal to find the real position of the signal; subtract the signal component corresponding to the cross-correlation peak found each time from the received signal, that is, to eliminate the multipath components in the received signal one by one, and repeat the above process until the Doppler offset of the synchronization head signal exceeds the signal range;

[0020] The Doppler coefficient of the channel δ = 1 + dk i ; Channel amplitude a = peak ÷ ∑ (sync dki ×sync dki *), where peak is the interpolated synchronization header signal sync dki The peak-to-peak value after cross-correlation with the intercepted received signal; the channel delay τ = (lags x -1)÷Fs, where laggs x is the starting point of the true position of the signal found, Fs is the sampling rate of the signal; the channel impulse response (a, τ) and Doppler parameter (δ) calculated for one of the multipath solutions.

[0021] Furthermore, in step S5, the channel parameters calculated are added to the original random number sequence to generate a reference signal, all the generated reference signals are stored row by row, a reference matrix X is constructed, and the least squares formula (X T X)b=X T h is decoded, where b is the coefficient to be solved and h is a signal segment intercepted from the received signal with minimum delay. QPSK demodulation is performed on b, and the demodulated result is the final decoding result.

[0022] Furthermore, the specific method of adding the calculated channel parameters to the original random number sequence to generate the reference signal in step S5 is:

[0023] The obtained M×N signals are spectrum shifted; zeros are padded in front of the spectrum shifted signals according to the calculated channel delay parameters, and the number of zero padding points is the delay of each multipath minus the minimum value of the channel delay parameter and then multiplied by Fs; then, the spectrum shifted signals are interpolated according to the calculated channel Doppler coefficient, and the interpolated signals are multiplied by the amplitude parameter of the channel; the above-mentioned zero padding, interpolation and coefficient multiplication operations are performed on each multipath; all multipath signals that take into account the influence of channel parameters (a, τ, δ) are superimposed to obtain a reference signal.

[0024] Beneficial effects: Compared with the prior art, the present invention directly modulates information into the delay-Doppler stretch domain, which is consistent with the actual physical process and takes into account the coupling effect between the Doppler domain and the delay domain, so the effect is better, the communication rate is increased, and the bit error rate can be reduced;

[0025] Regarding the problem of signal distortion and increased bit error rate caused by high peak-to-average ratio, the present invention uses a complex Gaussian random sequence as the carrier signal. Compared with the traditional FFT method, the peak-to-average ratio of this signal is lower because the statistical characteristics of the complex Gaussian signal are more uniform and will not produce high peak values, thereby reducing signal distortion. Regarding the problem of the Doppler effect, the present invention modulates the information directly into the delay-Doppler stretch domain, which is highly consistent with the physical laws of the actual Doppler occurrence, and takes into account the coupling effect between the Doppler domain and the delay domain, and can resist strong Doppler interference. Therefore, the present invention solves the above problems at the same time, and the OTFS modulation method of the present invention is suitable for underwater high-speed mobile communications. BRIEF DESCRIPTION OF THE DRAWINGS

[0026] Figure 1 The figure is a flowchart of the Random-OTFS modulation and demodulation process in the method of the present invention.

[0027] Figure 2 It is a specific flow chart of the method of the present invention.

[0028] Figure 3 This is a diagram of the spectrum shifting result obtained in this embodiment.

[0029] Figure 4 FIG. 4 is a diagram of the multi-path reduction process obtained in this embodiment.

[0030] Figure 5 This is a channel result diagram obtained in this embodiment.

[0031] Figure 6 1 is a decoding constellation diagram under different signal-to-noise ratios obtained in this embodiment.

[0032] Figure 7 This is a bit error rate curve diagram obtained in this embodiment. DETAILED DESCRIPTION

[0033] The present invention is further explained below in conjunction with the accompanying drawings and specific embodiments. It should be understood that these embodiments are only used to illustrate the present invention and are not used to limit the scope of the present invention. After reading the present invention, various equivalent forms of modifications to the present invention by those skilled in the art all fall within the scope defined by the claims attached to this application.

[0034] like Figure 1 As shown, the present invention provides an OTFS modulation method suitable for underwater high-speed mobile communication. The entire Random-OTFS modulation and demodulation process is as follows Figure 1 As shown, it includes generating a binary random number and then digitally encoding it, then performing Random-OTFS modulation on the encoded signal, performing Random-OTFS demodulation on it after passing through a shallow sea channel, then decoding it, and finally calculating the bit error rate.

[0035] Based on the above Random-OTFS modulation and demodulation process, the present invention provides an OTFS modulation method suitable for underwater high-speed mobile communication, such as Figure 2 As shown, it includes the following steps:

[0036] S 1: Generate a real random number sequence and an imaginary random number sequence, add these two parts to get a complex random number sequence, divide the delay grid and Doppler grid, and generate binary random data;

[0037] The delay grid is divided into equally spaced linear grids, starting from 0. The delay value of each grid increases by a fixed step of every n points, where n is greater than the delay resolution of the signal itself × the sampling rate, resulting in M ​​signals, which are suitable for capturing the delay spread characteristics in the channel. The N Doppler grids stretch the signal in equal proportions, causing the Doppler frequency shift to increase exponentially between different grids, ultimately resulting in M ​​× N signals.

[0038] S2: Digitally modulate the binary random data and generate a coded signal in the delay-Doppler domain by accumulating the modulated signal of each grid point;

[0039] The digital modulation is QPSK modulation, and the generated binary data is mapped to the corresponding QPSK constellation points, with every two bits mapped to one constellation point, and the modulated data is stored in an M*N matrix.

[0040] S3: Time-reverse the generated random number sequence to construct a synchronization header signal, perform spectrum shifting on the coded signal and the synchronization header signal, and arrange them front and back in the time domain to form a complete signal to be transmitted;

[0041] The specific steps of spectrum shifting are as follows: calculate the required interpolation factor, that is, divide the sampling frequency of the signal by the bandwidth of the signal (the bandwidth of the signal is determined according to the bandwidth range of the system); then interpolate the encoded signal according to the interpolation factor to achieve bandwidth compression of the signal, and then multiply it by ejω in the time domain t , where ω=2*pi*fc, fc is the center frequency, which is determined by the center frequency of the system. The coded signal is moved to the position of the center frequency fc; finally, the constructed synchronization header signal is processed in the same way as the above-mentioned coding signal, so that the spectrum of the coding signal and the synchronization header signal can be shifted.

[0042] The starting part of the signal to be transmitted is the synchronization header signal after spectrum shifting, the middle part has a zero interval and its length is m-1 times the length of the synchronization header signal, and the last part is the coded signal after spectrum shifting; the synchronization header is designed to use the synchronization header signal to find the exact position of the signal to be decoded during decoding, and the zero interval is designed to reduce interference between signals.

[0043] S4: The transmitted signal reaches the receiving end through the underwater acoustic channel, and the received signal is obtained;

[0044] S5: The receiving end first finds the specific position of the required signal in the received signal through the synchronization header signal, then performs channel estimation, further solves the impulse response and Doppler parameters of the channel, thereby accurately constructing the channel model, and finally uses the least squares method for decoding to achieve high-speed communication in the underwater acoustic channel;

[0045] The method for determining the specific position of the required signal in the received signal is:

[0046] Different interpolation factors ka are used to interpolate the synchronization header signal, and then cross-correlate it with the received signal; ka = 1 + σ, where (v is the moving speed of the target, c is the speed of sound, which is 1500m / s) and is calculated; the interpolated signal is cross-correlated with the received signal, and the interpolation factor corresponding to the maximum moment of the cross-correlation peak is extracted to determine it as the Doppler frequency shift deviation d k a′; then, d ka′ is used as a new interpolation factor, and the Doppler compensation of the synchronization header signal is performed by interpolation. The compensated signal is cross-correlated with the received signal. The position of the cross-correlation peak is the starting position of the signal, so as to find the real position of the signal.

[0047] The channel impulse response and Doppler parameters are calculated as follows:

[0048] First, a rough Doppler estimation is performed, and the interpolation factor corresponding to the maximum time of the cross-correlation peak is extracted as a rough estimate of the Doppler frequency shift d k a′, thereby determining a rough frequency shift range; then, based on the rough estimate, further narrow the search range and use a smaller step size to accurately estimate the Doppler frequency shift; in the same way as the rough Doppler estimate, interpolate the synchronization head signal according to different interpolation factors ki, make a more precise adjustment to the Doppler frequency shift, calculate the correlation again, and find a more accurate Doppler frequency shift d k i′; finally, the Doppler frequency shift dk is accurately estimated i 'Interpolate the synchronization header signal so that the interpolated synchronization header signal sync dki Cross-correlate with the intercepted received signal to find the real position of the signal; subtract the signal component corresponding to the cross-correlation peak found each time from the received signal, that is, to eliminate the multipath components in the received signal one by one, and repeat the above process until the Doppler offset of the synchronization head signal exceeds the signal range;

[0049] The Doppler coefficient of the channel δ = 1 + dk i ; Channel amplitude a = peak ÷ ∑ (sync dki ×sync dki *), where peak is the interpolated synchronization header signal sync dki The peak-to-peak value after cross-correlation with the intercepted received signal; the channel delay τ = (lags x -1)÷Fs, where lags x is the starting point of the true position of the signal found, Fs is the sampling rate of the signal; the channel impulse response (a, τ) and Doppler parameter (δ) calculated for one of the multipath solutions.

[0050] The calculated channel parameters are added to the original random number sequence to generate a reference signal. All the generated reference signals are stored row by row to construct a reference matrix X. The least squares formula (X T X)b=X T h is decoded, where b is the coefficient to be solved and h is a signal segment intercepted from the received signal with minimum delay. QPSK demodulation is performed on b, and the demodulated result is the final decoding result.

[0051] The specific method of adding the calculated channel parameters to the original random number sequence to generate the reference signal is:

[0052] The obtained M×N signals are spectrum shifted; zero padding is performed in front of the spectrum shifted signals according to the calculated channel delay parameters, and the number of zero padding points is the delay of each multipath minus the minimum value of the channel delay parameter multiplied by Fs; then, the spectrum shifted signals are interpolated according to the calculated channel Doppler coefficient, and the interpolated signals are multiplied by the amplitude parameter of the channel; the above-mentioned zero padding, interpolation and coefficient multiplication operations are performed on each multipath; all multipath signals that take into account the influence of channel parameters (a, τ, δ) are superimposed to obtain a reference signal.

[0053] In order to verify the effectiveness of the method of the present invention, the above-mentioned solution of the present invention is applied as an example in this embodiment, as follows:

[0054] Step (1): Construct a complex random number sequence and map it to the delay-Doppler domain.

[0055] Set the number of sample points to 2700, generate a real random number sequence and an imaginary random number sequence with a length of 2700, add these two parts to get the required complex random number sequence x; then divide the delay grid into 30 and the Doppler grid into 20; add 30 delays to x (τ1=0, τ2=10, τ3=20, ...), each delay increases by 10 points, and get 30 delay signals; then, 20 Doppler domains are proportionally divided. The signal is stretched incrementally (δ1=1, δ2=1.0037, δ3=1.0074, …) to obtain 30×20 signals yi,j (i=1,2,…,30; j=1,2,…,20).

[0056] Step (2): QPSK modulation.

[0057] Generate a binary random number matrix with 60 rows and 20 columns, map the generated binary data to the corresponding QPSK constellation points, map every two bits to a constellation point, and save the modulated data in a 30×20 matrix middle.

[0058] Step (3): modulate the information code onto the delay-Doppler domain divided in step (1) and superimpose them; by accumulating the modulated signal of each grid point, generate a signal y′ modulated in the delay-Doppler domain, as shown in formula (1).

[0059]

[0060] Step (4): Spectrum migration.

[0061] Calculate the required interpolation factor kd = Fs ÷ B = 20, where Fs is the sampling rate of the signal, which is 192kHz, and B is the bandwidth of the signal, which is 9.6×10 3 bps, B is determined by the bandwidth range of the system. Interpolate y′ according to kd to achieve bandwidth compression of the signal, and then multiply it by ejω in the time domain t (where ω=2*pi*fc, fc is the center frequency of 12kHz, determined by the center frequency of the system), the frequency of y′ is shifted to fc to obtain a new signal y″. Time-reverse x to construct signal sync1, and perform the same spectrum shifting process on sync1 as y′ to generate a synchronization header signal sync for marking the start of data.

[0062] In this embodiment, the Figure 3 Shown is the result of shifting the y′ spectrum.

[0063] Step (5): Time domain arrangement.

[0064] The starting part is sync, the middle part is a zero interval, the length is 1 / 4 of the length of the sync signal, and the last part is y". In this way, the time domain arrangement of the sync and y" after the spectrum is moved is completed to generate the final signal sig to be transmitted.

[0065] Step (6): sig reaches the receiving end after passing through the shallow sea channel, and the received signal rec is obtained.

[0066] Step (7): Find the true location of the signal.

[0067] Select different interpolation factors ka to interpolate sync and perform cross-correlation operation with rec. In the range of [-0.01, 0.01], it increases by σ (the range selected by ka must cover the maximum movement speed of the target), where (where v is the moving speed of the target, c is the speed of sound, which is 1500 m / s). The interpolated signals syncka and rec are cross-correlated, and the interpolation factor ka′ corresponding to the maximum moment of the cross-correlation peak is extracted to determine it as the Doppler frequency shift deviation d ka '=-0.0068. Then, d ka ′ is used as a new interpolation factor, and Doppler compensation is performed on sync by interpolation. The compensated signal is cross-correlated with rec, and the position of the cross-correlation peak is the starting position of the signal, so as to find the real position of the signal.

[0068] Step (8): Channel estimation.

[0069] Set the maximum number of multipaths to 70. First, make a rough Doppler estimate of one of the multipaths. ka ′=--0.0068 as a rough estimate of the Doppler frequency shift, and determine an approximate frequency shift range. Then, based on the rough estimate, further narrow the search range so that the interpolation factor ki is within [d ka ′-0.0002, d ka ′+0.0002], use a smaller step size Perform an accurate Doppler shift estimate. In the same way as the rough Doppler estimate, interpolate sync according to ki, make a finer adjustment to the Doppler shift, calculate the correlation again, and determine a more accurate Doppler shift amount dk i '=-0.0069. Based on this precise estimate, sync is interpolated and cross-correlated with rec to find the true position of the signal. The signal components corresponding to the cross-correlation peaks found each time are subtracted from rec one by one, that is, the multipath components in rec are subtracted one by one, and the above process is repeated until the Doppler offset of sync exceeds the range of rec.

[0070] The Doppler coefficient of the channel δ = 1 + dk i ′=0.9931;channel amplitude a=peak÷∑(sync dki ×sync dki *)=0.1066+0.0368i, where peak is sync dki The peak-to-peak value after cross-correlation with rec is 5.6550×10 10 +1.9524×10 10 i; channel delay τ = (lags x -1)÷Fs=0.0104, where lags x is the starting point of the true position of the found signal, and its size is 2004. This is the channel impulse response (a, τ) and Doppler parameter (δ) calculated for one of the multipath solutions.

[0071] In this embodiment, the Figure 4 The results of multipath reduction are shown as well as Figure 5 Shown is the result of acquiring the channel.

[0072] Step (9): Construct a reference matrix for decoding. Perform the spectrum shift described in step (4) on each signal in yij to generate a new signal y ij ″. According to the channel delay parameter τ in y ij " before the zero padding, the number of zero padding points c = (τ-Tmi n )×Fs, where Tmi nis the minimum value of the channel delay parameter, which is 0.0092. Then, according to the channel Doppler coefficient δ, yi j ″ is interpolated, and the interpolated signal is multiplied by the amplitude parameter a of the channel. The above zero padding, interpolation and coefficient multiplication operations are performed on each multipath. All multipath signals that take into account the influence of channel parameters (a, τ, δ) are superimposed to obtain a reference signal xdd. Finally, all the generated reference signals are stored row by row to construct a reference matrix X (l = δm ax ×(2700+10×(30-1))×δm ax ×kd+c max =88030).

[0073] Step (10): Decoding. Using the least squares formula (X T X)b=X T h is decoded, where X is the reference matrix constructed in step (9), h is a signal segment intercepted from the received signal with minimum delay, with a length of l, and b is the coefficient to be solved (600). QPSK phase division decision is made on b, and the result after the decision is the final binary decoding result.

[0074] like Figure 6 The decoded constellation diagrams under different signal-to-noise ratios are shown. As the signal-to-noise ratio increases, the points in the constellation diagram become more concentrated and the phase distinction of each point becomes clearer.

[0075] like Figure 7 The bit error rate curve is shown in Figure 2. As the signal-to-noise ratio increases, the bit error rate decreases. The final bit error rate is calculated by counting the number of erroneous bits sent by the transmitting point and dividing it by the total number of bits sent by the transmitting point.

Claims

1. An OTFS modulation method suitable for underwater high-speed mobile communication, characterized in that: The steps include: S1: Generate a real random number sequence and an imaginary random number sequence, add these two parts to get a complex random number sequence, divide the delay grid and Doppler grid, and generate binary random data; S2: Digitally modulate the binary random data and generate a coded signal in the delay-Doppler domain by accumulating the modulated signal of each grid point; S3: Time-reverse the generated random number sequence to construct a synchronization header signal, perform spectrum shifting on the coded signal and the synchronization header signal, and arrange them front and back in the time domain to form a complete signal to be transmitted; S4: The transmitted signal reaches the receiving end through the underwater acoustic channel, and the received signal is obtained; S5: The receiving end first finds the specific position of the required signal in the received signal through the synchronization header signal, then performs channel estimation, and further solves the impulse response and Doppler parameters of the channel, so as to accurately construct the channel model, and finally uses the least squares decoding to achieve high-speed communication in the underwater acoustic channel.

2. The OTFS modulation method suitable for underwater high-speed mobile communication according to claim 1, characterized in that: In step S1, the delay grid is divided into equally spaced linear grids, starting from 0, and the delay value of each grid increases by a fixed step of every n points, where n is greater than the delay resolution of the signal itself × the sampling rate, and M signals are obtained, which are suitable for capturing the delay spread characteristics in the channel; N Doppler grids stretch the signal in an equal proportion, so that the Doppler frequency shift increases exponentially between different grids, and finally M×N signals are obtained.

3. The OTFS modulation method suitable for underwater high-speed mobile communication according to claim 1, characterized in that: In step S2, the digital modulation is QPSK modulation, and the generated binary data is mapped to the corresponding QPSK constellation points, with every two bits mapped to one constellation point, and the modulated data is stored in an M*N matrix.

4. The OTFS modulation method suitable for underwater high-speed mobile communication according to claim 1, characterized in that: The spectrum shifting in step S3 is specifically as follows: calculating the required interpolation factor, that is, dividing the sampling frequency of the signal by the bandwidth of the signal; then interpolating the encoded signal according to the interpolation factor to achieve bandwidth compression of the signal; then multiplying it by e in the time domain. j ωt in, ω=2*pi*fc, fc is the center frequency, which is determined by the center frequency of the system. The coded signal is moved to the position of the center frequency fc. Finally, the constructed synchronization head signal is processed in the same way as the above-mentioned coding signal, so that the spectrum of the coding signal and the synchronization head signal can be shifted.

5. The OTFS modulation method suitable for underwater high-speed mobile communication according to claim 4, characterized in that: The starting part of the signal to be transmitted in step S3 is the synchronization header signal after spectrum shifting, the middle part has a zero interval and a length that is m-1 times the length of the synchronization header signal, and the last part is the coded signal after spectrum shifting.

6. The OTFS modulation method suitable for underwater high-speed mobile communication according to claim 1, characterized in that: The method for determining the specific position of the required signal in the received signal in step S5 is: Different interpolation factors ka are used to interpolate the synchronization header signal, and then cross-correlate it with the received signal; ka = 1 + σ, where Calculate and cross-correlate the interpolated signal with the received signal, extract the interpolation factor corresponding to the maximum moment of the cross-correlation peak, and determine it as the Doppler frequency shift deviation d ka′ ; Then, d ka′ As a new interpolation factor, the synchronization head signal is Doppler compensated by interpolation, and the compensated signal is cross-correlated with the received signal. The position of the cross-correlation peak is the starting position of the signal, so as to find the real position of the signal.

7. The OTFS modulation method suitable for underwater high-speed mobile communication according to claim 6, characterized in that: The method for calculating the impulse response and Doppler parameters of the channel in step S5 is: First, a rough Doppler estimation is performed, and the interpolation factor corresponding to the maximum time of the cross-correlation peak is extracted as a rough estimate of the Doppler frequency shift d ka′ , thereby determining a rough frequency shift range; then, based on the rough estimate, further narrow the search range and use a smaller step size to accurately estimate the Doppler frequency shift; in the same way as the rough Doppler estimate, interpolate the synchronization head signal according to different interpolation factors ki, make a more precise adjustment to the Doppler frequency shift, calculate the correlation again, and find a more accurate Doppler frequency shift d ki′ ; Finally, according to the accurate estimation of Doppler frequency shift d ki′ Interpolate the synchronization header signal to make the interpolated synchronization header signal sync d_ki Cross-correlate with the intercepted received signal to find the true position of the signal; Subtract the signal component corresponding to the cross-correlation peak value found each time from the received signal, that is, to eliminate the multipath components in the received signal one by one, and repeat the above process until the Doppler shift of the synchronization head signal exceeds the signal range; The Doppler coefficient of the channel δ = 1 + d ki ; Channel amplitude a = peak ÷ ∑ (sync d_ki ×sync d_ki * ), where peak is the interpolated synchronization header signal sync d_ki The peak-to-peak value after cross-correlation with the intercepted received signal; the channel delay τ = (lags x -1)÷Fs, where lags x is the starting point of the true position of the signal found, Fs is the sampling rate of the signal; the channel impulse response (a, τ) and Doppler parameter (δ) calculated for one of the multipath solutions.

8. The OTFS modulation method suitable for underwater high-speed mobile communication according to claim 7, characterized in that: In step S5, the calculated channel parameters are added to the original random number sequence to generate a reference signal, all the generated reference signals are stored row by row, a reference matrix X is constructed, and the least squares formula (X) is used. T X)b=X T h is decoded, where b is the coefficient to be solved and h is a signal segment intercepted from the received signal with minimum delay. QPSK demodulation is performed on b, and the demodulated result is the final decoding result.

9. The OTFS modulation method suitable for underwater high-speed mobile communication according to claim 8, characterized in that: The specific method of adding the calculated channel parameters to the original random number sequence to generate the reference signal in step S5 is: Perform spectrum shifting on the obtained M×N signals; perform zero padding in front of the spectrum shifted signals according to the calculated channel delay parameters, and the number of zero padding points is the delay of each multipath minus the minimum value of the channel delay parameter multiplied by Fs; then, interpolate the spectrum shifted signals according to the calculated channel Doppler coefficient, and multiply the interpolated signals by the amplitude parameter of the channel; perform the above zero padding, interpolation and coefficient multiplication operations on each multipath; superimpose all multipath signals that take into account the influence of channel parameters (a, τ, δ) to obtain a reference signal.

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