A wireless power transmission topology with strong anti-deviation performance based on multi-frequency energy parallel transmission

Abstract

The present invention discloses a wireless power transmission topology with strong anti-offset performance based on multi-frequency energy parallel transmission. The control module outputs a driving square wave signal of a high-frequency inverter, so that the output voltage of the high-frequency inverter is a mixed superposition of dual-frequency or multi-frequency voltages. After passing through the primary multi-frequency shared compensation network, the dual-frequency or multi-frequency current simultaneously excites the primary transmitting coil, and the multi-frequency energy is transmitted to the receiving module in parallel through magnetic field coupling. The receiving module realizes dual-frequency or multi-frequency energy separation and decoupled transmission, and outputs AC voltage or current with each frequency respectively, which are rectified and filtered respectively and then combined and output to power the load. The design of the resonance parameters makes the output at different frequencies inconsistent with the change trend of mutual inductance. After the combined output, an output voltage or current that is insensitive to the change of mutual inductance is obtained, which greatly reduces the output fluctuation caused by the change in the relative position of the transmitting coil and the receiving coil, and ensures the stable output of the wireless power transmission system under high-offset conditions.

Description

Technical Field

[0001] The invention relates to a wireless power transmission topology with strong anti-deviation performance, and belongs to the field of electric energy conversion. Background Art

[0002] Inductive wireless power transmission uses magnetic field coupling to achieve "wireless power supply", that is, a non-contact transformer with completely separated primary and secondary sides is used to transmit power through coupling of high-frequency magnetic fields, so that there is no physical connection between the primary side (power supply side) and the secondary side (power consumption side) during the energy transfer process. Compared with traditional contact power supply, non-contact power supply is convenient and safe to use, without sparks and electric shock hazards, dust accumulation and contact loss, mechanical wear and corresponding maintenance problems, can adapt to a variety of severe weather and environments, and is easy to realize automatic power supply, with broad application prospects.

[0003] However, the change of the relative position between the primary and secondary sides of the non-contact transformer causes the transformer parameters to change over a large range, resulting in fluctuations in the system output and a significant decrease in transmission efficiency, which limits the promotion and application of WPT technology. In order to improve the anti-deviation capability of the wireless power transmission system, Mickel Budhia, John T. Boys, Grant A. Covic and Chang-Yu Huang, "Development of a Single-Sided Flux Magnetic Coupler for Electric Vehicle IPT Charging Systems" IEEE Transactions on Industrial Electronics, vol. 60, no. 1, January 2013 proposed to superimpose a third winding (referred to as Q winding) overlapping the secondary winding between the two windings (referred to as DD winding) of the non-contact transformer secondary side to reduce the lateral misalignment sensitivity of the secondary output power, and better solve the problem of the "inductive blind spot" of "complete cancellation of the inflow and outflow fluxes" when misaligned, which affects the power transmission capacity of the transformer. However, this DDQ winding structure can only improve the output characteristics of the non-contact transformer under the condition of lateral misalignment. The output characteristics of this "DDQ" winding structure still change greatly when the vertical distance between the primary and secondary sides changes (i.e., the air gap changes). Considering the uncertainty of the air gap size and misalignment between the primary and secondary sides of the non-contact transformer in practical applications, further research is still needed.

[0004] Chinese Patent 201720241345.5, a voltage source and current source composite excitation non-contact conversion circuit uses a non-contact converter, the output characteristics under constant voltage source excitation are inversely proportional to the primary-secondary coupling coefficient (mutual inductance) of the non-contact transformer, and the output characteristics under constant current source excitation are proportional to the primary-secondary coupling coefficient (mutual inductance) of the non-contact transformer. The voltage source and the current source are compositely excited and combined to output, thereby reducing the system output fluctuation caused by the change of mutual inductance. L. Zhao, DJ Thrimawithana, UK Madawala, AP Hu, and CC Mi. A misalignment-tolerant series-hybrid wireless EV switching system with integrated magnetics, IEEE Transactions on Power Electronics, 2019, 34 (2): 1276-1285. It is proposed to connect the LCC / LCC compensation topology and the S / S compensation topology in series on the input side and in series on the output side to obtain the output characteristics that change non-monotonic with the mutual inductance and improve the misalignment tolerance. However, the existing technical solutions for improving the tolerance of combined output misalignment all require two sets of transformer windings, which consume a large amount of copper and are costly. In addition, the external characteristics are limited by the structure of the contactless transformer. The magnetic flux coupling between the primary and secondary sides cannot be ignored under complex misalignment conditions such as angular and diagonal. This leads to complex output characteristics of the combined converter and difficulty in parameter design. Stable output can only be obtained under certain specific offset directions. Considering that the relative position offset of the primary and secondary sides may be in any direction in actual applications, a new solution needs to be proposed. Summary of the invention

[0005] Purpose of the invention: In view of the above-mentioned prior art, a wireless power transmission topology with strong anti-deviation performance based on multi-frequency energy parallel transmission is proposed, which has small output fluctuations in any offset direction.

[0006] Technical solution: A wireless power transmission topology with strong anti-offset performance based on multi-frequency energy parallel transmission, including a DC power supply, a high-frequency inverter, a primary multi-frequency shared compensation network, a primary transmitting coil, a receiving module, a load, and a control module;

[0007] The control module outputs a driving square wave signal of the high-frequency inverter; the output of the high-frequency inverter is a mixed superposition of dual-frequency or multi-frequency voltage or current, and the frequencies include f1, f2, ... f n , n≥2; the primary multi-frequency shared compensation network adopts LC high-order compensation topology, and forms a primary resonant network with the primary transmitting coil, and the resonant frequencies are f1, f2, ... f n, multi-frequency energy drives the primary transmitting coil simultaneously and couples to the receiving module through the magnetic field;

[0008] The receiving module includes receiving units of the same number as the frequency number, each receiving unit includes a frequency selection network, a compensation network and an energy coupling coil, and the n receiving units respectively output frequencies of f1, f2, ... f n The AC voltage or current is rectified and filtered by the rectification and filtering units respectively, and then combined and output to supply power to the load.

[0009] Furthermore, the control module controls the switching frequency or conduction time of the switch tube in the high-frequency inverter to achieve the output frequency of the high-frequency inverter being f1, f2, ... and f n A mixed superposition of voltages or currents.

[0010] Furthermore, the control module outputs a square wave signal with a frequency of f to drive the high-frequency inverter, and the high-frequency inverter outputs a square wave voltage superimposed by a fundamental wave of the frequency f and an nth harmonic sinusoidal voltage, wherein the frequencies f1, f2, ..., f n It is the fundamental and / or third and / or fifth or any higher order odd harmonic frequency.

[0011] Furthermore, the control module outputs the frequencies f1, f2, ..., f in a time-sharing manner within a period T. n The lengths of the square wave signals of each frequency are t1, t2, ... t n , satisfying t1+t2+…+t n ≤T, realize time-division multiplexing, and adjust t1, t2, ... and t n The output is regulated by the time ratio.

[0012] Furthermore, by designing the parameters of the LC high-order compensation topology, the output voltage gain or current gain of the primary resonant network is made independent of the equivalent load of the primary resonant network.

[0013] Furthermore, the frequency selection network is an LC hybrid network.

[0014] Furthermore, the primary-side multi-frequency common compensation network includes a switch for implementing parameter switching of the LC high-order compensation topology or uses switched capacitor or inductor tuning, and the control module outputs a drive signal of the switch, switched capacitor or inductor of the primary-side multi-frequency common compensation network, so that the primary-side resonant network resonates at frequency f1 within time t1, resonates at frequency f2 within time t2, ..., t n Time at frequency f n Resonance, satisfying t1+t2+…+t n≤T, multi-frequency energy time-sharing transmission is realized, and its output voltage gain or current gain is independent of the equivalent load of the primary side resonant network.

[0015] Furthermore, the primary transmitting coil is a single winding, double winding or multi-winding structure, two or more windings share a primary multi-frequency shared compensation network, and there is no flux coupling between any two windings or a series LC frequency selection network to achieve decoupling control.

[0016] A wireless power transmission topology with strong anti-offset performance based on multi-frequency energy parallel transmission, including a DC power supply, a high-frequency inverter, a primary multi-frequency shared compensation network, a primary transmitting coil, a receiving module, a load, and a control module;

[0017] The control module outputs a driving square wave signal of the high-frequency inverter; the output of the high-frequency inverter is a mixed superposition of dual-frequency or multi-frequency voltage or current, and the frequencies include f1, f2, ... f n , n≥2; the primary multi-frequency shared compensation network adopts LC high-order compensation topology, and forms a primary resonant network with the primary transmitting coil, and the resonant frequencies are f1, f2, ... f n , and the output voltage gain or current gain of the primary resonant network is independent of the equivalent load of the primary resonant network, and the multi-frequency energy drives the primary transmitting coil at the same time and is coupled to the receiving module through the magnetic field;

[0018] The receiving module includes a secondary receiving coil, a secondary multi-frequency common compensation network, and a multi-frequency common rectification and filtering circuit; the secondary multi-frequency common compensation network is a high-order LC network, which forms a secondary resonant network with the secondary receiving coil, and the secondary multi-frequency common compensation network includes a switch for realizing parameter switching; the control module outputs a switch driving signal of the secondary multi-frequency common compensation network, so that the secondary resonant network is switched at t i The resonant frequency in time is f i , and the output voltage gain or current gain of the secondary resonant network has nothing to do with the equivalent load of the secondary resonant network; the secondary multi-frequency shared compensation network is at t i The output frequency during this time is f i The AC signal is rectified and filtered before supplying power to the load.

[0019] A wireless power transmission topology with strong anti-offset performance based on multi-frequency energy parallel transmission, including a DC power supply, a high-frequency inverter, a primary multi-frequency shared compensation network, a primary transmitting coil, a receiving module, a load, and a control module;

[0020] The control module outputs the frequencies f1, f2, ..., f in a time-sharing manner within a period T. nThe square wave signal of each frequency is used as the driving signal of the high frequency inverter, and the lengths of the square wave signals of each frequency are t1, t2, ..., t n , and satisfy t1+t2+……+t n ≤T, and by adjusting t1, t2, ... and t n Output regulation is achieved by adjusting the time ratio;

[0021] The primary multi-frequency shared compensation network adopts an LC high-order compensation topology and forms a primary resonance network with the primary transmitting coil, and the resonance frequencies are f1, f2, ... f n , and the output voltage gain or current gain of the primary resonant network is independent of the equivalent load of the primary resonant network, so as to realize multi-frequency energy time-sharing driving of the primary transmitting coil and couple it to the receiving module through the magnetic field;

[0022] The receiving module includes a secondary receiving coil, a secondary multi-frequency common compensation network and a rectifier filter unit connected in sequence. The secondary multi-frequency common compensation network is an LC high-order compensation topology, and forms a secondary resonant network with the secondary receiving coil. i The output frequency during this time is f i The AC signal is rectified and filtered to supply power to the load.

[0023] A wireless power transmission topology with strong anti-offset performance based on multi-frequency energy parallel transmission, including a DC power supply, a high-frequency inverter, a primary side transmitting module, a receiving module, a load and a control module;

[0024] The control module outputs a driving square wave signal of the high-frequency inverter. The output of the high-frequency inverter is a mixed superposition of dual-frequency or multi-frequency voltage or current. The frequencies include f1, f2, ..., f n , n≥2; the primary transmitting module includes the same number of transmitting units as the number of frequencies, each transmitting unit includes a frequency selection network, a compensation network and a transmitting coil, and the n transmitting units respectively output frequencies of f1, f2, ... f n The output voltage gain or current gain is independent of the equivalent load of the primary transmitting module;

[0025] The receiving module includes receiving units of the same number as the frequency, each receiving unit includes a frequency selection network, a compensation network and a receiving coil, and outputs frequencies f1, f2, ... f n The AC voltage or current is rectified and filtered by the rectification and filtering units respectively, and then combined and output to supply power to the load.

[0026] Beneficial effects: 1. The present invention combines multi-frequency energy for output, and makes the output at different frequencies inconsistent with the change trend of mutual inductance. After the combined output, an output voltage or current or power that is insensitive to changes in mutual inductance is obtained, thereby reducing the output fluctuation caused by the change in the relative position of the transmitting coil and the receiving coil, and ensuring the stable output of the wireless power transmission system under high offset conditions.

[0027] 2. The present invention adds a frequency selection network to realize the decoupling control of multi-frequency energy, avoiding the mutual influence of cross-coupling, so that the external characteristics are not limited by the transformer structure. When an offset in any direction occurs, the system can still have a stable output voltage or output current under constant input conditions, thereby improving the system's anti-offset capability.

[0028] 3. The present invention uses waveform control, time-division multiplexing and comprehensive utilization of fundamental harmonics to share the inverter, introduce a multi-frequency shared resonant network, and share the primary excitation coil, so that only one set of windings is required for the primary side of the transformer, saving materials and helping to reduce the size and weight of the transformer. BRIEF DESCRIPTION OF THE DRAWINGS

[0029] Figure 1 The wireless power transmission topology circuit structure of the present invention Figure 1 ;

[0030] Figure 2 The wireless power transmission topology circuit structure of the present invention Figure 2 ;

[0031] Figure 3 The wireless power transmission topology circuit structure of the present invention Figure 3 ;

[0032] Figure 4 The wireless power transmission topology circuit structure of the present invention Figure 4 ;

[0033] Figure 5 It is a circuit diagram of a dual-frequency signal shared primary inverter in a wireless power transmission topology of the present invention;

[0034] Figure 6 The topology and harmonic components of the wireless power transmission topology of the present invention using harmonics to achieve multi-frequency energy parallel transmission, wherein Figure 6 (a) is the circuit diagram, Figure 6 (b) is a schematic diagram of harmonic components;

[0035] Figure 7 yes Figure 1 to Figure 6 The equivalent schematic diagram of the primary circuit of the topology shown, where Figure 7 (a) is the primary side equivalent circuit diagram using multi-frequency voltage signal excitation; Figure 7 (b) is the primary side equivalent circuit diagram using multi-frequency current signal excitation;

[0036] Figure 8 It is a circuit diagram of a wireless power transmission topology of the present invention with a dual-winding structure on the secondary side;

[0037] Fig. 9 It is a circuit schematic diagram of a wireless power transmission topology of the present invention with a single winding structure on the secondary side;

[0038] Fig.10 The wireless power transmission topology of the present invention realizes the circuit structure of multi-frequency energy time-sharing transmission Figure 1 ;

[0039] Fig.11 The wireless power transmission topology of the present invention realizes the circuit structure of multi-frequency energy time-sharing transmission Figure 2 ;

[0040] Fig.12 The circuit structure of the wireless power transmission topology of the present invention to realize multi-frequency energy time-sharing transmission Figure 3 ;

[0041] Fig.13 The wireless power transmission topology of the present invention realizes the circuit structure of multi-frequency energy time-sharing transmission Figure 4 ;

[0042] Fig.14 yes Figure 8 is a specific circuit structure diagram of the topology and its equivalent circuit diagram, wherein Fig.14 (a) is a schematic diagram of the circuit structure; Fig.14 (b) is the equivalent circuit diagram;

[0043] Fig.15 This is a circuit diagram of the present invention using a primary-side multi-frequency shared compensation network structure and receiving module outputs in parallel;

[0044] Fig.16 This is a circuit diagram of the present invention using a primary-side multi-frequency shared compensation network structure 2 and receiving module outputs in parallel;

[0045] Fig.17 It is a circuit diagram of the present invention using a primary-side multi-frequency shared compensation network structure 3 and receiving module outputs in parallel;

[0046] Fig.18 It is a circuit diagram of the present invention using a primary-side multi-frequency shared compensation network structure 4 and receiving module outputs in parallel;

[0047] Fig.19 This is a circuit diagram of the present invention using a primary-side multi-frequency shared compensation network structure 5 and receiving module outputs in parallel;

[0048] Fig. 20It is a circuit diagram of a receiving module structure adopted in the wireless power transmission topology of the present invention;

[0049] Fig.21 It is a circuit diagram of a second receiving module structure of the wireless power transmission topology of the present invention;

[0050] Fig. 22 It is a schematic diagram of three circuits of a receiving module structure of a wireless power transmission topology of the present invention;

[0051] Fig.23 This invention is a circuit diagram of a primary-side multi-frequency shared compensation network structure and a receiving module output connected in series;

[0052] Fig.24 It is a circuit diagram of the present invention using a primary-side multi-frequency shared compensation network structure 2 and receiving module outputs connected in series;

[0053] Fig.25 It is a circuit diagram of the present invention using a primary-side multi-frequency shared compensation network structure three and receiving module outputs connected in series;

[0054] Fig.26 It is a circuit diagram of the present invention using a primary-side multi-frequency shared compensation network structure 4 and receiving module outputs connected in series;

[0055] Fig. 27 It is a circuit diagram of the present invention using a primary-side multi-frequency shared compensation network structure 5 and receiving module outputs connected in series;

[0056] Fig.28 The wireless power transmission of the present invention is based on Fig. 9 A specific implementation circuit of the topology shown is one;

[0057] Fig.29 The wireless power transmission of the present invention is based on Fig. 9 Specific implementation circuit 2 of the topology shown;

[0058] Fig.30 The wireless power transmission of the present invention is based on Fig. 9 Specific implementation circuit three of the topology shown;

[0059] Fig.31 The wireless power transmission of the present invention is based on Fig. 9 Specific implementation circuit of the topology shown in Figure 4;

[0060] Fig.32 The wireless power transmission of the present invention is based on Fig. 9 The specific implementation circuit of the topology shown is five;

[0061] Fig.33 The wireless power transmission of the present invention is based on Fig.13A specific implementation circuit of the topology shown is one;

[0062] Fig.34 The wireless power transmission of the present invention is based on Fig.13 Specific implementation circuit 2 of the topology shown;

[0063] Fig.35 The wireless power transmission of the present invention is based on Fig.13 Specific implementation circuit three of the topology shown;

[0064] Fig.36 The wireless power transmission of the present invention is based on Fig.11 A specific implementation circuit of the topology shown is one;

[0065] Fig.37 The wireless power transmission of the present invention is based on Fig.11 Specific implementation circuit 2 of the topology shown;

[0066] Fig.38 The wireless power transmission of the present invention is based on Fig.11 The specific implementation circuit 3 of the topology shown and the schematic diagram of the switch control signal; Fig.38 (a) is a schematic diagram of the circuit; Fig.38 (b) is the switch control signal;

[0067] Fig.39 It is a schematic diagram of a primary multi-winding structure adopted in the wireless power transmission topology of the present invention;

[0068] Fig.40 It is a schematic diagram of the wireless power transmission topology of the present invention adopting the primary multi-winding structure;

[0069] Fig.41 3 is a schematic diagram of the wireless power transmission topology of the present invention using a primary multi-winding structure;

[0070] Fig.42 4 is a schematic diagram of the wireless power transmission topology of the present invention using a primary multi-winding structure;

[0071] Fig.43 Schematic diagram of the wireless power transmission topology of the present invention using a primary multi-winding structure;

[0072] Fig.44 It is the simulation result of the output voltage of the wireless power transmission topology test example of the present invention;

[0073] Fig.45 It is the output voltage waveform of the secondary side receiving module of the wireless power transmission topology test example of the present invention when the coupling coefficient is 0.15;

[0074] Fig.46It is the output voltage waveform of the secondary side receiving module of the wireless power transmission topology test example of the present invention when the coupling coefficient is 0.35;

[0075] Wherein: 1-DC power supply; 2-high frequency inverter; 3-primary multi-frequency common compensation network; 4-primary transmitting coil; 5_1-f1# receiving module, in which the resonant network outputs a voltage or current with a frequency of f1; 5_2-f2# receiving module, in which the resonant network outputs a voltage or current with a frequency of f1; 6-load; 7-control module; 5-receiving module, in which the resonant network outputs a voltage or current with a frequency of f1 or f2 in a time-sharing manner; L p -Self-inductance of the primary transmitting coil, L s -Self-inductance of the secondary transmitting coil, V in -DC input voltage, v AB -High frequency inverter outputs square wave voltage, i 12 -High frequency inverter output current, V AB_1 -The effective value of the fundamental voltage output by the high-frequency inverter, V AB_3 -The effective value of the third harmonic voltage output by the high-frequency inverter, I AB_1 -The effective value of the fundamental current output by the high-frequency inverter, I AB_3 -The effective value of the third harmonic current output by the high-frequency inverter, i p - Current flowing through the transmitting coil, i p1 - Current flowing through the fundamental wave transmitting coil, i p2 - Current flowing through the harmonic transmitting coil, i s1 - Current flowing through the fundamental wave receiving coil, i s2 - Current flowing through the harmonic receiving coil, i1-f1# receiving module rectifier bridge input current, v1-f1# receiving module rectifier bridge input voltage, i2-f2# receiving module rectifier bridge input current, v2-f2# receiving module rectifier bridge input voltage, I p_1 -The effective value of the fundamental current flowing through the transmitting coil, I p_3 -The effective value of the third harmonic current flowing through the transmitting coil, I p1 -The effective value of the current flowing through the fundamental wave transmitting coil, I p2 -The effective value of the current flowing through the harmonic transmitting coil, I s1_1 -The effective value of the fundamental current flowing through the fundamental receiving coil, I s1_3 -The effective value of the third harmonic current flowing through the fundamental wave receiving coil, I s2_1 -The effective value of the fundamental current flowing through the harmonic receiving coil, I s2_3-The effective value of the third harmonic current flowing through the harmonic receiving coil, the effective value of the input current of the rectifier bridge of the receiving module I1-f1#, the effective value of the input voltage of the rectifier bridge of the receiving module V1-f1#, the effective value of the input current of the rectifier bridge of the receiving module I2-f2#, the effective value of the input voltage of the rectifier bridge of the receiving module V2-f2#, ω _1 - Fundamental resonant frequency, ω _3 -Third harmonic resonant frequency,ω _m -mth harmonic resonant frequency, I o -DC output current, V o -DC output voltage, R L - Load resistance. DETAILED DESCRIPTION

[0076] The present invention will be further explained below in conjunction with the accompanying drawings.

[0077] The present invention proposes a wireless power transmission topology with strong anti-offset performance based on multi-frequency energy parallel transmission, which is suitable for different types of magnetic coupling mechanisms and winding forms (such as circular, square, DD and guide rail types, etc.), and aims to achieve decoupling control of multi-frequency energy, avoid the influence of cross-coupling, and improve the misalignment tolerance of the wireless power transmission system.

[0078] Figure 1 The basic circuit composition of the wireless power transmission topology based on multi-frequency energy parallel transmission of the present invention is given. The system structure includes a DC power supply 1, a high-frequency inverter 2, a primary multi-frequency common compensation network 3, a primary transmitting coil 4, a receiving module 5, a load 6, and a control module 7. In terms of connection relationship, the DC power supply 1, the high-frequency inverter 2, the primary multi-frequency common compensation network 3 and the primary transmitting coil 4 are connected in series in sequence, and the control module 7 outputs a driving square wave signal of the high-frequency inverter 2, so that the output voltage of the high-frequency inverter 2 is a mixed superposition of dual-frequency or multi-frequency voltages or currents, and the frequencies are recorded as f1, f2, ... f n , n≥2. The primary side multi-frequency shared compensation network 3 has frequency f1, frequency f2 and...frequency f n Resonance, dual-frequency or multi-frequency currents simultaneously excite the primary transmitting coil 4, and transmit the multi-frequency energy in parallel to the receiving module 5 through magnetic field coupling. The receiving module 5 includes the same number of receiving units as the number of frequencies, each receiving unit includes a frequency selection network and a compensation network, and the frequency selection network, the compensation network and the corresponding receiving coil form a resonant network, which is adjusted to different resonant frequencies to achieve dual-frequency or multi-frequency energy separation and decoupled transmission, and the output frequencies are f1, f2, ... f n The AC voltage or current is rectified and filtered respectively and then combined for output to supply power to the load.

[0079] The high frequency inverter 2 may be a combination of n inverters, such as Figure 2As shown, the input terminals of n inverters are connected in parallel to the DC power supply V in , the control module outputs n driving signals with frequencies of f1, f2, ... f n , n is an integer greater than or equal to 2. Introduce the same number of isolation transformers T r1 , T r2 ,…T rn The square wave voltages output by n inverters are superimposed in series on the secondary side to achieve multi-frequency input of the system. Figure 3 As shown, each inverter is connected in series with an LC resonant network, which resonates at the corresponding excitation frequency, namely Where i=1,2,……,n,L (i) is the inductance value of the LC resonant network in series with the i-th inverter, C (i) is the capacitance value of the LC resonant network in series with the i-th inverter, which converts the output voltage of the corresponding inverter into a current source input. The isolation transformer T is introduced, and the frequencies flowing through the n primary windings are f1, f2, ... and f n The excitation current induces a voltage of the corresponding frequency in the secondary winding, thereby realizing multi-frequency input of the system. Figure 4 As shown, the output of each inverter is connected to the corresponding isolation transformer T ri The series resonant capacitor C (i) ,satisfy Where i=1,2,……,n,L TP (i) Isolation transformer T ri The self-inductance of the primary winding induces a frequency f in the secondary winding. i The secondary side of the isolation transformer is connected in parallel to superimpose the multi-frequency currents in parallel, thereby realizing the multi-frequency input of the system.

[0080] In addition, the output voltage or current waveform of the high-frequency inverter can be modulated into an ideal waveform by controlling the switching frequency and conduction time of each switch tube in the inverter, so that it can be decomposed into the superposition of dual-frequency or multi-frequency voltage or current. Taking the dual-frequency system as an example, Figure 5 As shown, the high-frequency inverter adopts a full-bridge inverter, and the control module 7 outputs square wave voltages with frequencies f1 and f2 respectively, which drive the two bridge arms of the full-bridge inverter respectively. The midpoints of the two bridge arms output square wave voltages with frequencies f1 and f2 respectively. The output voltage of the high-frequency inverter is the voltage difference between the midpoints of the two bridge arms, which can be decomposed into the superposition of two voltages with frequencies f1 and f2.

[0081] like Figure 6 As shown in (a), the control module 7 outputs a square wave signal with a frequency of f to drive the high-frequency inverter 2. The working frequency of the high-frequency inverter is f, and the square wave voltage v with a frequency of f is output. 12 ,like Figure 6 As shown in (b), according to Fourier decomposition, v 12 Expanded into the superposition of the fundamental wave and all n-order odd harmonics, satisfying

[0082]

[0083] where ω _m =mω _1 ,ω _1 =2πf, then the fundamental effective value of the inverter output voltage is V AB_1 And (2n-1)th harmonic effective value V AB_2n-1 for:

[0084]

[0085] The resonant frequencies of the primary multi-frequency common compensation network 3 are f1, f2, ..., f n , where f n =(2n-1)f1, that is, f1 is the fundamental resonance frequency, f n is the (2n-1)th harmonic resonant frequency. In order to illustrate the working principle below, we assume that f1=f. However, it should be noted that in practical applications, in order to obtain the input inductive phase angle and realize soft switching, the operating frequency is usually designed to deviate slightly from the fundamental resonant frequency, allowing a certain deviation between the operating frequency and the resonant frequency.

[0086] According to the superposition theorem, we can Figure 1 to Figure 6 The primary circuit of the wireless power transmission topology of the present invention is equivalent to Figure 7 As shown, the output of the high-frequency inverter 2 is a series superposition of dual-frequency or multi-frequency voltage sources or a parallel superposition of dual-frequency or multi-frequency current sources, which serves as the input of the primary multi-frequency common compensation network 3. The primary multi-frequency common compensation network 3 and the primary transmitting coil 4 form a primary resonant network with resonant frequencies of f1, f2, ..., f n , realizing multi-frequency energy driving the transmitting coil simultaneously.

[0087] In order to highlight the design focus of the present invention and illustrate the working principle, the following takes the wireless power transmission topology of f1 and f2 dual-frequency energy parallel transmission as an example to illustrate the specific composition and parameter design method of the circuit. The derivation of other dual-frequency or multi-frequency topologies is similar to the relevant conclusions and will not be repeated. The basic composition of the circuit is as follows: Figure 8 and Fig. 9 shown. Figure 8 and Fig. 9The primary circuit is the same as that of the transmitter coil, which transmits dual-frequency energy of frequencies f1 and f2 at the same time. The primary multi-frequency shared compensation network 3 and the transmitter coil 4 constitute a primary resonant network, which satisfies that at the frequency f1, the output voltage gain of the resonant network is independent of the equivalent load of the network, and at the frequency f2, the output current gain of the resonant network is independent of the equivalent load of the network; or that at the frequency f1, the output current gain of the resonant network is independent of the equivalent load of the network, and at the frequency f2, the output voltage gain of the resonant network is independent of the equivalent load of the network. The difference is Figure 8 Central edge transmitting coil L p The energy receiving coil L of the receiving unit 5_1 s1 , the energy receiving coil L of the receiving unit 5_2 s2 Constitute a non-contact transformer, M1 is L p With L s1 Mutual inductance between, M2 is L p With L s2 The mutual inductance between s12 For L s1 With L s2 The mutual induction between Fig. 9 Central edge transmitting coil L p The energy receiving coil L of the receiving unit 5_1 s Construct a non-contact transformer, M is L p With L s The mutual inductance between the receiving coil L of the receiving unit 5_2 b With inductance L a And mutual inductance M s The coupled inductor, L s With inductance L a Series connection, L s The dual-frequency energy in the coupling primary excitation coil is coupled to the L b Generate an induced voltage or current with a frequency of f2, and then use the f1 frequency selection network and the f2 frequency selection network to separate and extract the energy of the f1 and f2 frequencies and combine them for output.

[0088] like Fig.10 As shown, the primary multi-frequency shared resonant network 3 can also introduce a switch to achieve LC parameter switching or introduce a switch capacitor or inductor tuning. The control module 7 outputs a drive signal of the switch, switch capacitor or inductor in the primary multi-frequency shared resonant network 3, so that the resonant network resonates at frequency f1 within time t1 and resonates at frequency f2 within time t2, t1+t2≤T, to achieve multi-frequency energy time-sharing transmission. Figure 8 Similarly, the receiving unit 5_1 outputs a voltage or current with a frequency of f1 within a time period of t1, and the receiving unit 5_2 outputs a voltage or current with a frequency of f2 within a time period of t2. It should be noted that the secondary receiving module here can also adopt Fig. 9 The circuit structure is shown. Fig.11 Another implementation of the secondary side receiving module is given, including a receiving coil L s , multi-frequency shared compensation network, rectifier filter unit, omitting the frequency selection network, similar to the primary multi-frequency shared compensation network 3, the secondary multi-frequency shared compensation network also adopts LC high-order compensation topology, and the receiving coil L s A resonant network is formed with resonant frequencies of f1 and f2. i The output frequency during this time is f i The AC signal is connected to the load after rectification and filtering, where 1≤i≤n. Conversely, Fig.11 It is also possible to introduce switches in the secondary-side multi-frequency shared compensation network to implement LC parameter switching or introduce switch capacitor or inductor tuning. The primary-side multi-frequency shared compensation network adopts an LC high-order compensation topology.

[0089] Fig.12 , Fig.13 The present invention provides two other circuit implementations of the wireless power transmission topology based on multi-frequency energy parallel transmission using multi-frequency energy time-division multiplexing. Fig.10 , Fig.11 The difference lies in the different implementation of the primary circuit. The control module 7 outputs square wave signals with frequencies f1 and f2 in a time-sharing manner within a period T, wherein the length of the square wave signal with frequency f1 is t1, and the length of the square wave signal with frequency f2 is t2, satisfying t1+t2≤T, thereby realizing time-sharing multiplexing, and realizing output regulation by adjusting the time ratio of t1 and t2.

[0090] around Figure 8 to Figure 13 The following are five basic circuit structures shown in the figure. The following specific embodiments are given to more clearly illustrate the technical solution of the present invention. Considering that the two receiving coils in the secondary double winding structure realize decoupling control, the following embodiments do not consider the mutual inductance M s12 , and cannot be used to limit the protection scope of the present invention.

[0091] Figure 14 to Figure 27 The wireless power transmission topology based on multi-frequency energy parallel transmission of the present invention is given. Figure 8 The specific circuit implementation of the structural form shown, Figure 28 to Figure 32 The wireless power transmission topology based on multi-frequency energy parallel transmission of the present invention is given. Fig. 9 The specific circuit implementation of the structural form shown, Figure 33 to Figure 35 The wireless power transmission topology based on multi-frequency energy parallel transmission of the present invention is given. Fig.13 The specific circuit implementation of the structural form shown, Figure 36 to Figure 38 The wireless power transmission topology based on multi-frequency energy parallel transmission of the present invention is given. Fig.10The specific circuit implementation of the structural form shown. Figure 33 to Figure 38 The purpose is to illustrate the specific circuit implementation and parameter design method when the wireless power transmission topology of the present invention adopts the time division multiplexing method.

[0092] In the following embodiments, unless otherwise specified, the frequencies f1 and f2 are taken as the fundamental wave and the third harmonic by default, and the derivation of the other frequencies is similar to the relevant conclusions. The operating frequency is f, and a certain deviation is allowed between f and f1. The resonant frequency satisfies:

[0093] ω _1 =2πf1,ω _3 =2πf2=6πf1 (3)

[0094] Fig.14 (a) The present invention is based on Figure 8 A specific circuit topology, the primary multi-frequency shared compensation network is composed of inductor L1 and capacitor C, L1, C and primary excitation coil L p To form a T-type resonant network, such as Fig.13 As shown in (b), v p 、i p The output voltage and current of the resonant network are respectively. The secondary side of the non-contact transformer is a double-winding structure, and the f1# frequency selection network consists of capacitor C sx1 With inductance L sx1 The f1# compensation network is an LCC compensation network, which consists of capacitor C s1 , C s2 With inductance L sr1 Composition, receiving coil L s1 Connected in series with the f1# frequency selection network; the f2# frequency selection network consists of capacitor C sx2 With inductance L sx2 The f2# compensation network is a series compensation network. s3 To compensate the capacitance, the receiving coil L s2 The output of f1# receiving module 5_1 and f2# receiving module 5_2 are connected in parallel to the load R L powered by.

[0095] The parameters of the primary side multi-frequency shared compensation network meet the following requirements:

[0096]

[0097] Then the output current effective value I at the fundamental frequency can be obtained p_1 And the output voltage effective value V at the third harmonic frequency p_3 They are:

[0098]

[0099] It can be seen that the resonant frequency is ω _1 When the output current of the resonant network is independent of the equivalent load of the network, the resonant frequency is ω _3 When , the output voltage of the resonant network has nothing to do with the equivalent load of the network.

[0100] The parameters of the frequency selection network components meet the following requirements:

[0101]

[0102] Impedance Z of f1# frequency selection network and f2# frequency selection network x1_m With Z x2_m They are:

[0103]

[0104] where ω _m is the mth harmonic frequency, ω _m =mω _1 . Due to L sx1 , C sx1 With L sx2 , C sx2 They are all parallel resonances, and the impedance is infinite at the resonant frequency, which hinders the transmission of the third harmonic in the f1# receiving module and the fundamental wave in the f2# receiving module, thereby realizing energy decoupling transmission in the secondary side network.

[0105] The parameters of the secondary resonant components meet the following requirements:

[0106]

[0107] According to the basic circuit theory, the effective value I1 of the input current of the rectifier bridge of the f1# receiving module and the effective value I2 of the input current of the rectifier bridge of the f2# receiving module can be deduced as follows:

[0108]

[0109] DC output current I o for:

[0110]

[0111] It can be seen that the output current is independent of the load, and constant current output can be achieved under variable load conditions. In addition, the output current of the f1# receiving module changes in direct proportion to the value of the mutual inductance M1, and the output current of the f2# receiving module changes in inverse proportion to the value of the mutual inductance M2. When the output current of the f1# receiving module and the output current of the f2# receiving module change in the offset condition, that is, when they increase or decrease at the same time, the output current change of the f1# receiving module and the output current change of the f2# receiving module can offset each other to a certain extent, reducing the sensitivity of the output to the offset. Taking the increase of the air gap as an example, both M1 and M2 decrease, the corresponding I1 decreases, and I2 increases. By reasonably designing the parameters of the transformer and the resonant element, the increase of I2 compensates for the decrease of I1, then a constant current output can be achieved within a certain offset condition, improving the anti-offset characteristics of the wireless power transmission system.

[0112] Fig.14 The primary side multi-frequency shared compensation network 3 can also be used Figure 15 to Figure 19 The primary side multi-frequency shared compensation network shown in the figure is designed Figure 15 to Figure 19 The parameters of the primary multi-frequency shared compensation network in are shown in Table 1, making the resonant frequency ω _1 When the output current of the resonant network is independent of the equivalent load of the network, the resonant frequency is ω _3 When , the output voltage of the resonant network is independent of the equivalent load of the network. Based on equations (9) and (10), we can further deduce that Figures 15 to 20 The external characteristics of Figure 15 to Figure 19 The circuit topology shown has Fig.14 The external characteristics of the circuit topology shown are similar, that is, the output current of the f1# receiving module changes in direct proportion to the value of the mutual inductance M1, and the output current of the f2# receiving module changes in inverse proportion to the value of the mutual inductance M2. Under the offset condition, the change trends of M1 and M2 are the same, that is, when they increase or decrease at the same time, the output current changes of the f1# receiving module and the output current changes of the f2# receiving module can offset each other to a certain extent, reducing the sensitivity of the output to the offset. It should be noted that compared with Fig.14 , Figure 15 to Figure 19 The primary side multi-frequency shared compensation network has multiple adjustable free variables. Fig.15 For example, compared to Fig.14 , Fig.15 A compensation capacitor C2 is introduced, and the soft switching of the high-frequency inverter can be ensured by adjusting the capacitance of the compensation capacitor C2.

[0113] Table 1: Figure 15 to Figure 19 Compensation parameters of primary-side multi-frequency shared compensation network and output DC current

[0114]

[0115] also, Fig.14The receiving module 5 in can also be Figure 20 to Figure 22 The circuit topology is given in . The following describes the circuit parameter values ​​and working principles respectively:

[0116] like Fig. 20 As shown, the secondary side of the non-contact transformer is a double-winding structure, and the f1# frequency selection network consists of a capacitor C sx1 With inductance L sx1 The f1# compensation network is a series compensation network. s3 To compensate the capacitance, the receiving coil L s1 Connected in series with the f1# frequency selection network; the f2# frequency selection network consists of capacitor C sx2 With inductance L sx2 The f2# compensation network is an LCC compensation network, which consists of capacitor C s1 , C s2 With inductance L sr1 Composition, receiving coil L s2 The output of f1# receiving module 5_1 and f2# receiving module 5_2 are connected in series to the load R L Power supply. The parameters of the frequency selection network components meet the following requirements:

[0117]

[0118] Impedance Z of f1# frequency selection network and f2# frequency selection network x1_m With Z x2_m They are:

[0119]

[0120] where ω _m is the mth harmonic frequency, ω _m =mω _1 . Due to L sx1 , C sx1 With L sx2 , C sx2 They are all parallel resonances, and the impedance is infinite at the resonant frequency, which hinders the transmission of the third harmonic in the f1# receiving module and the fundamental wave in the f2# receiving module, thereby realizing energy decoupling transmission in the secondary side network.

[0121] The parameters of the secondary resonant components meet the following requirements:

[0122]

[0123] According to the basic circuit theory, the DC output voltage V o for:

[0124]

[0125] It can be seen that the output voltage is independent of the load, and constant voltage output can be achieved under variable load conditions. In addition, the output voltage of the f1# receiving module changes in direct proportion to the value of the mutual inductance M1, and the output voltage of the f2# receiving module changes in inverse proportion to the value of the mutual inductance M2. Under offset conditions, the change trends of M1 and M2 are the same, that is, when they increase or decrease at the same time, the output voltage change of the f1# receiving module and the output voltage change of the f2# receiving module can offset each other to a certain extent, reducing the sensitivity of the output to offset. Taking the increase of the air gap as an example, both M1 and M2 decrease, the corresponding V1 decreases, and V2 increases. By reasonably designing the parameters of the transformer and resonant elements, the increase in V2 compensates for the decrease in V1, then a constant voltage output can be achieved within a certain offset condition, improving the anti-offset characteristics of the wireless power transmission system.

[0126] like Fig.21 As shown, the secondary side of the non-contact transformer is a double-winding structure, and the f1# frequency selection network consists of a capacitor C sx1 With inductance L sx1 The f1# compensation network is an LCC compensation network, which consists of capacitor C s1 With inductance L sr1 Composition, receiving coil L s1 With C s1 After the series connection, it is connected in parallel with the f1# frequency selection network, and then connected in series with L sr1 ; f2# frequency selection network consists of capacitor C sx2 With inductance L sx2 The f2# compensation network is a series compensation network. s3 To compensate the capacitance, the receiving coil L s2 The output of f1# receiving module 5_1 and f2# receiving module 5_2 are connected in parallel to the load R L Power supply. The parameters of the frequency selection network components meet the following requirements:

[0127]

[0128] Impedance Z of f1# frequency selection network and f2# frequency selection network x1_m With Z x2_m They are:

[0129]

[0130] where ω _m is the mth harmonic frequency, ω _m =mω _1 . Due to L sx2 With C sx2 Parallel resonance, the impedance is infinite at the resonant frequency, which hinders the transmission of fundamental wave energy in the f2# receiving module. sx1 With C sx1In series resonance, the impedance is zero at the resonant frequency, providing a low-resistance loop for the third harmonic current, bypassing the third harmonic energy in the fundamental wave receiving coil, thereby realizing the separation and decoupling transmission of the fundamental wave and harmonic energy in the secondary network.

[0131] The parameters of the secondary resonant components meet the following requirements:

[0132]

[0133] According to the basic circuit theory, the DC output current I can be deduced o for:

[0134]

[0135] It can be seen that the output current expression is the same as Fig.14 The output current expression is the same as Fig.14 Have similar external characteristics.

[0136] like Fig. 22 As shown, the secondary side of the non-contact transformer is a double-winding structure, and the f1# frequency selection network consists of a capacitor C sx1 With inductance L sx1 The f1# compensation network is a series compensation network. s3 To compensate the capacitance, the receiving coil L s1 Connected in series with the f1# frequency selection network; the f2# frequency selection network consists of capacitor C sx2 With inductance L sx2 The f2# compensation network is an LCC compensation network, which consists of capacitor C s1 , C s2 With inductance L sr1 Composition, receiving coil L s2 With C s1 After connecting in series with f2# frequency selection network, capacitor C s2 Connect in parallel, then connect in series L sr1 The outputs of f1# receiving module block 5_1 and f2# receiving module block 5_2 are connected in series to provide a load R L Power supply. The parameters of the frequency selection network components meet the following requirements:

[0137]

[0138] Impedance Z of f1# frequency selection network and f2# frequency selection network x1_m With Z x2_m They are:

[0139]

[0140] where ω _m is the mth harmonic frequency, ω _m =mω_1 . Due to L sx1 With C sx1 Parallel resonance, the impedance is infinite at the resonant frequency, which hinders the transmission of harmonic energy in the f1# receiving module. sx2 With C sx2 Series resonance, the impedance is zero at the resonant frequency, providing a low-resistance loop for the fundamental current in the f2# receiving module, bypassing the fundamental energy, thereby achieving separation and decoupling transmission of fundamental and harmonic energy in the secondary network.

[0141] The parameters of the secondary resonant components meet the following requirements:

[0142]

[0143] According to the basic circuit theory, the DC output voltage V o for:

[0144]

[0145] Output voltage expression and Fig. 20 The output voltage expression is the same as Fig. 20 With similar external characteristics, by reasonably designing the parameters of the transformer and resonant components, a constant voltage output can be achieved within a certain offset condition, thus improving the anti-offset characteristics of the wireless power transmission system.

[0146] akin, Figure 15 to Figure 19 The receiving module in can also use Figure 20 to Figure 22 The circuit form in has similar output characteristics and will not be described here.

[0147] Figure 23 to Figure 27 The present invention is based on Figure 8 The other five circuit implementation forms of the secondary side receiving module topology are all the same Fig.13 The receiving module 5 given in the above is the same, and the parameter design is also the same, which will not be repeated here. The difference is Fig.13 The output of the f1# receiving module is connected in parallel with the output of the f2# receiving module to supply power to the load. Figure 23 to Figure 27 The output of the f1# receiving module is connected in series with the output of the f2# receiving module to supply power to the load. Fig.23 Take the circuit composition, resonant network parameter selection and working principle as an example:

[0148] like Fig.23 As shown, the primary multi-frequency shared compensation network consists of inductor L1, capacitor C and capacitor C2. L1, C, C2 and the primary excitation coil L p Form a T-type resonant network, and Fig.15 The primary multi-frequency shared compensation network in is the same, but the resonant component parameters are different. Here, the parameters of the primary multi-frequency shared compensation network satisfy:

[0149]

[0150] The effective value of the output current I at the harmonic frequency can be obtained p_3 And the output voltage RMS value V at the fundamental frequency p_1 They are:

[0151]

[0152] The resonant frequency is ω _3 When the output current of the resonant network is independent of the equivalent load of the network, the resonant frequency is ω _1 When , the output voltage of the resonant network has nothing to do with the equivalent load of the network.

[0153] Further, from formula (8), according to the circuit, it can be deduced that the effective value V1 of the input voltage of the rectifier bridge of the f1# receiving module and the effective value V2 of the input voltage of the rectifier bridge of the f2# receiving module are respectively:

[0154]

[0155] DC output voltage V o for:

[0156]

[0157] It can be seen that the output voltage is independent of the load, and constant voltage output can be achieved under variable load conditions. In addition, the output voltage of the f1# receiving module changes inversely with the value of the mutual inductance M1, and the output voltage of the f2# receiving module changes in direct proportion to the value of the mutual inductance M2. When the output voltage of M1 and M2 change in the same trend under the offset condition, that is, when they increase or decrease at the same time, the output voltage change of the f1# receiving module and the output voltage change of the f2# receiving module can offset each other to a certain extent, reducing the sensitivity of the output to the offset.

[0158] Similarly, we can design Figure 24 to Figure 27 The parameters of the primary compensation network in make the resonant frequency ω _3 When the output current of the resonant network is independent of the equivalent load of the network, the resonant frequency is ω _1 When the output voltage of the resonant network is independent of the equivalent load of the network. Table 2 lists Figure 24 to Figure 27 The primary compensation network parameter values ​​and DC output voltage can be seen Figure 24 to Figure 27 The circuit topology shown has Fig.23 The external characteristics of the circuit topology shown are similar. It should be noted that here Fig. 27 The dual frequency here does not refer to the fundamental frequency and harmonic frequency. High-frequency inverters can use Figure 5The circuit topology is given in , and the driving frequency given by the control module 7 satisfies f2≈1.3f1.

[0159] Table 2 Figure 24 to Figure 27 Compensation parameters of primary-side multi-frequency shared compensation network and output DC current

[0160]

[0161] Figure 28 to Figure 32 The wireless power transmission topology based on multi-frequency energy parallel transmission of the present invention is given. Fig. 9 The specific circuit implementation of the structure shown in the figure, in which the primary side multi-frequency compensation network adopts Fig.18 In the circuit structure, the primary resonant network outputs the effective value of the current I at the fundamental frequency. p_1 And the output voltage effective value V at the third harmonic frequency p_3 They are:

[0162]

[0163] It can be seen that the resonant frequency is ω _1 When the output current of the resonant network is independent of the equivalent load of the network, the resonant frequency is ω _3 When , the output voltage of the resonant network is independent of the equivalent load of the network. Correspondingly, a fundamental induced voltage ω is generated on the secondary winding that is independent of the load. _1 MI p_1 , which changes in direct proportion to the mutual inductance M; at the same time, a harmonic induced current V that is independent of the load will be generated on the secondary winding p_3 / (ω _3 M), which changes inversely with the mutual inductance M. Figure 28 to Figure 32 The induced current on the secondary winding is converted into voltage through coupled inductance, and then the frequency selection network is used to separate the harmonic energy to obtain an output voltage or current that changes inversely with the mutual inductance M; at the same time, the f1# frequency selection network separates the fundamental wave energy to obtain an output voltage or current that changes in direct proportion to the mutual inductance M. The fundamental wave and harmonic energy are decoupled and transmitted and then combined for power supply to obtain an output voltage or current that changes non-monotonic with M, which can effectively reduce the output fluctuation under offset conditions and improve the anti-offset characteristics. It should be noted that Figure 28 to Figure 32 The primary side multi-frequency compensation network 3 in can also be used Figure 14 to Figure 17 , Fig.19 as well as Figure 23 to Figure 27 The circuit structure in .

[0164] like Fig.28 As shown, the contactless transformer adopts a primary single winding and a secondary single winding structure. s At the same time, it picks up the dual-frequency energy of the fundamental wave f1 and the harmonic f2, and the primary side excitation winding L pThe mutual inductance between them is M. Inductance L a 、Inductance L b Mutual inductance M s The f1# frequency selection network consists of capacitor C sx1 With inductance L sx1 The f1# compensation network is a T-type compensation network consisting of an inductor L a , L sr1 With capacitor C s1 , C s2 Composition, L s , L a With C s1 After the series connection, it is connected in parallel with the f1# frequency selection network. sr2 Connect in parallel at both ends of the f1# frequency selection network, and then connect to L sr1 The f2# frequency selection network consists of capacitor C sx2 With inductance L sx2 The f2# compensation network is a series compensation network, and the compensation capacitor is C s3 , L b The output of f1# receiving module 5_1 and f2# receiving module 5_2 are connected in series to the load R L Power supply. The parameters of the frequency selection network components meet the following requirements:

[0165]

[0166] Impedance Z of f1# frequency selection network and f2# frequency selection network x1_m With Z x2_m They are:

[0167]

[0168] where ω _m is the mth harmonic frequency, ω _m =mω _1 . Due to L sx2 With C sx2 Parallel resonance, the impedance is infinite at the resonant frequency, which hinders the transmission of fundamental wave energy in the f2# receiving module. sx1 With C sx1 In series resonance, the impedance is zero at the resonant frequency, providing a low-resistance loop for the third harmonic current, bypassing the third harmonic energy in the fundamental wave receiving coil, thereby realizing the separation and decoupling transmission of the fundamental wave and harmonic energy in the secondary network.

[0169] The parameters of the secondary resonant components meet the following requirements:

[0170]

[0171] According to the basic circuit theory, the DC output voltage Vo for:

[0172]

[0173] It can be seen that the output voltage is independent of the load, and constant voltage output can be achieved under variable load conditions. In addition, the output voltage of the f1# receiving module changes in direct proportion to the value of the mutual inductance M, and the output voltage of the f2# receiving module changes in inverse proportion to the value of the mutual inductance M. The output voltage changes non-monotonic with M, which can effectively reduce output fluctuations. Furthermore, by reasonably designing the parameters of the transformer and the resonant element, the system can still have a stable output voltage under constant input conditions when an offset occurs, thereby improving the system's anti-offset capability.

[0174] like Fig.29 As shown, the f1# frequency selection network consists of capacitor C sx1 With inductance L sx1 The f1# compensation network is a T-type compensation network consisting of an inductor L a , L sr1 With capacitor C s1 , C s2 Composition, L s , L a With C s1 After the series connection, it is connected in parallel with the f1# frequency selection network. sr2 Connect in parallel at both ends of the f1# frequency selection network, and then connect to L sr1 The f2# frequency selection network consists of capacitor C sx2 With inductance L sx2 The f2# compensation network consists of a capacitor C s3 , C s4 Composition, inductance L b Connected in parallel with f2# frequency selection network and f2# compensation network. a 、Inductance L b Mutual inductance M s The outputs of the f1# receiving module 5_1 and the f2# receiving module 5_2 are connected in series to provide a load R L Power supply. The parameters of the frequency selection network components meet the following requirements:

[0175]

[0176] Impedance Z of f1# frequency selection network and f2# frequency selection network x1_m With Z x2_m They are:

[0177]

[0178] where ω _m is the mth harmonic frequency, ω _m =mω_1 .L sx2 With C sx2 , L sx1 With C sx1 They are all series resonant, with zero impedance at the resonant frequency, providing a low-resistance loop for the third harmonic current in the f1# receiving module and the fundamental current in the f2# receiving module, thereby achieving separation and decoupling transmission of fundamental and harmonic energy in the secondary network.

[0179] The parameters of the secondary resonant components meet the following requirements:

[0180]

[0181] According to the basic circuit theory, the DC output voltage V o for:

[0182]

[0183] and Fig.28 The output voltage expression of the wireless power transmission system is the same and has similar external characteristics. By reasonably designing the parameters of the transformer and resonant components, a constant voltage output can be achieved within a certain offset condition, thus improving the anti-offset characteristics of the wireless power transmission system.

[0184] like Fig.30 As shown, the circuit structure of f1# receiving module and f2# receiving module is the same as Fig.29 Similar to similar, the difference is that the inductor L sr2 Replace with capacitor C s2 , the capacitor C s3 Replace with inductor L sr2 , the output of f1# receiving module 5_1 and f2# receiving module 5_2 are connected in parallel to provide a load R L Power supply. The secondary side resonant component parameters meet:

[0185]

[0186] in, According to the basic circuit theory, the DC output current I can be deduced o for:

[0187]

[0188] It can be seen that the output current is independent of the load, constant current output can be achieved under variable load conditions, and the output current changes non-monotonicly with M. By reasonably designing the parameters of the transformer and resonant components, constant current output can be achieved within a certain offset condition, thereby improving the anti-offset characteristics of the wireless power transmission system.

[0189] like Fig.31 As shown, the frequency selection network and Fig.29Similarly, the f1# compensation network consists of an inductor L a , L sr1 With capacitor C s1 Composition, L s , L a With C s1 After the series connection, it is connected in parallel with the f1# frequency selection network, and then connected to L sr1 The f2# compensation network consists of a capacitor C s2 , C s3 Composition, harmonic receiving coil L b With C s2 After the series connection, it is connected in parallel with the f2# frequency selection network, and then connected to C s3 The output of f1# receiving module 5_1 and f2# receiving module 5_2 are connected in parallel to provide a load R L Power supply. The secondary side resonant component parameters meet:

[0190]

[0191] According to the basic circuit theory, the DC output current I o for:

[0192]

[0193] It can be seen that the output current is independent of the load, constant current output can be achieved under variable load conditions, and the output current changes non-monotonicly with M. By reasonably designing the parameters of the transformer and resonant components, constant current output can be achieved within a certain offset condition, thereby improving the anti-offset characteristics of the wireless power transmission system.

[0194] like Fig.32 As shown, the circuit structure of f1# receiving module is similar to Fig.30 The f1# receiving module circuit is the same as that of the Fig.28 The f2# frequency selection network is the same as the f2# compensation network, and the f2# compensation network consists of a capacitor C s3 and inductor L sr2 Composition, harmonic receiving coil L b After being connected in series with the f2# frequency selection network, it is connected with the capacitor C s3 In parallel, and then with the inductor L sr2 The output of f1# receiving module 5_1 and f2# receiving module 5_2 are connected in parallel to provide a load R L Power supply. The secondary side resonant component parameters meet:

[0195]

[0196] in, According to the basic circuit theory, the DC output current I o for:

[0197]

[0198] It can be seen that the output current is independent of the load, constant current output can be achieved under variable load conditions, and the output current changes non-monotonicly with M. By reasonably designing the parameters of the transformer and resonant components, constant current output can be achieved within a certain offset condition, thereby improving the anti-offset characteristics of the wireless power transmission system.

[0199] Fig.33 The wireless power transmission topology based on multi-frequency energy parallel transmission of the present invention is given. Fig.13 The specific circuit implementation of the structural form shown in the figure. The control module 7 outputs square wave signals of frequencies f1 and f2 in time sharing within the period T, wherein the length of the square wave signal of frequency f1 is t1, and the length of the square wave signal of frequency f2 is t2, satisfying t1+t2≤T, realizing time sharing multiplexing, and realizing output regulation by adjusting the time ratio of t1 and t2. The non-contact transformer adopts a primary single winding and a secondary single winding structure, and the frequency f1 energy transmission channel and the frequency f2 energy transmission channel share the primary compensation network 3, the transmitting coil L p , receiving coil L s , secondary side compensation network and rectifier filter network, and no frequency selection network is required. Fig.33 Primary circuit and Fig.18 The primary circuit is the same as in the example above. The secondary multi-frequency common compensation network adopts the LCC compensation network structure. The receiving coil L s With capacitor C s1 After being connected in series, with the inductor L sr1 Connect in parallel, then in series with capacitor C s2 The secondary side resonant network parameters satisfy:

[0200]

[0201] According to the basic circuit theory, it can be deduced that the effective value V1 of the input voltage of the rectifier bridge of the receiving module at the frequency f1 and the effective value V2 of the input voltage of the rectifier bridge of the receiving module at the frequency f2 are respectively:

[0202]

[0203] The resonant network outputs an AC voltage with a frequency of f1 during time t1 and an AC voltage with a frequency of f2 during time t2. After being rectified and filtered, it is connected to the load. The DC output voltage V o for:

[0204]

[0205] It can be seen that the output voltage is independent of the load, and constant voltage output can be achieved under variable load conditions, and the output voltage changes non-monotonicly with M. By reasonably designing the parameters of the transformer and resonant components, constant voltage output can be achieved within a certain offset condition, improving the anti-offset characteristics of the wireless power transmission system. At the same time, the time ratio of t1 and t2 can be adjusted to achieve output regulation.

[0206] It should be noted that since the secondary side multi-frequency shared compensation network and the primary side multi-frequency shared compensation network are both at frequencies f1, f2, ..., f n The lower resonance and similar characteristics, so the secondary multi-frequency shared compensation network in this embodiment can also be used Figure 13 to Figure 27 The primary side multi-frequency shared compensation network topology in the figure is different in that the secondary side multi-frequency shared compensation network requires that at multiple resonant frequencies, the output voltage of the resonant network is independent of the equivalent load of the network, or the output current is independent of the equivalent load of the network, which will not be elaborated here. Fig.34 and Fig.35 Two examples are given, Fig.34 The middle and secondary side multi-frequency shared compensation network and Fig.24 The multi-frequency sharing compensation network in the central and southern regions is symmetrical. Fig.35 The middle and secondary side multi-frequency shared compensation network and Fig.23 The central and peripheral multi-frequency sharing compensation network is symmetrical. Fig.33 The primary side multi-frequency shared compensation network can also be used Figure 13 to Figure 27 The primary side multi-frequency shared compensation network topology and compensation parameter design are the same.

[0207] Figure 14 to Figure 35 The primary-side multi-frequency shared compensation network is designed with fixed parameters to achieve resonance at frequencies f1 and f2. It can also achieve resonance at frequencies f1 and f2 by parameter switching or switching capacitor / inductor network tuning, which also falls within the protection scope of the present invention. Figure 36 to Figure 38 The wireless power transmission topology based on multi-frequency energy parallel transmission of the present invention is given. Fig.11 A specific circuit implementation example of the structural form shown.

[0208] like Fig.36 As shown, the control module 7 outputs square wave signals with frequencies f1 and f2 in time sharing within the period T, wherein the length of the square wave signal with frequency f1 is t1, and the length of the square wave signal with frequency f2 is t2, satisfying t1+t2≤T, thus realizing time sharing multiplexing. The primary multi-frequency shared compensation network is composed of an inductor L1, capacitors C1, C2 and a switch tube K. The switch tube K is connected in series with the capacitor C1, and the capacitor C2 is connected to the transmitting coil L p Series connection, L p After the branch is connected in parallel with the capacitor C1, it is connected in series with the inductor L1. The parameters of the resonant components of the secondary multi-frequency shared compensation network meet the following requirements:

[0209]

[0210] The primary switch tube K is disconnected, and the primary multi-frequency shared compensation network 3 is equivalent to series compensation. The switch tube K is closed, and the primary multi-frequency shared compensation network 3 is equivalent to LCL compensation. The parameters of the resonant components meet the following requirements:

[0211]

[0212] Output voltage effective value V at fundamental frequency p_1 And the output current effective value I at the third harmonic frequency p_3 They are:

[0213]

[0214] It can be seen that when the switch tube K is disconnected, the resonant frequency is ω _1 , at this time, the output voltage of the resonant network has nothing to do with the equivalent load of the network; when the switch tube K is closed, the resonant frequency is ω _3 , at this time, the output current of the resonant network has nothing to do with the equivalent load of the network. The control module 7 controls the switch tube K to close within time t2, then the resonant network outputs an AC current with a frequency of f1 within time t1, and an AC current with a frequency of f2 within time t2, which is connected to the load after rectification and filtering. According to the basic circuit theory, the DC output current I can be deduced o for

[0215]

[0216] The output current is independent of the load, and constant current output can be achieved under variable load conditions, and the output current changes non-monotonically with M. By reasonably designing the parameters of the transformer and resonant components, constant current output can be achieved within a certain offset condition, improving the anti-offset characteristics of the wireless power transmission system, and at the same time, the time ratio of t1 and t2 can be adjusted to achieve output regulation. It should be noted that the secondary multi-frequency shared compensation network can also introduce a similar switch tube K to switch the compensation parameters.

[0217] like Fig.37 As shown, the primary multi-frequency shared compensation network consists of inductor L1, capacitor C1, and transmitting coil L p It is connected in parallel with the capacitor C1 and then connected in series with the inductor L1. The inductor L1 is an adjustable inductor, and the inductance value can be dynamically adjusted. Similarly, the capacitor C1 can also be in the form of a switched capacitor. Fig.37 The purpose is to illustrate that the wireless power transmission topology based on multi-frequency energy parallel transmission of the present invention can also introduce switched capacitor or inductor tuning. Fig.36 The parameters of the primary resonant element satisfy:

[0218]

[0219] Among them L1 (1) is the equivalent inductance value of L1 during time t1, L1 (2) is the equivalent inductance of L1 during time t2, correspondingly, the effective value of the output voltage V at frequency f1 p_1 and the output current effective value I at the f2 harmonic frequency p_3 They are:

[0220]

[0221] It can be seen that the resonant frequency during the t1 period is ω _1 , at this time the output voltage of the resonant network has nothing to do with the equivalent load of the network; the resonant frequency during time t2 is ω _3 , at this time, the output current of the resonant network has nothing to do with the equivalent load of the network. The control module (7) controls the inductance value of the inductor L1 to be L1 within the time t1 (1) , the induction value is L2 within t2 time (1) , the resonant network outputs an AC current with a frequency of f1 during time t1, and an AC current with a frequency of f2 during time t2. After being rectified and filtered, it is connected to the load. According to the basic circuit theory, the DC output current I can be deduced. o for

[0222]

[0223] The output current is independent of the load, and constant current output can be achieved under variable load conditions, and the output current changes non-monotonically with M. By reasonably designing the parameters of the transformer and resonant components, constant current output can be achieved within a certain offset condition, improving the anti-offset characteristics of the wireless power transmission system, and the time ratio of t1 and t2 can be adjusted to achieve output regulation. Similarly, the secondary multi-frequency shared compensation network can also introduce switch capacitor or inductor tuning.

[0224] like Fig.38 As shown in (a), the control module outputs a square wave signal with a frequency of f as the driving signal of the high-frequency inverter. The primary multi-frequency shared compensation network consists of inductors L1, L2, capacitors C1, C2, C3 and switches K1, K2. The transmitting coil L p Connected in series with capacitor C3, capacitor C2 and switch K2 are connected in series, capacitor C1 and switch K1 are connected in series and then connected in series with inductor L2, L p The series branch is connected in parallel with the C2 series branch and then in series with L2.

[0225] The control signals of switches K1 and K2 are as follows: Fig.38 As shown in (b), switches K1 and K2 are turned on complementary, and within Δt1, v g(K1) is positive, K1 is closed and K2 is open; during Δt2, v g (K2) is positive, K1 is open and K2 is closed. The parameters of the primary resonant element meet the following conditions:

[0226]

[0227] During the Δt1 time, the primary multi-frequency shared compensation network is equivalent to series compensation, and the resonant frequency is f1; during the Δt2 time, the primary multi-frequency shared compensation network is equivalent to LCL compensation, and the resonant frequency is f2, and multi-frequency energy is transmitted in time. According to the basic circuit theory, the output voltage effective value V at the frequency f1 can be obtained p_1 The output current effective value I at the frequency f2 p_3 They are:

[0228] I p_3 =ω _3 C2V AB_3 =ω _1 C2V AB_1 , V p_1 =V AB_1 (93)

[0229] You can see ω _1 The output voltage of the resonant network at the resonant frequency has nothing to do with the equivalent load of the network; ω _3 At the resonant frequency, the output current of the resonant network has nothing to do with the equivalent load of the network. Furthermore, the DC output current I o for:

[0230]

[0231] It can be seen that the output current is independent of the load, and constant current output can be achieved under variable load conditions, and the output current changes non-monotonicly with M. By reasonably designing the parameters of the transformer and resonant components, constant current output can be achieved within a certain offset condition, improving the anti-offset characteristics of the wireless power transmission system, and the time ratio of t1 and t2 can be adjusted to achieve output regulation. Similarly, the secondary multi-frequency shared compensation network can also introduce switch capacitor or inductor tuning.

[0232] The wireless power transmission topology of the multi-frequency energy parallel transmission of the present invention can also adopt a primary multi-winding structure. Here, taking the primary double-winding structure as an example, several circuit structure examples are given, such as Figure 39 to Figure 43 As shown. Where L p1 , L p2 are the self-inductance of the two primary coils, L s1 , L s2 is the self-inductance of the two receiving coils on the secondary side, M1 is L p1 With L s1 Mutual inductance between, M2 is Lp2 With L s2 The mutual inductance between them is defined as M s12 For L s1 With L s2 The mutual inductance between p12 For L p1 With L p2 The DC power supply 1, high frequency inverter 2 and resonant network are connected in series in sequence; the output end of f1# receiving module 5_1 is connected in series or in parallel with the output of f2# receiving module 5_2 and then connected to the load. It should be noted that the two transmitting coils L p1 , L p2 To achieve decoupling control, the coil structure and connection relationship of the two primary windings can be set so that the transmitting coil L p1 With the transmitting coil L p2 The relative relationship of the magnetic flux coupling of each part is inconsistent, the forward coupling and reverse coupling exist at the same time, and the transmitting coil L p1 The current in the transmitting coil L p2 The magnetic flux generated in each part cancels out, making L p1 Current in L p2 The algebraic sum of the magnetic flux generated in is close to zero, and the mutual inductance M p12 Close to zero; also can be in L p1 With L p2 In the circuit, the f1# frequency selection network and the f2# frequency selection network are connected in series to realize the electrical decoupling control of the fundamental wave and the harmonics. Similarly, the two coils of the mutual inductance also realize the decoupling control. Figure 28 to Figure 32 , the secondary side can also adopt a single winding structure here, which will not be repeated.

[0233] like Fig.39 As shown, the two primary transmitting coils L p1 , L p2 Each of them has an independent compensation network, where L p1 LCC compensation is adopted, and the compensation network is composed of inductors L1, C1, and C2; L p2 Series capacitor compensation is adopted, and the compensation network is capacitor C3. The input ends of the two-part compensation network are connected in parallel to the output end of the high-frequency inverter. p1 , L p2 There are also f1# frequency selection networks (L x1 With C x1 parallel) and f2# frequency selection network (L x2 With C x2 In parallel), the frequency selection network component parameters meet:

[0234]

[0235] Impedance Z of f1# frequency selection network and f2# frequency selection networkx1_m With Z x2_m They are:

[0236]

[0237] where ω _m is the mth harmonic frequency, ω _m =mω _1 . Due to L x1 With C x1 , L sx2 , L x2 With C x2 They are all parallel resonant, and the impedance is infinite at the resonant frequency, which hinders the transmission of the third harmonic in the f1# receiving module and the fundamental wave in the f2# receiving module, thereby realizing energy decoupling transmission in the primary network. The parameters of the primary resonance element meet the following requirements:

[0238]

[0239] According to the basic circuit theory, it can be deduced that the fundamental frequency flows through the transmitting coil L p1 Current effective value I p_1 The secondary winding current at the third harmonic frequency is in the transmitting coil L p2 The effective value of the induced voltage V ind_3 They are:

[0240]

[0241] It can be seen that the resonant frequency is ω _1 When the current flows through the transmitting coil L p1 The current is constant, independent of the load, and the resonant frequency is ω _3 When the secondary winding current is in the transmitting coil L p2 The induced voltage generated is constant and has nothing to do with the load.

[0242] According to the circuit fundamental wave theory, the DC output current I can be deduced o for:

[0243]

[0244] The output current is independent of the load, and constant current output can be achieved under variable load conditions. In addition, the output current of the f1# receiving module 5_1 changes in direct proportion to the value of the mutual inductance M1, and the output current of the f2# receiving module 5_2 changes in inverse proportion to the value of the mutual inductance M2. When the output current of M1 and M2 have the same change trend under the offset condition, that is, when they increase or decrease at the same time, the output current change of the f1# receiving module and the output current change of the f2# receiving module can offset each other to a certain extent, reducing the sensitivity of the output to the offset.

[0245] like Fig.40As shown, the primary side of the non-contact transformer adopts a double winding structure, and the compensation capacitors C1 and L p1 Connect in series, compensation capacitor C2 and L p2 Series connection; L x1 With C x1 The series connection forms the f1# frequency selection network, and the coil L p2 The roads are connected in parallel, L x2 With C x2 Connect in parallel to form f2# frequency selection network, and coil L p2 The secondary side receiving circuit is connected in series. Fig.23 Similarly, it should be noted that this embodiment requires that the secondary side f1# frequency selection network is connected in series with the fundamental wave receiving coil. The primary side frequency selection network component parameters satisfy:

[0246]

[0247] Impedance Z of f1# frequency selection network and f2# frequency selection network x_m With Z x_m They are:

[0248]

[0249] where ω _m is the mth harmonic frequency, ω _m =mω _1 . Due to L x1 With C x1 Series resonance, the impedance is zero at the resonant frequency, providing a low-resistance loop for the fundamental current; L x2 With C x2 They are all parallel resonances, and the impedance is infinite at the resonant frequency, which hinders the fundamental current from flowing through the harmonic transmitting coil, thereby realizing energy decoupling transmission in the primary network.

[0250] The parameters of the primary resonant element meet the following requirements:

[0251]

[0252] According to the basic circuit theory, it can be deduced that the third harmonic frequency flows through the transmitting coil L p2 Current effective value I p_3 The secondary winding current at the fundamental frequency is in the transmitting coil L p1 The effective value of the induced voltage V ind_1 They are:

[0253]

[0254] It can be seen that the resonant frequency is ω _1 When the secondary winding current is in the fundamental wave transmitting coil L p1The induced voltage generated is constant and has nothing to do with the load; the resonant frequency is ω _3 When the current flows through the harmonic transmitting coil L p2 The current is constant regardless of the load.

[0255] The parameters of the secondary resonant element satisfy equation (8), and the DC output voltage V o for:

[0256]

[0257] It can be seen that the output voltage of the f1# receiving module changes inversely with the value of the mutual inductance M1, and the output voltage of the f2# receiving module changes directly with the value of the mutual inductance M2. By reasonably designing the parameters of the transformer and resonant components, a constant voltage output can be achieved within a certain offset condition, thereby improving the anti-offset characteristics of the wireless power transmission system.

[0258] In particular, when there is no magnetic flux coupling between the fundamental wave transmitting coil and the harmonic transmitting coil, L p1 With L p2 Decoupled from each other, L x2 With C x2 The f2# frequency selection network composed of parallel connections can be omitted, such as Fig.41 As shown; but in L p1 With L p2 There is a magnetic flux coupling M p12 When L p2 The third harmonic energy in the p12 Coupling to L p1 , affecting the output characteristics. In order to achieve the decoupled transmission of dual-frequency energy, L x1 With C x1 The frequency selection network composed of parallel connection is essential, otherwise it will work at the fundamental frequency ω _1 When L p1 The current in the secondary winding will also generate an induced voltage on the secondary winding, causing the induced voltage on the secondary winding at the fundamental frequency to be related to the load and lose the constant voltage characteristic. In addition, in order to increase the freedom of parameter design, it is also possible to x1 , C x1 The capacitors are connected in parallel at both ends of the series path, such as Fig.42 As shown, the parameters of the primary resonant element satisfy:

[0259]

[0260] in At this time, the third harmonic frequency flows through the transmitting coil L p2 Current effective value I p_3 The secondary winding current at the fundamental frequency is in the transmitting coil L p1 The effective value of the induced voltage V ind_1They are:

[0261]

[0262] The corresponding DC output voltage V o for:

[0263]

[0264] like Fig.43 As shown, the primary side of the contactless transformer adopts a double winding structure, and the secondary side receiving circuit is Fig.14 Similarly, it should be noted that this embodiment requires that the secondary side f1# frequency selection network is connected in series with the fundamental wave receiving coil. Inductor L1, capacitor C1, inductor L2 and capacitor C2 form an LCLC dual-frequency resonant network, and the parameters meet the following requirements:

[0265]

[0266] Flowing through the fundamental wave transmitting winding L p1 The fundamental current effective value I p1_1 The third harmonic current effective value I p1_3 They are:

[0267]

[0268] You can see I p1_1 with I p1_3 All have nothing to do with the load. x1 With C x1 The series connection forms the f1# frequency selection network, and the transmitting coil L p2 Parallel connection, parameters meet:

[0269]

[0270] L x1 With C x1 Series resonance, the impedance is zero at the resonant frequency, providing a low resistance loop for the fundamental current, then L p2 Only harmonic current flows through it. It can be deduced that the secondary harmonic receiving winding current at the third harmonic frequency is L p2 The effective value of the induced voltage V ind_3 for:

[0271] V ind_3 =I p1_3 ω _3 L p2 (111)

[0272] Induction voltage V ind_3 Constant, independent of the load. The secondary resonant element parameters satisfy equation (8), and the DC output current I o for:

[0273]

[0274] It can be seen that the output current is independent of the load, and constant current output can be achieved under variable load conditions. In addition, the output current of the f1# receiving module changes in direct proportion to the value of the mutual inductance M1, and the output current of the f2# receiving module changes in direct proportion to the value of the mutual inductance M2. By reasonably designing the parameters of the transformer and resonant components, constant current output can be achieved within a certain offset condition, improving the anti-offset characteristics of the wireless power transmission system.

[0275] Test example:

[0276] In order to verify the feasibility of the present invention, Fig.28 The wireless power transmission topology shown in the figure is used as an example for simulation verification. The following table shows the specific values ​​of the resonant inductor and capacitor used in this test.

[0277] Table 3: Resonant element parameters

[0278] <![CDATA[L1(μH)]]> <![CDATA[C1(nF)]]> <![CDATA[C3(nF)]]> <![CDATA[C2(nF)]]> <![CDATA[L p (μH)]]> <![CDATA[L s (μH)]]> <![CDATA[C s1 (nF)]]> <![CDATA[C sx1 (nF)]]> <![CDATA[M s (μH)<!-- 21 --> ]]> 9.78 194.77 13.912 39.84 30 38 36.51 57.286 10.02 <![CDATA[C sx2 (nF)]]> <![CDATA[L sx2 (μH)]]> <![CDATA[C s3 (nF)]]> <![CDATA[L a (μH)]]> <![CDATA[L sr1 (μH)]]> <![CDATA[L sr2 (μH)]]> <![CDATA[L sx1 (μH)]]> <![CDATA[L b (μH)]]> <![CDATA[L2(μH)]]> 62.61 56 40.16 16.7 48 54.4 6.8 16.7 7

[0279] The input voltage V in is 30V, the operating frequency f0 is 85kHz, then the fundamental resonance frequency ω _1 =2πf0, harmonic resonance frequency ω _3 =6πf0. Take the load resistance R L 20Ω and 40Ω, different coupling coefficients The simulation results of the DC output voltage are as follows: Fig.44 As shown. It can be seen that the output voltage of the wireless power transmission topology based on multi-frequency energy parallel transmission proposed by the present invention fluctuates very little under variable load conditions, remains approximately constant, and has an output characteristic that is independent of the load; in addition, the wireless power transmission topology proposed by the present invention can also effectively improve the anti-deviation capability of the wireless power supply system. Fig.44 As shown in the figure, the coupling coefficient of the non-contact transformer changes from 0.1 to 0.5. Within the range of 5 times the coupling coefficient, the output voltage V o The fluctuation ξ is only 1.37 (definition ξ=V om1x / V omin ). For this wireless power transmission topology using a non-contact transformer with a single primary winding and a single secondary winding structure, the output fluctuation is only related to the coupling coefficient range, and is not limited by the relative offset direction of the primary and secondary sides of the non-contact transformer. Therefore, it can maintain a stable output in any offset direction, greatly improving the system's anti-offset performance.

[0280] Fig.45 and Fig.46The waveforms of the output voltages of the fundamental wave output module and the harmonic output module are given respectively when the coupling coefficients of this test example are k=0.15 and k=0.35. It can be seen that the fundamental wave and the harmonic are decoupled. The output voltage amplitude of the fundamental wave output module increases with the increase of the coupling coefficient, while the output voltage of the harmonic output module decreases with the increase of the coupling coefficient, which is consistent with the theoretical analysis.

[0281] The above is only a preferred embodiment of the present invention. It should be pointed out that for ordinary technicians in this technical field, several improvements and modifications can be made without departing from the principle of the present invention. These improvements and modifications should also be regarded as the scope of protection of the present invention.

Claims

1. A wireless power transmission topology with strong anti-deviation performance based on multi-frequency energy parallel transmission, characterized in that: It comprises a direct current power supply (1), a high frequency inverter (2), a primary multi-frequency common compensation network (3), a primary transmitting coil (4), a receiving module (5), a load (6), and a control module (7); The control module (7) outputs a driving square wave signal of the high-frequency inverter (2); the output of the high-frequency inverter (2) is a mixed superposition of dual-frequency or multi-frequency voltage or current, and the frequencies include f 1. f 2.…… f n , n≥2; the primary multi-frequency shared compensation network (3) adopts an LC high-order compensation topology and forms a primary resonance network with the primary transmitting coil (4), and the resonance frequency is f 1. f 2.…… f n , the multi-frequency energy simultaneously drives the primary transmitting coil (4) and is coupled to the receiving module (5) through the magnetic field; The receiving module (5) comprises receiving units of the same number as the frequency, each receiving unit comprises a frequency selection network, a compensation network and an energy coupling coil, and the n receiving units respectively output frequencies of f 1. f 2.…… f n The AC voltage or current is then rectified and filtered by the rectifying and filtering units and then combined and output to supply power to the load (6).

2. The wireless power transmission topology according to claim 1, characterized in that: The control module (7) controls the switching frequency or conduction time of the switch tube in the high-frequency inverter (2) to achieve an output frequency of the high-frequency inverter (2) of f 1. f 2……with f n A mixed superposition of voltages or currents.

3. The wireless power transmission topology according to claim 1, characterized in that: The control module (7) outputs a frequency of f The square wave signal drives the high frequency inverter (2), and the high frequency inverter (2) outputs a frequency f The square wave voltage superimposed with the fundamental wave and the nth harmonic sinusoidal voltage, the frequency f 1. f 2.…… f n It is the fundamental and / or third and / or fifth or any higher order odd harmonic frequency.

4. The wireless power transmission topology according to claim 1, characterized in that: The control module (7) outputs a time-sharing frequency of f 1. f 2.…… f n The lengths of the square wave signals of each frequency are t1, t2, ... t n , satisfying t1+t2+…+t n ≤T, realize time-division multiplexing, and adjust t1, t2, ... and t n The output is regulated by the time ratio.

5. The wireless power transmission topology according to any one of claims 1 to 4, characterized in that: By designing the parameters of the LC high-order compensation topology, the output voltage gain or current gain of the primary resonant network is made independent of the equivalent load of the primary resonant network.

6. A wireless power transmission topology according to any one of claims 1 to 4, characterized in that: The frequency selection network is an LC hybrid network.

7. The wireless power transmission topology according to any one of claims 1 to 4, characterized in that: The primary-side multi-frequency common compensation network (3) includes a switch for implementing parameter switching of the LC high-order compensation topology or uses switch capacitor or inductor tuning, and the control module (7) outputs a drive signal of the switch, switch capacitor or inductor of the primary-side multi-frequency common compensation network (3) so that the primary-side resonant network is at a frequency within a time t1. f 1 resonance, frequency within t2 time f 2 resonance, ..., t n Frequency in time f n Resonance, satisfying t1+t2+…+t n ≤T, multi-frequency energy time-sharing transmission is realized, and its output voltage gain or current gain is independent of the equivalent load of the primary side resonant network.

8. The wireless power transmission topology according to any one of claims 1 to 4, characterized in that: The primary transmitting coil (4) is a single winding, double winding or multi-winding structure, two or more windings share a primary multi-frequency common compensation network (3), and there is no magnetic flux coupling between any two windings or a series LC frequency selection network to achieve decoupling control.

9. The wireless power transmission topology according to claim 1, characterized in that: The output of the high-frequency inverter (2) is a mixed superposition of dual-frequency voltages or currents, with a frequency of f 1 and f 2; and f The output voltage gain of the primary resonant network at a frequency of 1 is independent of the equivalent load of the primary resonant network. f The output current gain of the primary resonance network at the 2nd frequency is independent of the equivalent load of the primary resonance network, and the dual-frequency energy simultaneously drives the primary transmitting coil (4) and is coupled to the receiving end through the magnetic field; The receiving module (5) comprises a receiving unit 1 and a receiving unit 2. The energy coupling coil of the receiving unit 1 and the primary transmitting coil (4) form a contactless transformer. The coupling frequency of the energy coupling coil of the receiving unit 1 is f 1. f 2 AC voltage or current; the energy coupling coil of the receiving unit 2 is coupled to the receiving unit 1 to obtain a frequency of f 2 AC voltage or current; the receiving unit 1 and the receiving unit 2 respectively output a frequency of f 1. f The AC voltage or current of 2 is then rectified and filtered by the rectifying and filtering units and then combined and output to supply power to the load (6).

10. A wireless power transmission topology with strong anti-deviation performance based on multi-frequency energy parallel transmission, characterized in that: It comprises a direct current power supply (1), a high frequency inverter (2), a primary multi-frequency common compensation network (3), a primary transmitting coil (4), a receiving module (5), a load (6), and a control module (7); The control module (7) outputs a driving square wave signal of the high-frequency inverter (2); the output of the high-frequency inverter (2) is a mixed superposition of dual-frequency or multi-frequency voltage or current, and the frequencies include f 1. f 2.…… f n , n≥2; the primary multi-frequency shared compensation network (3) adopts an LC high-order compensation topology and forms a primary resonance network with the primary transmitting coil (4), and the resonance frequency is f 1. f 2.…… f n , and the output voltage gain or current gain of the primary resonant network is independent of the equivalent load of the primary resonant network, and the multi-frequency energy simultaneously drives the primary transmitting coil (4) and is coupled to the receiving module (5) through a magnetic field; The receiving module (5) comprises a secondary side receiving coil, a secondary side multi-frequency common compensation network, and a multi-frequency common rectification and filtering circuit; the secondary side multi-frequency common compensation network is a high-order LC network, and forms a secondary side resonant network with the secondary side receiving coil; the secondary side multi-frequency common compensation network comprises a switch for realizing parameter switching; the control module (7) outputs a switch driving signal of the secondary side multi-frequency common compensation network, so that the secondary side resonant network is switched at t i The resonant frequency in time is f i , and the output voltage gain or current gain of the secondary resonant network has nothing to do with the equivalent load of the secondary resonant network; the secondary multi-frequency shared compensation network is at t i The output frequency during the time is f i The AC signal is rectified and filtered before supplying power to the load (6).

11. A wireless power transmission topology with strong anti-deviation performance based on multi-frequency energy parallel transmission, characterized in that: It comprises a direct current power supply (1), a high frequency inverter (2), a primary multi-frequency common compensation network (3), a primary transmitting coil (4), a receiving module (5), a load (6), and a control module (7); The control module (7) outputs a time-sharing frequency of f 1. f 2.…… f n A square wave signal of a frequency is used as a driving signal of the high frequency inverter (2), and the lengths of the square wave signals of each frequency are t1, t2, ... t n , and satisfy t1+t2+……+t n ≤T, and by adjusting t1, t2, ... and t n Output regulation is achieved by the time ratio; The primary multi-frequency shared compensation network (3) adopts an LC high-order compensation topology and forms a primary resonance network with the primary transmitting coil (4). The resonance frequency is f 1. f 2.…… f n , and the output voltage gain or current gain of the primary resonance network is independent of the equivalent load of the primary resonance network, so as to achieve multi-frequency energy time-sharing driving of the primary transmitting coil (4) and couple to the receiving module (5) through a magnetic field; The receiving module (5) comprises a secondary receiving coil, a secondary multi-frequency common compensation network and a rectifier filter unit connected in sequence. The secondary multi-frequency common compensation network is an LC high-order compensation topology and forms a secondary resonant network with the secondary receiving coil. i The output frequency during the time is f i The AC signal is rectified and filtered to supply power to the load (6).

12. A wireless power transmission topology with strong anti-deviation performance based on multi-frequency energy parallel transmission, characterized in that: It comprises a direct current power supply (1), a high-frequency inverter (2), a primary transmitting module, a receiving module (5), a load (6) and a control module (7); The control module (7) outputs a driving square wave signal of the high-frequency inverter (2), and the output of the high-frequency inverter (2) is a mixed superposition of dual-frequency or multi-frequency voltage or current, and the frequencies include f 1. f 2.…… f n , n≥2; the primary transmitting module includes the same number of transmitting units as the number of frequencies, each transmitting unit includes a frequency selection network, a compensation network and a transmitting coil, and the n transmitting units respectively output frequencies of f 1. f 2.…… f n The output voltage gain or current gain is independent of the equivalent load of the primary transmitting module; The receiving module (5) comprises receiving units of the same number as the frequency, each receiving unit comprising a frequency selection network, a compensation network and a receiving coil, and outputs frequencies of f 1. f 2.…… f n The AC voltage or current is then rectified and filtered by the rectifying and filtering units and then combined and output to supply power to the load (6).

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