A marine microgrid inverter, modulation strategy and control method

Abstract

The present invention discloses a marine microgrid inverter, modulation strategy and control method. The inverter includes a DC support capacitor, an A-phase bridge inverter unit, a B-phase bridge inverter unit, a C-phase bridge inverter unit, a three-winding transformer and a three-phase AC filter. The external input DC voltage is applied to both ends of the DC support capacitor to form a DC bus. The three bridge inverter units have the same circuit topology, and the input DC voltage is converted into an AC PWM voltage, which is then superimposed in series through the transformer windings. The DC voltage utilization rate and output voltage quality of the inverter are improved by matching the transformer windings, and the power density is improved by the magnetic integration technology of the transformer; the modulation strategy of the present invention suppresses the common mode voltage and the primary circulating current of the transformer while realizing the equivalent frequency doubling of the switching frequency, reduces the vibration noise of the inverter, and improves the power density; the control method of the present invention can improve the output voltage quality of the inverter with nonlinear load, and realizes the effective suppression of the primary current harmonics of the transformer, and reduces the vibration energy level of the inverter.

Description

Technical Field

[0001] The invention belongs to the technical field of inverter power supplies, and in particular relates to a marine microgrid inverter, a modulation strategy and a control method. Background Art

[0002] At present, the known microgrid inverter generally adopts the three-phase three-bridge arm inverter topology, which has the advantages of simple circuit topology, small number of components, low cost and mature control strategy. However, special ship microgrid inverters have higher requirements than general microgrid inverters in terms of power density, DC voltage utilization, output voltage quality and vibration and noise indicators. The known circulating current suppression measures mainly include methods such as serially connecting circulating current limiting inductors and changing the carrier phase of the inverter unit. Among them, serially connecting circulating current filter inductors inevitably increases the size and cost of the inverter; although changing the carrier phase of the inverter unit does not increase the hardware cost, the carrier phase determines the multiple harmonic superposition characteristics, which may bring new problems such as the decline of output voltage waveform quality and electromagnetic compatibility indicators while suppressing the circulating current.

[0003] At the same time, three-phase inverters generally adopt a voltage-current dual closed-loop control strategy in a synchronous rotating coordinate system, and their regulators generally use PI regulators, which can achieve zero-static tracking of the fundamental voltage command. However, the PI regulator has limited ability to suppress harmonics, so when carrying nonlinear loads, the quality of the inverter output voltage is significantly reduced. In order to improve the harmonic suppression capability of the inverter, the closed-loop control bandwidth can be increased, but the system stability margin may be reduced, resulting in an increase in voltage overshoot or even instability during the dynamic load process. In addition, the transformer-based multiple superposition topology is adopted, and the transformer primary current harmonics are relatively rich, which will inevitably increase the vibration level of the transformer, so effective primary current harmonic suppression measures need to be taken. Summary of the invention

[0004] In view of the above technical defects, the present invention proposes a marine microgrid inverter, modulation strategy and control method which can improve the power density, DC voltage utilization and output voltage quality of the inverter while reducing the vibration and noise energy level.

[0005] To achieve the above-mentioned purpose, the present invention proposes a marine microgrid inverter, comprising a DC support capacitor 1, an A-phase bridge inverter unit 2, a B-phase bridge inverter unit 3, a C-phase bridge inverter unit 4, a three-winding transformer 5 and a three-phase AC filter 6, wherein an externally input DC voltage is applied to both ends of the DC support capacitor 1 to form a DC bus; the three bridge inverter units have the same circuit topology, each bridge inverter unit is composed of four inverter bridge arms connected in parallel to the DC bus, each inverter bridge arm is composed of two series switching devices, and two groups of inverter bridge arms constitute an H-bridge inverter unit.

[0006] Furthermore, the A-phase bridge inverter unit 2 includes a first switch device 20 and a second switch device 21 connected in series to form a first inverter bridge arm, and a third switch device 22 and a fourth switch device 23 connected in series to form a second inverter bridge arm. The two inverter bridge arms form an H-bridge inverter unit connected in parallel with the DC bus; another H-bridge inverter unit formed by a fifth switch device 24 and a sixth switch device 25 connected in series, and a seventh switch device 26 and an eighth switch device 27 connected in series is connected in parallel with the DC bus; one of the two H-bridge inverter units is connected to the first winding 50 on the primary side of the A-phase, and the other is connected to the second winding 51 on the primary side of the A-phase; the PWM voltages output by the two H-bridge inverter units are superimposed in series through the transformer windings, and a dualized PWM voltage is output at the A-phase secondary winding 52; the dualized PWM voltage is input to the three-phase AC filter 6 of the inverter, and the industrial frequency output voltage is obtained after filtering.

[0007] A modulation strategy for the marine microgrid inverter as described above is also provided, wherein the modulation strategy is as follows: the phase difference between the first inverter bridge arm triangle carrier and the second inverter bridge arm triangle carrier, and the third inverter bridge arm triangle carrier and the fourth inverter bridge arm triangle carrier of each phase bridge inverter unit is 180°, and the phase difference between the first inverter bridge arm triangle carrier and the third inverter bridge arm triangle carrier, and the second inverter bridge arm triangle carrier and the fourth inverter bridge arm triangle carrier of each phase bridge inverter unit is 90°; taking the phase of the first inverter bridge arm triangle carrier of the A-phase bridge inverter unit as a reference, the phase difference between the inverter bridge arm triangle carriers of the B-phase bridge inverter unit and the inverter bridge arm triangle carriers corresponding to the A-phase bridge inverter unit is α, and the phase difference between the inverter bridge arm triangle carriers of the C-phase bridge inverter unit and the inverter bridge arm triangle carriers corresponding to the A-phase bridge inverter unit is β.

[0008] Furthermore, α=180°, β=180°.

[0009] A control method for the marine microgrid inverter as described above is also provided, wherein the sampled three-phase voltage u oa (t),u ob (t) and u oc (t) The positive sequence fundamental voltage of the d-axis in the positive synchronous rotating coordinate system is obtained by decomposing the positive sequence and negative sequence fundamental voltage of the DSC and performing dq transformation. Q-axis positive sequence fundamental voltage per unit value Unit value of negative sequence fundamental voltage of d-axis in counter-rotating synchronous coordinate system And the q-axis positive sequence fundamental voltage per unit value and As the feedback value of the positive sequence voltage controller 82, it participates in the inverter output positive sequence voltage control. and As the feedback value of the negative sequence voltage controller 83, it participates in the inverter output negative sequence voltage control; the d-axis per unit voltage reference value of the positive sequence voltage controller 82 is 1, minus the feedback value Then it is input to the d-axis positive-sequence voltage PI regulator 821 of the positive-sequence voltage controller 82. The q-axis per-unit voltage reference value of the positive-sequence voltage controller 82 is 0, minus the feedback value Then it is input to the q-axis positive sequence voltage PI regulator 822 of the positive sequence voltage controller 82;

[0010] The d-axis per unit voltage reference value of the negative sequence voltage controller 83 is 0, minus the feedback value The output of the d-axis negative sequence voltage PI regulator 831 is input to the negative sequence voltage controller 83. Multiply by the cross decoupling term ωC f After subtraction, it is output to the d-axis input port of the negative-sequence dq to positive-sequence dq conversion module 86; the q-axis per-unit voltage reference value of the negative-sequence voltage controller 83 is 0, minus the feedback value Then, it is input to the q-axis negative-sequence voltage PI regulator 832 of the negative-sequence voltage controller 83. The output of the q-axis negative-sequence voltage PI regulator 832 is Multiply by the cross decoupling term ωC f After subtraction, the output is sent to the q-axis input port of the negative-sequence dq to positive-sequence dq transformation module 85;

[0011] The negative sequence dq to positive sequence dq conversion module 85 converts -ω o The dq component of the negative sequence dq coordinate system of velocity rotation passes through -2ω o Transform to ω o In the positive sequence dq coordinate system of speed rotation, the transformed dq axis component is superimposed with the dq axis component output of the positive sequence voltage controller 82 to obtain the d axis current reference value of the current controller 84 and q-axis current reference value

[0012] Add the d-axis output current Feedforward minus d-axis inductor current feedback The input is to the d-axis current PI+R regulator 843 which is formed by adding a resonant control link on the basis of the PI regulator; the q-axis current Cross-decoupling term ωL m And the output voltage The feedforward term is then superimposed with the output of the third harmonic proportional resonant PR controller 841 of the transformer d-axis primary zero-sequence current to obtain the modulation wave d-axis component u d ; Add the q-axis output current Feedforward minus q-axis inductor current feedback Input to q-axis current PI+R regulator 844; d-axis current Cross-decoupling term ωL m And the output voltage The feedforward term is then superimposed with the output of the third harmonic proportional resonant PR controller 842 of the transformer q-axis primary zero-sequence current to obtain the modulation wave q-axis component u q ;

[0013] For the modulated wave d-axis component u d With the q-axis component u q Perform the dq inverse transformation to obtain the ABC three-phase modulation wave voltage u A 、u B and u C , and then the driving pulse of the corresponding switching device is obtained through SPWM modulation.

[0014] Compared with the prior art, the beneficial effects of the present invention are as follows: the multiple combined inverter topology improves the DC voltage utilization rate and output voltage quality of the inverter through the coordination of transformer windings, and improves the power density through the magnetic integration technology of the transformer; the modulation strategy of the present invention suppresses the common mode voltage and the primary circulating current of the transformer while realizing the equivalent frequency doubling of the switching frequency, reduces the vibration noise of the inverter, and improves the power density; the control method of the present invention can improve the output voltage quality of the inverter with nonlinear load, and realizes the effective suppression of the primary current harmonics of the transformer, thereby reducing the vibration energy level of the inverter. BRIEF DESCRIPTION OF THE DRAWINGS

[0015] Figure 1 This is a topological structure diagram of the marine microgrid inverter of the present invention;

[0016] Figure 2 It is the power frequency modulation wave of the H-bridge inverter unit and two triangle carrier waveforms;

[0017] Figure 3 Output PWM voltage waveform diagram for H-bridge inverter unit;

[0018] Figure 4 It is a bridge inverter unit power frequency modulation wave and four triangle carrier waveforms;

[0019] Figure 5 The PWM voltage waveform diagram output by a bridge inverter unit;

[0020] Figure 6 The influence of the phase shift angle of the inter-phase carrier on the output common mode voltage amplitude of the inverter;

[0021] Figure 7 The influence of the phase shift angle of the inter-phase carrier on the effective value of the inverter output common mode voltage;

[0022] Figure 8 is the influence of the phase shift angle of the inter-phase carrier on the effective value of the circulating current of phase A;

[0023] Fig. 9 is the influence of the phase shift angle of the inter-phase carrier on the effective value of the circulating current of phase B;

[0024] Fig.10 is the influence of the phase shift angle of the inter-phase carrier on the effective value of the C phase circulating current;

[0025] Fig.11 This is a block diagram of the inverter control system of the present invention.

[0026] In the figure, 1. DC support capacitor, 2. A-phase bridge inverter unit, 3. B-phase bridge inverter unit, 4. C-phase bridge inverter unit, 5. multi-winding transformer, 6. three-phase AC filter, 20. A-phase bridge inverter unit first switch device, 21. A-phase bridge inverter unit second switch device, 22. A-phase bridge inverter unit third switch device, 23. A-phase bridge inverter unit fourth switch device, 24. A-phase bridge inverter unit fifth switch device, 25 .A-phase bridge inverter unit sixth switch device, 26.A-phase bridge inverter unit seventh switch device, 27.A-phase bridge inverter unit eighth switch device, 50.A-phase primary side first winding, 51.A-phase primary side second winding, 52.A-phase secondary side winding, 53.B-phase primary side first winding, 54.B-phase primary side second winding, 55.B-phase secondary side winding, 56.C-phase primary side first winding, 57.C-phase primary side second winding, 58.C Phase secondary winding, 70. Power frequency modulation wave, 71. First inverter bridge arm triangle carrier, 72. Third inverter bridge arm triangle carrier, 73. Second inverter bridge arm triangle carrier, 74. Fourth inverter bridge arm triangle carrier, 75. Transformer primary winding PWM voltage, 76. Transformer secondary winding PWM voltage, 82. Positive sequence voltage controller, 83. Negative sequence voltage controller, 84. Current controller, 85. Negative sequence dq to positive sequence dq transformation, 821. d-axis positive sequence voltage PI regulator, 822. q-axis positive sequence voltage PI regulator 822, 831. d-axis negative sequence voltage PI regulator, 832. q-axis negative sequence voltage PI regulator, 841. Third harmonic proportional resonant PR controller of transformer d-axis primary zero-sequence current, 842. Third harmonic proportional resonant PR controller of transformer q-axis primary zero-sequence current, 843. d-axis current PI+R regulator, 844. q-axis current PI+R regulator. DETAILED DESCRIPTION

[0027] In order to make the purpose, technical solutions and advantages of the present invention more clearly understood, the present invention is further described in detail below with reference to the accompanying drawings.

[0028] The present invention analyzes the influence of the carrier phase shift angle between the three phases of the inverter unit on the output common-mode voltage and the circulating current, and obtains a carrier phase shift angle combination that takes into account both common-mode voltage and circulating current suppression. It is relatively easy to implement and can reduce the cost of circulating current filter inductance and EMI filter. It is further explained below in conjunction with the accompanying drawings and specific implementation methods.

[0029] Figure 1 The topological structure diagram of the marine microgrid inverter of the present invention includes a DC support capacitor 1, an A-phase bridge inverter unit 2, a B-phase bridge inverter unit 3, a C-phase bridge inverter unit 4, a three-winding transformer 5 and a three-phase AC filter 6, wherein an externally input DC voltage is applied to both ends of the DC support capacitor 1 to form a DC bus. The three bridge inverter units have the same circuit topology, and each bridge inverter unit is composed of four inverter bridge arms connected in parallel to the DC bus, each inverter bridge arm is composed of two series-connected switch devices, and two groups of inverter bridge arms constitute an H-bridge inverter unit. Take the A-phase bridge inverter unit 2 as an example: the first switch device 20 and the second switch device 21 are connected in series to form a first inverter bridge arm, the third switch device 22 and the fourth switch device 23 are connected in series to form a second inverter bridge arm, and these two inverter bridge arms form an H-bridge inverter unit connected in parallel with the DC bus; similarly, another H-bridge inverter unit formed by the fifth switch device 24 and the sixth switch device 25 connected in series, and the seventh switch device 26 and the eighth switch device 27 connected in series is connected in parallel with the DC bus; one of these two H-bridge inverter units is connected to the first winding 50 on the primary side of the A-phase, and the other is connected to the second winding 51 on the primary side of the A-phase; the PWM voltages output by the above two H-bridge inverter units are superimposed in series through the transformer windings, and a dualized PWM voltage is output at the A-phase secondary winding 52; the dualized PWM voltage is input to the three-phase AC filter 6 of the inverter, and the industrial frequency output voltage is obtained after filtering.

[0030] To convert the input DC voltage into AC PWM voltage, Figure 1 The inverter topology shown can adopt a carrier phase shift modulation strategy based on unipolar frequency multiplication modulation: the phase difference between the first inverter bridge arm triangle carrier and the second inverter bridge arm triangle carrier, and the third inverter bridge arm triangle carrier and the fourth inverter bridge arm triangle carrier of each phase bridge inverter unit is 180°, and the phase difference between the first inverter bridge arm triangle carrier and the third inverter bridge arm triangle carrier, and the second inverter bridge arm triangle carrier and the fourth inverter bridge arm triangle carrier of each phase bridge inverter unit is 90°; taking the phase of the first inverter bridge arm triangle carrier of the A-phase bridge inverter unit as a reference, the phase difference between each inverter bridge arm triangle carrier of the B-phase bridge inverter unit and each inverter bridge arm triangle carrier corresponding to the A-phase bridge inverter unit is α, and the phase difference between each inverter bridge arm triangle carrier of the C-phase bridge inverter unit and each inverter bridge arm triangle carrier corresponding to the A-phase bridge inverter unit is β, and α=180°, β=180°.

[0031] It is precisely because of the existence of the triangular carrier phase shift angle that the high-frequency harmonics near the triangular carrier and its multiple frequencies are coupled through the windings of the common transformer core, which will generate high-frequency circulating currents. In order to suppress the high-frequency circulating currents between the inverter units, the present invention uses the control freedom of the inverter three-phase carrier phase shift angle to analyze the influence of the phase-to-phase carrier phase shift angle on the circulating current. Finally, in order to take into account the inverter AC output common mode voltage suppression, the present invention provides the optimized phase angles of the A-phase, B-phase and C-phase carriers. The use of this carrier phase angle combination can not only suppress the circulating currents between the inverter units, but also reduce the output common mode voltage and improve the electromagnetic compatibility performance.

[0032] In this embodiment, the triangular carrier phase of the first inverter bridge arm of the A-phase bridge inverter unit is 0°, the triangular carrier phase of the second inverter bridge arm is -180°, the triangular carrier phase of the third inverter bridge arm is -90°, and the triangular carrier phase of the fourth inverter bridge arm is -270°. The triangular carrier phase of the first inverter bridge arm of the B-phase bridge inverter unit is α, the second bridge arm carrier phase is α-180°, the third bridge arm carrier phase is α-90°, and the fourth bridge arm carrier phase is α-270°. The triangular carrier phase of the first inverter bridge arm of the C-phase bridge inverter unit is β, the second bridge arm carrier phase is β-180°, the third bridge arm carrier phase is β-90°, and the fourth bridge arm carrier phase is β-270°.

[0033] The angles of α and β can be adjusted according to specific requirements, and do not affect the inverter output power frequency voltage and the equivalent quadruple frequency and five-level effects. However, different carrier phases will affect the circulating current between the three-phase inverter units and the AC output common mode voltage. The following is a specific implementation method for selecting the carrier phase between the three-phase inverter units:

[0034] The voltage of the first winding 50 on the primary side of phase A is V 1 , the current is I 1 , the voltage of the second winding 51 on the primary side of phase A is V 2 , the current is I 2 , the voltage of the secondary winding 52 of phase A is V 3 , the current is I 3 ; The voltage of the first winding 53 on the primary side of phase B is V 4 , the current is I 4 , the voltage of the second winding 54 on the primary side of phase B is V 5 , the current is I 5 , the voltage of the secondary winding 55 of phase B is V 6 , the current is I 6 ; The voltage of the first winding 56 on the primary side of phase C is V 7 , the current is I 7 , the voltage of the second winding 57 on the primary side of phase C is V 8 , the current is I 8 , the voltage of the secondary winding 58 of phase C is V 9, the current is I 9 According to the transformer port voltage and current equation, we can get:

[0035]

[0036] Where V = [V 1 V 2 V 3 V 4 V 5 V 6 V 7 V 8 V 9 ] T , I=[I 1 I 2 I 3 I 4 I 5 I 6 I 7 I 8 I 9 ] T ,

[0037]

[0038]

[0039] The transformer port voltage and current coupling relationship is complex. Among them, the resistance value R in the resistance matrix R i (i=1,2,3,4,5,6,7,8,9) is the resistance of the i-th winding of the transformer; the inductance matrix L is the transformer model established based on the coupled inductance. There are 81 inductance parameters, among which the inductance L on the diagonal of the inductance matrix ii (i=1,2,3,4,5,6,7,8,9) is the self-inductance of the 9 windings of the transformer, and the inductance L ij =L ji (i=1,2,3,4,5,6,7,8,9) is the mutual inductance between winding i and winding j, with a total inductance value of 72. Due to the large number of circuit parameters, it is difficult to intuitively use a calculation formula to express the relationship between the inverter AC output common mode voltage, the inter-unit circulating current and the power unit phase-to-phase carrier shift angle.

[0040] Define the inverter AC output common mode voltage V cm :

[0041] V cm =(V 3 +V 6 +V 9 ) / 3 (2)

[0042] Taking the A phase carrier as the reference, Figure 4The variation law of the inverter output common mode voltage amplitude with the BA phase-to-phase carrier phase-shift angle and the CA phase-to-phase carrier phase-shift angle is shown; Figure 5 The figure shows the variation of the inverter output common-mode voltage RMS value with the BA phase-to-phase carrier phase shift angle and the CA phase-to-phase carrier phase shift angle. According to the volt-second balance principle, changing the phase of the carrier does not affect the change of the output power frequency voltage amplitude. When the three-phase carrier phase is in the two combinations of (0, -120°, 120°) and (0, 120°, -120°), both the common-mode voltage amplitude and the RMS value are the smallest.

[0043] Define the circulating current between the A-phase inverter units as I cir_A , the circulating current between the B-phase inverter units is I cir_B , the circulating current between the C-phase inverter units is I cir_C :

[0044]

[0045] Figure 6 , Figure 7 and Figure 8 The effective values ​​of the circulating current of the inverter units of phases A, B and C are shown as the changes of the carrier phase shift angles between phases BA and CA. The phase of the carrier of phase A is also used as the reference. The simulation of the carrier phase angles of phases B and C can obtain the effective values ​​of the three-phase circulating current under different carrier phase combinations. When the phase shift angles of phases A, B and C are (0°, 0°, 0°), (0°, 0°, 180°), (0°, 180°, 0°) and (0°, 180°, 180°), the circulating current of each phase is the smallest.

[0046] Therefore, the analysis results of the AC output common mode voltage and the circulating current between the inverter units show that the phase shift angles that minimize the common mode voltage and the circulating current are not the same. If a compromise is made between the two, the priority is to suppress the circulating current and select (0, 180°, 180°) as the optimized carrier phase angle, that is, α=180°, β=180°.

[0047]

[0048] Since the three-phase bridge inverter unit of the inverter has three power frequency modulation waves, namely, the power frequency modulation wave of the A-phase bridge inverter unit, the power frequency modulation wave of the B-phase bridge inverter unit, and the power frequency modulation wave of the C-phase bridge inverter unit, and the three-phase power frequency modulation waves are 120° apart from each other; therefore, the power frequency modulation wave of the A-phase bridge inverter unit is used as the power frequency modulation wave shared by the four inverter bridge arms of the A-phase bridge inverter unit, the power frequency modulation wave of the B-phase bridge inverter unit is used as the power frequency modulation wave shared by the four inverter bridge arms of the B-phase bridge inverter unit, and the power frequency modulation wave of the C-phase bridge inverter unit is used as the power frequency modulation wave shared by the four inverter bridge arms of the C-phase bridge inverter unit.

[0049] Combination Figure 2 , 3 The modulation strategy of an H-bridge inverter unit shown in the figure is that when the power frequency modulation wave 70 is greater than the first inverter bridge arm triangle carrier 71, the upper switch tube of the first inverter bridge arm is driven to be turned on and the lower switch tube is turned off; the first inverter bridge arm triangle carrier 71 is inverted to obtain the second inverter bridge arm triangle carrier 73, and the same power frequency modulation wave 70 is used for comparison. When the power frequency modulation wave 70 is greater than the second inverter bridge arm triangle carrier 73, the upper switch tube of the second inverter bridge arm is driven to be turned on and the lower switch tube is turned off; since the phase difference between the first inverter bridge arm triangle carrier 71 and the second inverter bridge arm triangle carrier 73 is 180°, after adopting this modulation strategy, the polarity of the transformer primary winding PWM voltage 75 is the same as the power frequency modulation wave polarity, and its equivalent output harmonic frequency is equal to twice the carrier frequency.

[0050] Combination Figure 4 , 5 The carrier signals corresponding to the four inverter bridge arms in a bridge inverter unit are shown, and the third inverter bridge arm triangle carrier 72 and the fourth inverter bridge arm triangle carrier 74 lag the first inverter bridge arm triangle carrier 71 and the second inverter bridge arm triangle carrier 73 by 90 degrees respectively. The PWM output voltages of the two H-bridge inverter units are superimposed through the transformer windings, and the transformer secondary winding voltage 76 presents a five-level PWM characteristic of four times the carrier frequency. Therefore, the waveform of the transformer secondary winding PWM voltage 76 after transformer superposition presents four times the carrier frequency and five-level characteristics, which improves the power density of the inverter and reduces the vibration energy level within the inverter test range.

[0051] The present invention analyzes the inverter control system architecture to illustrate its ability to improve the output voltage waveform quality and suppress harmonics, which will be further explained below in conjunction with the accompanying drawings and specific implementation methods.

[0052] Fig. 9 The control algorithm block diagram of the present invention is shown, and the three-phase voltage u oa (t),u ob (t) and u oc (t) The positive sequence fundamental voltage of the d-axis in the positive synchronous rotating coordinate system is obtained by decomposing the positive sequence and negative sequence fundamental voltage of the DSC and performing dq transformation. Q-axis positive sequence fundamental voltage per unit value Unit value of negative sequence fundamental voltage of d-axis in counter-rotating synchronous coordinate system And the q-axis positive sequence fundamental voltage per unit value and As the feedback value of the positive sequence voltage controller 82, it participates in the inverter output positive sequence voltage control. and As the feedback value of the negative sequence voltage controller 83, it participates in the inverter output negative sequence voltage control.

[0053] The d-axis per unit voltage reference value of the positive sequence voltage controller 82 is 1, minus the feedback value Then it is input to the d-axis positive-sequence voltage PI regulator 821 of the positive-sequence voltage controller 82. The q-axis per-unit voltage reference value of the positive-sequence voltage controller 82 is 0, minus the feedback value Then it is input to the q-axis positive sequence voltage PI regulator 822 of the positive sequence voltage controller 82;

[0054] The d-axis per unit voltage reference value of the negative sequence voltage controller 83 is 0, minus the feedback value The output of the d-axis negative sequence voltage PI regulator 831 is input to the negative sequence voltage controller 83. Multiply by the cross decoupling term ωC f After subtraction, it is output to the d-axis input port of the negative-sequence dq to positive-sequence dq conversion module 86; the q-axis per-unit voltage reference value of the negative-sequence voltage controller 83 is 0, minus the feedback value Then, it is input to the q-axis negative-sequence voltage PI regulator 832 of the negative-sequence voltage controller 83. The output of the q-axis negative-sequence voltage PI regulator 832 is Multiply by the cross decoupling term ωC f After subtraction, the output is sent to the q-axis input port of the negative-sequence dq to positive-sequence dq transformation module 85;

[0055] The negative sequence dq to positive sequence dq conversion module 85 converts -ω o The dq component of the negative sequence dq coordinate system of velocity rotation passes through -2ω o Transform to ω o In the positive sequence dq coordinate system of speed rotation, the transformed dq axis component is superimposed with the dq axis component output of the positive sequence voltage controller 82 to obtain the d axis current reference value of the current controller 84 and q-axis current reference value

[0056] Add the d-axis output current Feedforward minus d-axis inductor current feedback The input is to the d-axis current PI+R regulator 843 which is composed of a resonant control link added to the PI regulator, which can effectively suppress the 5th and 7th harmonic distortion in the output voltage waveform; the q-axis current Cross-decoupling term ωL m And the output voltage The feedforward term is then superimposed with the output of the third harmonic proportional resonant PR controller 841 of the transformer d-axis primary zero-sequence current to obtain the modulation wave d-axis component u d . Add the q-axis output current Feedforward minus q-axis inductor current feedback Input to q-axis current PI+R regulator 844; d-axis current Cross-decoupling term ωL m And the output voltage The feedforward term is then superimposed with the output of the third harmonic proportional resonant PR controller 842 of the transformer q-axis primary zero-sequence current to obtain the modulation wave q-axis component u q .

[0057] For the modulated wave d-axis component u d With the q-axis component u q Perform the dq inverse transformation to obtain the ABC three-phase modulation wave voltage u A 、u B and u C . Then, the drive pulse of the corresponding switching device is obtained through SPWM modulation. The use of this control algorithm can suppress the negative sequence component in the fundamental voltage and improve the inverter's ability to carry unbalanced loads; it can also suppress the harmonics caused by nonlinear loads and the harmonics of the primary current of the transformer, improve the quality of the output voltage waveform, and reduce the vibration energy level of the inverter. Based on the proposed topology, the present invention discloses a strategy for suppressing harmonics in the output voltage of the inverter. The control strategy is based on a voltage-current dual closed-loop control architecture in a synchronous rotating coordinate system, and uses a PI regulator to achieve zero-static error tracking of the fundamental voltage command. Furthermore, in order to suppress the harmonics caused by nonlinear loads, the present invention adds a resonant link on the basis of the PI regulator, thereby improving the control system's ability to suppress harmonics, improving the quality of the output voltage waveform, and reducing the vibration energy level of the transformer.

[0058] The special marine microgrid inverter disclosed in the present invention adopts a multiple combined inverter topology. The combined three-phase inverter is composed of three single-phase bridge inverter units. Each phase inverter unit is relatively independent. Its design and control method are similar to those of the single-phase inverter. Compared with the three-phase half-bridge inverter, the combined three-phase inverter has the advantages of high DC voltage utilization, flexible control, and strong unbalanced load capacity.

[0059] The present invention staggers the output phases of two or more inverter units and superimposes them through a transformer or inductor to increase the number of output voltage levels. Although the switching frequency of each unit remains unchanged, the overall output equivalent switching frequency of the inverter is doubled. Therefore, the present invention can reduce output voltage harmonics, reduce the volume and weight of the output filter, reduce the heat dissipation pressure of power semiconductor devices, and does not need to increase the inverter power level through a complex series-parallel design of power devices; by using transformer coupling at the output end, multiplexing and electrical isolation can be achieved on the secondary side.

[0060] The multiple combined inverter topology of the present invention improves the DC voltage utilization rate and output voltage quality of the inverter by matching the transformer windings, and improves the power density by the magnetic integration technology of the transformer; the modulation strategy of the present invention suppresses the common mode voltage and the primary circulating current of the transformer while achieving equivalent frequency doubling of the switching frequency, reduces the vibration noise of the inverter, and improves the power density; the control method of the present invention can improve the output voltage quality of the inverter with nonlinear load, and effectively suppresses the harmonics of the primary current of the transformer, thereby reducing the vibration energy level of the inverter.

Claims

1. A marine microgrid inverter, characterized in that: It comprises a DC support capacitor (1), an A-phase bridge inverter unit (2), a B-phase bridge inverter unit (3), a C-phase bridge inverter unit (4), a three-winding transformer (5) and a three-phase AC filter (6), wherein an externally input DC voltage is applied to both ends of the DC support capacitor (1) to form a DC bus; the three bridge inverter units have the same circuit topology, each bridge inverter unit is composed of four inverter bridge arms connected in parallel to the DC bus, each inverter bridge arm is composed of two series-connected switch devices, and two groups of inverter bridge arms constitute an H-bridge inverter unit; The A-phase bridge inverter unit (2) comprises a first switch device (20) and a second switch device (21) connected in series to form a first inverter bridge arm, a third switch device (22) and a fourth switch device (23) connected in series to form a second inverter bridge arm, and the two inverter bridge arms form an H-bridge inverter unit connected in parallel with a DC bus; another H-bridge inverter unit formed by a fifth switch device (24) and a sixth switch device (25) connected in series, and a seventh switch device (26) and an eighth switch device (27) connected in series is connected in parallel with the DC bus; one of the two H-bridge inverter units is connected to a first winding (50) on the primary side of the A-phase, and the other is connected to a second winding (51) on the primary side of the A-phase; the PWM voltages output by the two H-bridge inverter units are superimposed in series through transformer windings, and output at the A-phase secondary winding (52) to obtain a dualized PWM voltage; the dualized PWM voltage is input to a three-phase AC filter (6) of the inverter, and after filtering, an industrial frequency output voltage is obtained; The modulation strategy of the marine microgrid inverter is as follows: the phase difference between the first inverter bridge arm triangle carrier and the second inverter bridge arm triangle carrier, and the third inverter bridge arm triangle carrier and the fourth inverter bridge arm triangle carrier of each phase bridge inverter unit is 180°, and the phase difference between the first inverter bridge arm triangle carrier and the third inverter bridge arm triangle carrier, and the second inverter bridge arm triangle carrier and the fourth inverter bridge arm triangle carrier of each phase bridge inverter unit is 90°; taking the phase of the first inverter bridge arm triangle carrier of the A-phase bridge inverter unit as a reference, the phase difference between each inverter bridge arm triangle carrier of the B-phase bridge inverter unit and each inverter bridge arm triangle carrier corresponding to the A-phase bridge inverter unit is α, and the phase difference between each inverter bridge arm triangle carrier of the C-phase bridge inverter unit and each inverter bridge arm triangle carrier corresponding to the A-phase bridge inverter unit is β.

2. The modulation strategy of the marine microgrid inverter according to claim 1 is characterized in that: The α=180° and β=180°.

3. A control method for a marine microgrid inverter according to claim 1, characterized in that: The sampled three-phase voltage u oa (t),u ob (t) and u oc (t) The positive sequence fundamental voltage of the d-axis in the positive synchronous rotating coordinate system is obtained by decomposing the positive sequence and negative sequence fundamental voltage of the DSC and performing dq transformation. Q-axis positive sequence fundamental voltage per unit value Unit value of negative sequence fundamental voltage of d-axis in counter-rotating synchronous coordinate system And the q-axis positive sequence fundamental voltage per unit value and As the feedback value of the positive sequence voltage controller (82), it participates in the inverter output positive sequence voltage control. and As the feedback value of the negative sequence voltage controller (83), it participates in the inverter output negative sequence voltage control; the d-axis per unit voltage reference value of the positive sequence voltage controller (82) is 1, minus the feedback value The positive sequence voltage controller (82) is then input to the d-axis positive sequence voltage PI regulator (821). The q-axis per unit voltage reference value of the positive sequence voltage controller (82) is 0, minus the feedback value Then input to the q-axis positive sequence voltage PI regulator (822) of the positive sequence voltage controller (82); The d-axis per unit voltage reference value of the negative sequence voltage controller (83) is 0, minus the feedback value The output of the d-axis negative sequence voltage PI regulator (831) is then input to the negative sequence voltage controller (83). Multiply by the cross decoupling term ωC f After subtraction, the output is sent to the d-axis input port of the negative-sequence dq to positive-sequence dq conversion module (86); the q-axis per-unit voltage reference value of the negative-sequence voltage controller (83) is 0, minus the feedback value The output of the q-axis negative sequence voltage PI regulator (832) is then input to the negative sequence voltage controller (83). Multiply by the cross decoupling term ωC f After subtraction, the output is sent to the q-axis input port of the negative-sequence dq to positive-sequence dq conversion module (85); The negative sequence dq to positive sequence dq conversion module (85) converts -ω o The dq component of the negative sequence dq coordinate system of speed rotation passes through -2ω o Transform to ω o In the positive sequence dq coordinate system of speed rotation, the transformed dq axis component is superimposed with the dq axis component output of the positive sequence voltage controller (82) to obtain the d axis current reference value of the current controller (84) and q-axis current reference value Add the d-axis output current Feedforward minus d-axis inductor current feedback The input is to the d-axis current PI+R regulator (843) which is formed by adding a resonant control link on the basis of the PI regulator; the q-axis current Cross-decoupling term ωL m And the output voltage The feedforward term is then superimposed with the output of the third harmonic proportional resonant PR controller (841) of the transformer d-axis primary zero-sequence current to obtain the modulation wave d-axis component u d ; Add the q-axis output current Feedforward minus q-axis inductor current feedback Input to the q-axis current PI+R regulator (844); d-axis current Cross-decoupling term ωL m And the output voltage The feedforward term is then superimposed with the output of the third harmonic proportional resonant PR controller (842) of the transformer q-axis primary zero-sequence current to obtain the modulation wave q-axis component u q ; For the modulated wave d-axis component u d With the q-axis component u q Perform the dq inverse transformation to obtain the ABC three-phase modulation wave voltage u A 、u B and u C , and then the driving pulse of the corresponding switching device is obtained through SPWM modulation.

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