Ship propulsion motor sensorless control algorithm based on harmonic suppression

Abstract

The invention provides a ship propulsion motor sensorless control algorithm based on harmonic suppression, and the algorithm comprises the steps: introducing a compensation coefficient through the analysis of fifth and seventh harmonics, constructing a novel PIR controller, and suppressing the formulated frequency harmonics; the external disturbance of the system is regarded as a system expansion state, a novel observer is constructed by combining cascade ESO and PLL to estimate the position of a motor rotor, it is guaranteed that the system accurately compensates for the deviation angle in a dynamic state and a static state, and the performance of the position-sensorless driving system is improved. According to the invention, a compensation coefficient is introduced, a novel PIR controller is constructed, and control links for five-frequency multiplication and seven-frequency multiplication are added, so that the effects of suppressing PMSM phase current harmonics and reducing the total harmonic content of phase current are achieved. And then external disturbance of the system is regarded as a system expansion state, and the rotor angle of the motor is estimated by combining cascade ESO and PLL. Therefore, the system can accurately compensate the deviation angle in a steady state and a dynamic state, and the performance of the position-sensorless driving system is improved.

Description

Technical Field

[0001] The present invention relates to the technical field of permanent magnet synchronous motor control, and in particular to a position sensorless control algorithm for a ship propulsion motor based on harmonic suppression. Background Art

[0002] With the development of green power technology for ships, electric propulsion ships are becoming more and more mature. Permanent magnet synchronous propulsion motors, as the core of their propulsion systems, have the advantages of simple structure, high efficiency, and good reliability. The development of position sensorless control technology has solved the problems of high cost, large size, and low reliability of traditional mechanical sensors. Traditional ESO control can achieve fast dynamic response with high accuracy, but it handles acceleration disturbances slowly and even causes error accumulation. In addition, compared with land motors, the operating conditions of ship propulsion motors are more complicated, the motor system is affected by many internal and external unmodeled disturbances, and structures such as inverters will bring problems such as current harmonics. Traditional ESO control algorithms cannot meet actual needs. Therefore, it is necessary to seek new position sensorless control algorithms to meet the requirements of motor control systems for multi-condition response speed and estimation accuracy. Summary of the invention

[0003] In view of this, in order to meet the requirements of the motor control system for high convergence speed and estimation accuracy, the present invention provides a sensorless control algorithm for ship propulsion motors based on harmonic suppression, which performs transformation processing on the basis of traditional bilinear transformation, introduces compensation coefficients, constructs a new type of PIR controller, and adds control links for five and seven times the frequency, so as to achieve the effect of suppressing PMSM phase current harmonics and reducing the total harmonic content of phase current. The external disturbance to the system is then regarded as the system expansion state, and the cascaded ESO is combined with the PLL to estimate the rotor angle of the motor. The system can accurately compensate for the deviation angle in steady state and dynamic state, further improving the performance of the sensorless drive system.

[0004] To achieve the above object, the present invention provides the following technical solutions:

[0005] A position sensorless control algorithm for a ship propulsion motor based on harmonic suppression comprises the following steps:

[0006] By introducing compensation coefficients for the fifth and seventh harmonics analysis, a new PIR controller is constructed to suppress harmonics of designated frequencies.

[0007] The external disturbance to the system is regarded as the extended state of the system, and a new observer is constructed by combining the cascaded ESO and PLL to estimate the motor rotor position, ensuring that the system accurately compensates the deviation angle in dynamic and static conditions, thereby improving the performance of the position sensorless drive system.

[0008] Preferably, the method specifically comprises the following steps:

[0009] Step (1), designing a voltage model flux observer, and building a mathematical model of the ship's permanent magnet synchronous motor in an integral form in the α-β coordinate system;

[0010] Step (2), designing a current model flux observer, obtaining the stator flux in the α-β coordinate system according to the stator flux transformation under the dq axis, and building a hybrid flux observer in the full speed range by combining the advantages of both;

[0011] Step (3), for the hybrid flux observer in step (2), a phase-locked loop structure based on cascaded ESO is used to compensate the rotor position to obtain a rotor deviation angle;

[0012] Step (4), for the rotor position information obtained by the cascade ESO in step (3), the harmonic influence caused by the nonlinearity of the inverter can be obtained by analyzing it using the sum-difference formula;

[0013] Step (5), in view of the harmonic influence in step (4), an improved proportional integral resonant regulator is designed in the current loop to suppress the harmonic;

[0014] Step (6), combined with the improved proportional-integral resonant regulator in step (5), analyzes that the regulator can compensate for the fifth and seventh harmonic components while performing error-free tracking of the system fundamental current component.

[0015] Preferably, in step (1), the mathematical model of the permanent magnet synchronous motor in integral form in the α-β coordinate system is:

[0016]

[0017] In the formula, R s is the stator resistance; u α ,u β and i α ,i β are the stator voltage and current under α-β axis respectively; L s is the stator inductance; ω e is the electrical angular velocity; ψ f is the permanent magnet flux; θ is the electrical angle of the rotor position, ψ sα , ψ sβ is the stator flux of the α-β axis, ψ rα , ψ rβ is the rotor flux of the α-β axis.

[0018] Preferably, the voltage model flux observer constructed based on the rotor position information in the rotor flux is:

[0019]

[0020] The current model flux observer constructed based on the rotor position information in the rotor flux is:

[0021]

[0022] Preferably, in step (3), the PLL structure is changed to a PLL structure based on the cascaded ESO based on the hybrid flux observer, and the disturbance observed by ESO1 is added to ESO2 as a known quantity to compensate for the rotor position;

[0023] The first layer of the designed cascade ESO is the same as the design of the conventional ESO, which is:

[0024]

[0025] The disturbance observed by ESO1 is added to ESO2 as a known quantity, and the second-layer ESO is:

[0026]

[0027] In the formula, State variable x1 = i α , x2 = f; z1 is the observed estimated value of the system state variable x1; z2 is the observed estimated value of the system state variable x2, that is, the estimated value of the total disturbance f of the system; β1, β2 are the gains of the extended state observer;

[0028] The gains β1, β2, β3, and β4 are obtained based on the pole configuration method of ESO, using the bandwidth tuning method, and are taken as:

[0029] Then the rotor deviation angle is obtained as:

[0030]

[0031] In the formula, θ * is the estimated rotor position; is the observed value of the rotor flux on the α-β axis.

[0032] Preferably, in step (4), the rotor position information is:

[0033]

[0034] In the formula, ψ1, ψ 6k±1 are the fundamental amplitude and (6k±1)th harmonic amplitude of the rotor flux respectively; θ1, θ (6k±1) They are the fundamental phase and (6k±1)th harmonic phase of the rotor flux respectively; is the estimated value of the rotor position.

[0035] Preferably, in step (5), the s-function of the improved proportional-integral resonant regulator is:

[0036]

[0037] Then the transfer function is:

[0038]

[0039] Where ω0 is the resonant frequency ω0 = 6 ω e ; K P is the proportional gain; K i is the resonance gain; ω c is the cut-off frequency of the quasi-resonant controller; T s is the sampling period.

[0040] Preferably, in step (6), the fifth and seventh harmonic components are compensated as follows:

[0041]

[0042] Preferably, the differential equation of the improved proportional integral resonant regulator is:

[0043] u * (k)=(λ0 * +λ 0，n＝5 * +λ 0，n＝7 * ) i (k)+(λ2 * +λ 2，n＝5 * +λ 2，n＝7 * ) i (k-2)-(ξ1 * +ξ 1，n＝5 * +ξ 1，n＝7 * )u(k-1)-(ξ2 * +ξ 2，n＝5 * +ξ 2，n＝7 * )u(k-2).

[0044] Compared with the prior art, the present invention has the following beneficial effects:

[0045] The position sensorless control method based on cascade ESO and improved PIR disclosed in the present invention can realize the stable operation of ship propulsion motors under uncertain disturbances such as wind and waves. The present invention first proposes a hybrid flux observer to realize rotor position estimation in the full-speed domain, and changes the PLL part thereof to a PLL design based on cascade ESO, so as to ensure more accurate prediction and compensation of system state changes and reduce the impact of error accumulation. Then, a new PIR is designed in the current loop to perform harmonic analysis on the estimated rotor position. The present invention is more accurate in estimating the rotor position within the full-speed domain of the motor, and is superior in high-order harmonic suppression, which fully meets the requirements of the ship's three-phase permanent magnet synchronous propulsion motor used in the present invention for use under uncertain disturbances such as wind, waves and currents in water. BRIEF DESCRIPTION OF THE DRAWINGS

[0046] Figure 1 It is a structural block diagram of a hybrid flux observer for a voltage and current model of a permanent magnet synchronous motor in the present invention;

[0047] Figure 2 It is a block diagram of the phase-locked loop rotor position estimation based on the cascaded ESO in the present invention;

[0048] Figure 3 It is the overall structural block diagram of the novel position sensorless control method in the present invention;

[0049] Figure 4 is the bode diagram of the PIR controller in the present invention, wherein a is the bode diagram of the traditional PIR controller, and b is the bode diagram of the improved PIR controller;

[0050] Figure 5 It is a spectrum analysis diagram in the present invention, wherein a is the FFT result of the original system without compensation, b is the FFT result based on the traditional PIR control compensation, and c is the FFT result based on the improved PIR control compensation;

[0051] Figure 6 This is the motor speed diagram at a constant speed of 1000r / min in the present invention;

[0052] Figure 7 This is the motor speed diagram at a constant speed of 1000r / min in the present invention;

[0053] Figure 8 This is a speed error diagram when a sudden step and ramp disturbance is added in the present invention;

[0054] Fig. 9 This is a diagram of the rotor position error when a sudden step and ramp disturbance is added in the present invention;

[0055] Fig.10 It is a speed error diagram when a sudden acceleration disturbance occurs in the present invention;

[0056] Fig.11 This is a diagram of the rotor position error during sudden acceleration disturbance in the present invention. DETAILED DESCRIPTION

[0057] The present invention provides a position sensorless control algorithm for a ship propulsion motor based on harmonic suppression, specifically a position sensorless control method based on cascaded ESO and improved PIR applied to a ship permanent magnet synchronous propulsion motor.

[0058] Terminology explanation:

[0059] Cascade ESO: Cascade ESO is a technique used in computer science and engineering to improve the system's anti-interference ability and control accuracy. Specifically, Cascade ESO (Extended State Observer) is a method of observing the residual disturbance of the system, which is not affected by the system's measurement noise, thereby improving the system's anti-interference ability. This technique performs well in areas such as motor motion control, especially in improving the system's anti-interference performance.

[0060] The implementation of cascaded ESO relies on an improved active disturbance rejection control strategy (SnADRC), which uses a sequential output stacked extended state observer (SOS-ESO) to observe complex disturbances step by step, thereby improving the estimation accuracy of complex disturbances and the control accuracy of the system. The innovation of this method is that it first proposes a way to select the number of observers and establishes an intrinsic connection between the total disturbance order and the number of observers to enhance the system's anti-disturbance performance. Verified by simulation and experimental results, SnADRC has shown excellent performance in motor motion control.

[0061] PLL: Phase-locked loop.

[0062] PIR regulator: Proportional Integral Resonant Regulator.

[0063] PMSM: Permanent Magnet Synchronous Motor.

[0064] ESO: ExtendedStateObserver, extended state observer.

[0065] ESO1: ExtendedStateObserver1, the first layer in the cascade ESO.

[0066] ESO2: ExtendedStateObserver2, the second layer in the cascade ESO.

[0067] In order to more clearly illustrate the present invention or the technical solution in the prior art, the present invention will be explained in detail below in conjunction with the accompanying drawings, as follows:

[0068] The mathematical model of the ship's permanent magnet synchronous propulsion motor in the α-β coordinate system is:

[0069]

[0070] In the formula, R s is the stator resistance; u α ,u β and i α ,i β are the stator voltage and current under α-β axis respectively; L s is the stator inductance; w e is the electrical angular velocity; f is the permanent magnet flux; θ is the electrical angle of the rotor position. The PMSM flux equation is:

[0071]

[0072] Among them, sα ,Ψ sβ is the stator flux of the α-β axis.

[0073] Therefore, after taking the derivatives on both sides of formula (2) and integrating them, combined with formula (1), we can get the following integral form:

[0074]

[0075] According to the conversion relationship between the rotor flux and the stator flux of the permanent magnet motor, the rotor flux can be obtained as:

[0076]

[0077] Among them, rα ,Ψ rβ is the rotor flux of the α-β axis.

[0078] The rotor flux contains information about the rotor position, so the flux observer based on the voltage model is:

[0079]

[0080] Similarly, the flux observer based on the current model can be obtained as:

[0081]

[0082] Through the analysis of these two flux observers, the voltage model and the current model are connected together through the PI regulator. Combining the advantages of both, the optimal flux estimation value can be provided for the system in the full speed range. Figure 1 shown.

[0083] In order to solve the problem that the traditional ESO cannot handle acceleration disturbances, a cascade extended state observer is used on the basis of the original ESO to improve the disturbance observation capability. The basic idea is: using the disturbance observation characteristics of ESO, two ESOs, namely ESO1 and ESO2, are cascaded. First, the disturbance in the system is observed by ESO1, and the observed disturbance is used as the known part of ESO2. Then ESO2 only needs to observe the remaining part of the total disturbance, and the disturbance observed by the two is used as the total disturbance to compensate to the output of the controller, so as to achieve the purpose of eliminating the disturbance.

[0084] The mathematical model of the α-β voltage equation of formula (1) is transformed into the following α-β current differential equation:

[0085]

[0086] Taking the α-axis as an example, the standard state space equation of the α-axis current in the above formula is established as:

[0087] x.1=ax1+bu+f (8)

[0088] in, u=u α , take the state variable x1=i α ,x2=f,y is the coefficient

[0089] The system state equation can be obtained as follows:

[0090]

[0091] According to the linear extended state observer theory, its differential equation model can be obtained:

[0092]

[0093] The design of ESO1 is the same as that of conventional ESO. The state equation of the designed ESO1 is as follows:

[0094]

[0095] The perturbation observed by ESO1 is added to ESO2 as a known quantity:

[0096]

[0097] Gains β1, β2, β3, and β4 are based on the ESO pole placement method and bandwidth tuning method, and are taken as:

[0098]

[0099] Then, the deviation angle is obtained as:

[0100]

[0101] After analyzing the obtained magnetic flux, we can get

[0102]

[0103] Where Ψ1 and Ψ6k±1 are the fundamental amplitude and (6k±1)th harmonic amplitude of the rotor flux, respectively; θ1 and θ(6k±1) are the fundamental phase and (6k±1)th harmonic phase of the rotor flux, respectively.

[0104] The rotor position error signal is obtained using the sum-difference formula:

[0105]

[0106] According to formula (16), it can be clearly seen that the harmonic influence caused by the nonlinearity of the inverter, the obtained rotor position estimation error signal contains two parts in total, one part is the actual position estimation error Δθ, and the other part is the error influence caused by high-order harmonics.

[0107] Based on the traditional bilinear transformation, deformation processing is performed and the compensation coefficient is defined as ζ. The improved bilinear transformation method is obtained as follows:

[0108]

[0109] In order to analyze the resonant frequency deviation after the improved bilinear transformation is discretized, it is necessary to calculate the actual resonant frequency after discretization. The transfer function pole equation in the discrete domain is:

[0110]

[0111] The two poles of the transfer function in the discrete domain can be obtained as:

[0112]

[0113] According to the corresponding relationship of the discrete system z plane angular frequency, z = wT S , we can get the actual resonant angular frequency w corresponding to the resonant angular frequency obtained by the transfer function on the continuous system real for:

[0114]

[0115] In order to make the actual resonant angular frequency equal to the expected resonant frequency w0, and because the bandwidth w in the actual system c It is much smaller than w0, so it can be ignored here. Therefore, we can solve:

[0116]

[0117] Thus, the new s-function can be obtained as:

[0118]

[0119] The transfer function of the improved PIR controller is:

[0120]

[0121] Similarly, we can get:

[0122]

[0123] Therefore, the new difference equation is:

[0124] u * (k)=(λ0 * +λ 0，n＝5 * +λ 0，n＝7 * ) i (k)+(λ2 * +λ 2，n＝5 * +λ 2，n＝7 * ) i (k-2)-(ξ1 * +ξ 1，n＝5 * +ξ 1，n＝7 * )u(k-1)-(ξ2 * +ξ 2，n＝5 * +ξ 2，n＝7 * )u(k-2) (25)

[0125] The bode diagram of the PIR controller is as follows Figure 4 As shown. Figure 4 From (a) in the figure, it can be seen that the PIR controller has a high gain at the fundamental angular frequency ω0, so it can achieve zero-error tracking of the fundamental frequency AC signal; at the same time, the controller has a large bandwidth and can maintain a high gain within a certain range when the frequency is offset. However, this method has a small gain at the fundamental frequency multiple, so the PIR controller has limited ability to suppress current harmonic content. Figure 4 It can be seen from (b) that the improved PIR controller has a large gain at the fundamental wave and also has a high gain at the 5th and 7th harmonics. Therefore, the improved PIR controller can track the fundamental current component of the system without difference while also compensating for the 5th and 7th harmonic components.

[0126] Finally, the permanent magnet synchronous motor control system for electric propulsion ships is modeled and simulated. Figure 1The motor parameter table is shown in Table 1:

[0127] Table 1 Motor parameters required for simulation

[0128]

[0129] Figure 5 The FFT analysis results of the three-phase current harmonics of the PMSM under the original system without compensation, based on traditional PIR control compensation, and the improved PIR control compensation proposed in this patent are given respectively.

[0130] Under steady-state operation at a speed of 1000r / min, the motor fundamental frequency in FFT is 66.6Hz. The parameters of PIR are set as: Ki=1000, wc=5, w0=6we, Ts=1e-4. Figure 5 From (a) in the figure, it can be seen that when no compensation control is adopted, the 5th and 7th harmonic contents of the phase current are 5.94% and 2.06% respectively, and the THD is 6.30%; when the traditional PIR control is adopted, Figure 5 The 5th and 7th harmonic contents of phase current in (b) are reduced to 0.95% and 0.19%, respectively, and the THD is 2.24%. When the improved PIR current loop control is adopted, Figure 5 (c) The 5th and 7th harmonic contents of the phase current are reduced to 0.28% and 0.11%, and the THD is 0.34%.

[0131] In order to verify whether the improved PIR is effective in estimating the motor rotor position, the motor is set to no-load and the given speed is 1000r / min. The steady-state performance of the motor with and without the PIR controller is compared. Figure 6 Motor speed and Figure 7 The rotor position error can clearly show that the motor speed graph presents a more stable and continuous feature, with higher stability compared to the original result. In addition, the motor rotor position error graph shows a significant improvement, indicating that the improved PIR control technology can more effectively reduce the position estimation error and improve the accuracy and reliability of the system. This further verifies the effectiveness of the improved PIR technology in motor rotor position estimation.

[0132] In order to further verify the effectiveness of the cascade ESO on the system's anti-disturbance ability, the improved cascade ESO is compared with the traditional ESO based on the PIR control method. In ESO, w=100. When the motor runs to 0.15s, the given speed is changed to 1300r / min to simulate motor acceleration. When the motor runs to 0.25s, the given speed is changed to 800r / min to simulate motor deceleration. And when the motor runs to 0.2s, the load torque is changed to 10 to simulate a sudden load. When the motor runs to 0.45s, an acceleration load with a slope of 20 and an initial output of 10 is added. In practical applications, for example, when the control system where the motor is located needs to switch to a different operating mode or control scheme. It may cause a sudden change in load, causing the motor to bear different loads. The above situation is regarded as a step and ramp disturbance to the system simulation. By observing Figure 8 , Fig. 9 The results show that the motor speed fluctuation range of the traditional ESO is ±0.5r / min, the overshoot is 0.05%, and the rotor position error range is ±0.14, while the motor speed fluctuation range of the cascade ESO is ±0.2r / min, the overshoot is 0.02%, and the rotor position error is ±0.001. It can be seen that when sudden step and ramp disturbances are added, the system can operate stably in both ESO cases, but the improved cascade ESO control shows more superior characteristics than the traditional ESO.

[0133] When the motor runs to 0.2s, a load with an acceleration of 20 is added to simulate the environment where the motor is located, which may be subject to external interference, such as wind, waves, etc. These external interferences can be modeled as a disturbance signal in the shape of a sudden acceleration that affects the running state of the motor. Fig.10 , 11 It can be seen that the traditional ESO cannot keep the system stable after a sudden acceleration disturbance when the disturbance reaches a certain value, while the improved cascade ESO can significantly ensure the stability of the system after a sudden acceleration disturbance. It can be seen that the disturbance estimation method using the cascade ESO has zero steady-state estimation error for step, ramp, and acceleration disturbances, and has stronger disturbance suppression capabilities.

[0134] The above description is only a preferred specific implementation manner of the present invention, but the protection scope of the present invention is not limited thereto. Any technician familiar with the technical field can make equivalent replacements or changes according to the technical scheme and inventive concept of the present invention within the technical scope disclosed by the present invention, which should be covered by the protection scope of the present invention.

Claims

1. A sensorless control algorithm for ship propulsion motor based on harmonic suppression, characterized in that: The steps include: By introducing compensation coefficients for the fifth and seventh harmonics analysis, a new PIR controller is constructed to suppress harmonics of designated frequencies. The external disturbance to the system is regarded as the extended state of the system, and a new observer is constructed by combining the cascaded ESO and PLL to estimate the motor rotor position, ensuring that the system accurately compensates the deviation angle in dynamic and static conditions, thereby improving the performance of the position sensorless drive system.

2. According to the harmonic suppression-based position sensorless control algorithm for ship propulsion motors according to claim 1, it is characterized in that: The specific steps include: Step (1), designing a voltage model flux observer, and building a mathematical model of the ship's permanent magnet synchronous motor in an integral form in the α-β coordinate system; Step (2), designing a current model flux observer, obtaining the stator flux in the α-β coordinate system according to the stator flux transformation under the dq axis, and building a hybrid flux observer in the full speed range by combining the advantages of both; Step (3), for the hybrid flux observer in step (2), a phase-locked loop structure based on cascaded ESO is used to compensate the rotor position to obtain a rotor deviation angle; Step (4), for the rotor position information obtained by the cascade ESO in step (3), the harmonic influence caused by the nonlinearity of the inverter can be obtained by analyzing it using the sum-difference formula; Step (5), in view of the harmonic influence in step (4), an improved proportional integral resonant regulator is designed in the current loop to suppress the harmonic; Step (6), combined with the improved proportional-integral resonant regulator in step (5), analyzes that the regulator can compensate for the fifth and seventh harmonic components while performing error-free tracking of the system fundamental current component.

3. According to the harmonic suppression-based position sensorless control algorithm for ship propulsion motors according to claim 2, it is characterized in that: In step (1), the mathematical model of the permanent magnet synchronous motor in the integral form in the α-β coordinate system is: In the formula, R s is the stator resistance; u α ,u β and i α ,i β are the stator voltage and current under α-β axis respectively; L s is the stator inductance; ω e is the electrical angular velocity; ψ f is the permanent magnet flux; θ is the electrical angle of the rotor position, ψ sα , ψ sβ is the stator flux of the α-β axis, ψ rα , ψ rβ is the rotor flux of the α-β axis.

4. According to the harmonic suppression-based position sensorless control algorithm for ship propulsion motors of claim 3, it is characterized in that: The voltage model flux observer constructed based on the rotor position information in the rotor flux is: The current model flux observer constructed based on the rotor position information in the rotor flux is:

5. According to the harmonic suppression-based position sensorless control algorithm for ship propulsion motors in claim 2, it is characterized in that: In step (3), the PLL structure is changed to a PLL structure based on the cascaded ESO based on the hybrid flux observer, and the disturbance observed by ESO1 is added to ESO2 as a known quantity to compensate the rotor position; The first layer of the designed cascade ESO is the same as the design of the conventional ESO, which is: The disturbance observed by ESO1 is added to ESO2 as a known quantity, and the second-layer ESO is: In the formula, State variable x1 = i α , x2 = f; z1 is the observed estimated value of the system state variable x1; z2 is the observed estimated value of the system state variable x2, that is, the estimated value of the total disturbance f of the system; β1, β2 are the gains of the extended state observer; The gains β1, β2, β3, and β4 are obtained based on the pole configuration method of ESO and the bandwidth tuning method, and are taken as: Then the rotor deviation angle is obtained as: In the formula, θ * is the estimated rotor position; is the observed value of the rotor flux on the α-β axis.

6. The sensorless control algorithm for ship propulsion motor based on harmonic suppression according to claim 2 is characterized in that: In step (4), the rotor position information is: In the formula, ψ1, ψ 6k±1 are the fundamental amplitude and (6k±1)th harmonic amplitude of the rotor flux respectively; θ1,θ (6k±1) They are the fundamental phase and (6k±1)th harmonic phase of the rotor flux respectively; is the estimated value of the rotor position.

7. The sensorless control algorithm for ship propulsion motor based on harmonic suppression according to claim 2 is characterized in that: In step (5), the s-function of the improved proportional integral resonant regulator is: Then the transfer function is: Where ω0 is the resonant frequency ω0 = 6 ω e ; K P is the proportional gain; K i is the resonance gain; ω c is the cut-off frequency of the quasi-resonant controller; T s is the sampling period.

8. The sensorless control algorithm for ship propulsion motor based on harmonic suppression according to claim 7 is characterized in that: In step (6), the fifth and seventh harmonic components are compensated as follows:

9. The sensorless control algorithm for ship propulsion motor based on harmonic suppression according to claim 7 is characterized in that: The difference equation of the improved proportional integral resonant regulator is:

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