Speed sensorless control method for permanent magnet motor

Abstract

The invention provides a permanent magnet motor speed sensorless control method, which comprises the following steps of: firstly, designing a sliding mode observer to carry out back electromotive force observation; secondly, a low-pass filter is added for sliding mode noise suppression, and a permanent magnet motor back electromotive force observation signal with phase deviation is obtained; and then, designing a self-adaptive phase-locked loop, and carrying out phase locking on the back electromotive force observation signal of the permanent magnet motor with phase deviation to realize accurate observation and self-adaptive compensation of the rotor position angle of the permanent magnet motor. By designing the adaptive phase-locked loop, adaptive compensation of the rotor position angle of the permanent magnet motor can be realized, and the compensation method is not influenced by a rotating speed estimation error, so that the estimation precision of the rotor position angle is improved. Meanwhile, the low-pass filter is added in the self-adaptive phase-locked loop, so that sliding mode noise can be better filtered, smoother rotating speed and rotor position angle can be obtained, and the speed sensorless control precision of the permanent magnet motor is improved.

Description

Technical Field

[0001] The invention relates to the technical field of power electronics, and in particular to a speed sensorless control method for a permanent magnet motor. Background Art

[0002] In recent years, with the rapid development of technologies such as multi-rotor drones and new energy vehicles, permanent magnet motors have been widely used due to their high efficiency and high reliability. Under complex operating conditions such as strong electromagnetic interference, permanent magnet motors are prone to encoder disconnection and other faults, resulting in loss of speed and rotor position signals, affecting the operating reliability of permanent magnet motors. For this reason, the speed sensorless control technology of permanent magnet motors has received extensive attention and research. The conventional speed sensorless control method of permanent magnet motors based on sliding mode observers has the problem that the rotor position estimation is affected by the speed estimation error. As a result, once there is an estimation error in the speed, the rotor position will also have a significant error, affecting the operating stability of the speed sensorless control of the permanent magnet motor.

[0003] Reference [Lu Wenqi, Hu Yuwen, Du Xuyang, et al. Sensorless vector control speed control system of permanent magnet synchronous motor with new sliding mode observer [J]. Proceedings of the CSEE, 2010, 30(33):78-83.] proposed a speed sensorless control method for permanent magnet motor based on a new sliding mode observer, which improved the low-speed operation stability of the motor. However, this method requires the use of estimated speed for rotor position angle compensation, resulting in the estimated rotor position angle being affected by the estimated speed accuracy. Reference [Zhu Dehong, Zhao Mingyu. IPMSM full-speed domain position sensorless control strategy [J]. Combined Machine Tool and Automation Processing Technology, 2024, (03):96-100.] designed a speed sensorless control method for permanent magnet motor based on super-helical sliding mode observer, but since this method still uses a low-pass filter for sliding mode noise suppression, it also requires the use of estimated speed for rotor position angle compensation, resulting in the estimated rotor position angle being affected by the estimated speed accuracy. Reference [Zhang Qiang, Ma Qiang, Zhao Xianxin, et al. An improved permanent magnet synchronous motor speed sensorless control method [P]. Hubei Province: CN202010077705.9, 2023-05-02.] designed an improved permanent magnet motor speed sensorless control method, using nonlinear functions instead of sign functions to weaken the sliding mode chattering, and designed the approach rate to accelerate the sliding mode convergence speed. However, this method still uses a low-pass filter for sliding mode noise suppression, so it is also necessary to use the estimated speed for rotor position angle compensation, resulting in the estimated rotor position angle being affected by the estimated speed accuracy. Summary of the invention

[0004] In view of the problem that the rotor position angle estimation of traditional permanent magnet motor speed sensorless control is affected by the speed estimation, the present invention proposes a permanent magnet motor speed sensorless control method. By designing an adaptive phase-locked loop, adaptive compensation of the rotor position angle is achieved, thus overcoming the problem that the rotor position angle estimation accuracy is affected by the speed estimation accuracy.

[0005] In order to achieve the above object, the technical solution of the present invention is achieved as follows:

[0006] A speed sensorless control method for a permanent magnet motor, the steps of which are as follows:

[0007] Step 1: Use the voltage sensor to collect the three-phase voltage of the permanent magnet motor stator, including the a-phase voltage u a 、b phase voltage u b and c phase voltage u c , and the phase voltage u a 、u b 、u c Transformed to the two-phase stationary αβ coordinate system, the stator voltage u is obtained α 、u β ;

[0008] Step 2: Use current sensors to collect the three-phase current of the permanent magnet motor, including the a-phase current i a , b-phase current i b and c-phase current i c , and the current i a 、i b 、i c Transformed to the two-phase stationary αβ coordinate system, the current i is obtained α 、i β ;

[0009] Step 3: The stator voltage u obtained in step 1 α 、u β And the current i obtained in step 2 α 、i β , construct the following sliding mode observer, satisfying:

[0010]

[0011] in, represents the α-axis component of the estimated current, represents the β-axis component of the estimated current, represents the α-axis component of the estimated back EMF, represents the β-axis component of the estimated back electromotive force, k represents the sliding mode gain, sgn() is the sign function, L s Represents the stator inductance of the permanent magnet motor, R s Represents the stator resistance of the permanent magnet motor;

[0012] Step 4: Convert the back EMF estimated in step 3 After passing through a low-pass filter to remove the sliding mode noise, the filtered back electromotive force is obtained.

[0013] Step 5: Filtered back electromotive force obtained in step 4 Calculate the back EMF error e q ;

[0014] Step 6: Substitute the back EMF error e obtained in step 5 q After the proportional integral regulator, the estimated speed is obtained.

[0015] satisfy:

[0016]

[0017] Among them, k p is the proportional coefficient of the proportional-integral controller, k i is the integral coefficient of the proportional-integral controller, s represents the Laplace operator;

[0018] Step 7: The estimated speed obtained in step 6 After the integration link, the estimated rotor position angle is obtained satisfy:

[0019]

[0020] Step 8: Repeat steps 5 to 7 until the back EMF error e q When it converges to zero, the estimated speed of the permanent magnet motor is obtained and rotor position angle Will and As feedback of the permanent magnet motor vector control system, the speed sensorless control of the permanent magnet motor can be realized.

[0021] Preferably, the voltage u α 、u β The calculation method is:

[0022]

[0023] Among them, u α is the α-axis component of the three-phase stator voltage of the permanent magnet motor in the stationary αβ coordinate system, u β It is the β-axis component of the stator three-phase voltage of the permanent magnet motor in the stationary αβ coordinate system.

[0024] Preferably, the current i α 、i βThe calculation method is:

[0025]

[0026] Among them, i α is the α-axis component of the three-phase current of the permanent magnet motor in the stationary αβ coordinate system, i β It is the β-axis component of the three-phase current of the permanent magnet motor in the stationary αβ coordinate system.

[0027] Preferably, the filtered back electromotive force satisfy:

[0028]

[0029] Where s represents the Laplace operator, ω c Indicates the cutoff frequency of the low-pass filter.

[0030] Preferably, the back electromotive force error e q satisfy:

[0031]

[0032] Among them, e q is the back EMF error, is the estimated rotor position angle. When step 5 is executed for the first time, is zero, starting from the second execution of step 5, Obtained from step seven.

[0033] Beneficial effects of the present invention: The present invention designs an adaptive phase-locked loop, by adding the same low-pass filter used to filter out sliding mode noise in a conventional phase-locked loop, so that the rotor position angle estimation error caused by the low-pass filter is automatically offset, thereby realizing adaptive compensation of the rotor position angle that is independent of the estimated speed, and improving the accuracy of the rotor position angle estimation. At the same time, since a low-pass filter is added to the adaptive phase-locked loop, the sliding mode noise can also be further suppressed, so that a smoother estimated speed and rotor position angle can be obtained, and finally the speed sensorless control performance of the permanent magnet motor is improved. BRIEF DESCRIPTION OF THE DRAWINGS

[0034] In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the drawings required for use in the embodiments or the description of the prior art will be briefly introduced below. Obviously, the drawings described below are only some embodiments of the present invention. For ordinary technicians in this field, other drawings can be obtained based on these drawings without paying creative work.

[0035] Figure 1 This is the control block diagram of the conventional method.

[0036] Figure 2 This is a control block diagram of the method proposed in the present invention.

[0037] Figure 3 These are the experimental results of the conventional method.

[0038] Figure 4 These are the experimental results of the method proposed in the present invention. DETAILED DESCRIPTION

[0039] The following will be combined with the drawings in the embodiments of the present invention to clearly and completely describe the technical solutions in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0040] like Figure 2 As shown, the embodiment of the present invention provides a permanent magnet motor speed sensorless control method, and the specific steps are as follows:

[0041] Step 1: Use the voltage sensor to collect the three-phase voltage of the permanent magnet motor stator, including the a-phase voltage u a 、b phase voltage u b and c phase voltage u c , and the phase voltage u a 、u b 、u c Transformed to the two-phase stationary αβ coordinate system, the stator voltage u is obtained α 、u β ;

[0042] Voltage u α 、u β The calculation method is:

[0043]

[0044] Among them, u α is the α-axis component of the three-phase stator voltage of the permanent magnet motor in the stationary αβ coordinate system, u β It is the β-axis component of the stator three-phase voltage of the permanent magnet motor in the stationary αβ coordinate system.

[0045] Step 2: Use current sensors to collect the three-phase current of the permanent magnet motor, including the a-phase current i a , b-phase current i b and c-phase current i c , and the current i a 、i b 、i c Transformed to the two-phase stationary αβ coordinate system, the current i is obtainedα 、i β ;

[0046] Current i α 、i β The calculation method is:

[0047]

[0048] Among them, i α is the α-axis component of the three-phase current of the permanent magnet motor in the stationary αβ coordinate system, i β It is the β-axis component of the three-phase current of the permanent magnet motor in the stationary αβ coordinate system.

[0049] Step 3: The stator voltage u obtained in step 1 α 、u β And the current i obtained in step 2 α 、i β , construct the following sliding mode observer, satisfying:

[0050]

[0051] in, represents the α-axis component of the estimated current, represents the β-axis component of the estimated current, represents the α-axis component of the estimated back EMF, represents the β-axis component of the estimated back electromotive force, k represents the sliding mode gain, sgn() is the sign function, L s Represents the stator inductance of the permanent magnet motor, R s Represents the stator resistance of a permanent magnet motor.

[0052] Step 4: Convert the back EMF estimated in step 3 After passing through a low-pass filter to remove the sliding mode noise, the filtered back electromotive force is obtained. satisfy:

[0053]

[0054] Where s represents the Laplace operator, ω c Indicates the cutoff frequency of the low-pass filter.

[0055] Step 5: Filtered back electromotive force obtained in step 4 Calculate the back EMF error e q ,satisfy:

[0056]

[0057] Among them, e q is the back EMF error, is the estimated rotor position angle. When step 5 is executed for the first time, is zero, starting from the second execution of step 5, Obtained from step seven.

[0058] Step 6: Substitute the back EMF error e obtained in step 5 q After the proportional integral regulator, the estimated speed is obtained.

[0059] satisfy:

[0060]

[0061] Among them, k p is the proportional coefficient of the proportional-integral controller, k i is the integral coefficient of the proportional-integral controller, and s represents the Laplace operator.

[0062] Step 7: The estimated speed obtained in step 6 After the integration link, the estimated rotor position angle is obtained satisfy:

[0063]

[0064] Step 8: Repeat steps 5 to 7 until the back EMF error e q When it converges to zero, the estimated speed of the permanent magnet motor is obtained and rotor position angle Will and As the feedback of permanent magnet motor vector control, the speed sensorless control of permanent magnet motor can be realized. and The estimation method can be used to obtain the estimated value, which can be substituted into the permanent magnet motor vector control system to realize the dual closed-loop speed sensorless control of the speed outer loop and the current inner loop.

[0065] In order to verify the effectiveness of the present invention, an experimental verification is carried out. The motor parameters used in the experiment are shown in Table 1.

[0066] Table 1 Permanent magnet motor parameters

[0067]

[0068] During the experiment, the Figure 1 The conventional method shown and the method proposed in the present invention realize dual closed-loop vector control of the speed loop and the current loop of the permanent magnet motor. Figure 3 The experimental results of the conventional method are given when the load is 5Nm and the reference speed increases from 200r / min to 1000r / min. Figure 4The experimental results of the method proposed by the present invention under the same experimental conditions are given. Figure 3 and Figure 4 It can be seen that the rotor position angle estimation error (angle error for short) and speed estimation error (speed difference for short) of the method proposed in the present invention are significantly reduced. This is mainly because the method proposed in the present invention adds a low-pass filter to the phase-locked loop. This method can not only realize the adaptive compensation of the rotor position angle and overcome the influence of the speed estimation error on the rotor position angle estimation, but also can better filter out the sliding mode noise and obtain a smoother estimated speed and rotor position angle. The comparative experimental results reflect the superiority of the method proposed in the present invention.

[0069] The above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, improvements, etc. made within the spirit and principle of the present invention should be included in the protection scope of the present invention.

Claims

1. A speed sensorless control method for a permanent magnet motor, characterized in that: The steps are as follows: Step 1: Use the voltage sensor to collect the three-phase voltage of the permanent magnet motor stator, including the a-phase voltage u a 、b phase voltage u b and c phase voltage u c , and the phase voltage u a 、u b 、u c Transformed to the two-phase stationary αβ coordinate system, the stator voltage u is obtained α 、u β ; Step 2: Use current sensors to collect the three-phase current of the permanent magnet motor, including the a-phase current i a , b-phase current i b and c-phase current i c , and the current i a 、i b 、i c Transformed to the two-phase stationary αβ coordinate system, the current i is obtained α 、i β ; Step 3: The stator voltage u obtained in step 1 α 、u β And the current i obtained in step 2 α 、i β , construct the following sliding mode observer, satisfying: in, represents the α-axis component of the estimated current, represents the β-axis component of the estimated current, represents the α-axis component of the estimated back EMF, represents the β-axis component of the estimated back electromotive force, k represents the sliding mode gain, sgn() is the sign function, L s Represents the stator inductance of the permanent magnet motor, R s Represents the stator resistance of the permanent magnet motor; Step 4: Convert the back EMF estimated in step 3 After passing through a low-pass filter to remove the sliding mode noise, the filtered back electromotive force is obtained. Step 5: Filtered back electromotive force obtained in step 4 Calculate the back EMF error e q ; Step 6: Substitute the back EMF error e obtained in step 5 q After the proportional integral regulator, the estimated speed is obtained. satisfy: Among them, k p is the proportional coefficient of the proportional-integral controller, k i is the integral coefficient of the proportional-integral controller, s represents the Laplace operator; Step 7: The estimated speed obtained in step 6 After the integration link, the estimated rotor position angle is obtained satisfy: Step 8: Repeat steps 5 to 7 until the back EMF error e q When it converges to zero, the estimated speed of the permanent magnet motor is obtained and rotor position angle Will and As feedback of the permanent magnet motor vector control system, the speed sensorless control of the permanent magnet motor can be realized.

2. A permanent magnet motor speed sensorless control method according to claim 1, characterized in that: The voltage u α 、u β The calculation method is: Among them, u α is the α-axis component of the three-phase stator voltage of the permanent magnet motor in the stationary αβ coordinate system, u β It is the β-axis component of the stator three-phase voltage of the permanent magnet motor in the stationary αβ coordinate system.

3. A permanent magnet motor speed sensorless control method according to claim 1, characterized in that: The current i α 、i β The calculation method is: Among them, i α is the α-axis component of the three-phase current of the permanent magnet motor in the stationary αβ coordinate system, i β It is the β-axis component of the three-phase current of the permanent magnet motor in the stationary αβ coordinate system.

4. A permanent magnet motor speed sensorless control method according to claim 1, characterized in that: The filtered back EMF satisfy: Where s represents the Laplace operator, ω c Indicates the cutoff frequency of the low-pass filter.

5. A permanent magnet motor speed sensorless control method according to claim 4, characterized in that: The back EMF error e q satisfy: Among them, e q is the back EMF error, is the estimated rotor position angle. When step 5 is executed for the first time, is zero, starting from the second execution of step 5, Obtained from step seven.

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