Receiver baseband signal tracking processing method

Abstract

The invention discloses a tracking processing method for a baseband signal of a receiver, which comprises the following steps of: firstly, performing frequency correction through a frequency-locked loop, and performing phase correction through a phase-locked loop to realize accurate synchronization of carrier frequency and phase; then bit synchronization processing and incoherent integration are carried out, and the output signal-to-noise ratio of the loop and the measurement precision of the code phase are improved; and finally, the local pseudo-random code generator generates a local pseudo-random code, then the local pseudo-random code generator outputs three paths of pseudo-random codes which are mutually delayed by half a chip, namely a lead code, a real-time code and a lagging code, correlation values of the three paths of pseudo-random codes are obtained, a code phase error is obtained by calculating the correlation value difference of the lead code and the lagging code, and the local pseudo-random code is obtained. And after the code phase error is processed by a loop filter, an adjustment amount of a digital control oscillator is generated, so that the phase of the local pseudo-random code is adjusted. According to the invention, continuous adjustment of the digital control oscillator of the local carrier and the pseudo-random code is realized, so that correct navigation data information is demodulated for positioning calculation.

Description

Technical Field

[0001] The invention relates to the technical field of satellite navigation, and in particular to a tracking and processing method for a receiver baseband signal. Background Art

[0002] The Global Satellite Navigation System (GNSS) is an air-based radio navigation and positioning system that can provide users with all-weather three-dimensional coordinates, speed and time information at any location on the Earth's surface or in near-Earth space. It includes one or more satellite constellations and their augmentation systems required to support specific tasks. The four major satellite navigation system suppliers announced by the International Committee on Global Navigation Satellite Systems include the United States' Global Positioning System GPS, Russia's GLONASS, the European Union's Galileo Satellite Navigation System (GALILEO) and China's BeiDou Satellite Navigation System (BDS).

[0003] China's BeiDou-3 global satellite navigation system (hereinafter referred to as BeiDou-3 system) consists of 24 medium earth orbit satellites (MEO), 3 geostationary orbit satellites (GEO) and 3 inclined geosynchronous orbit satellites (IGSO), a total of 30 satellites. BeiDou-3 system broadcasts a new frequency signal B1C on MEO satellites and IGSO satellites. The B1C signal uses binary offset carrier (BOC, Binary Offset Carrier) modulation technology and is divided into a data transmission component and a pilot component. Because there is no modulation broadcast ephemeris on the pilot component, long-term coherent integration can be performed, thereby improving the sensitivity of the receiver. At the same time, the pilot component uses a layered code structure, so it has good cross-correlation characteristics. The above characteristics of the B1C signal have a significant impact on the improvement of the receiver's positioning performance and anti-interference performance.

[0004] Baseband algorithms are the core technology in digital communication systems, which are used to process and optimize low-frequency baseband signals containing original information. These algorithms are designed to improve signal transmission efficiency, reduce distortion, enhance anti-interference capabilities, and optimize spectrum utilization. Common baseband algorithms include pulse shaping, equalization, time synchronization, coding and modulation techniques. They are widely used in mobile communications, satellite communications, digital television, and computer networks, and play a vital role in the performance and reliability of modern digital communication systems. By applying these algorithms, communication systems can process and transmit information more efficiently, thereby meeting the growing demand for high-speed, high-quality communications.

[0005] The receiver obtains the rough value of the carrier Doppler frequency shift and pseudo-random code phase through the capture operation. The frequency error is within a few hundred Hz, and the pseudo-random code phase error is within half a chip range. The tracking loop needs to further precisely adjust the carrier and pseudo-random code phases to achieve loop lock state before data demodulation can be performed. Due to the Doppler frequency shift caused by the relative motion of the satellite and the receiver, the carrier and pseudo-random code phases in the signal received by the receiver are constantly changing, and the drift and random jitter of the local clock will also introduce changes. Therefore, the digital controlled oscillator NCO of the local carrier and the digital controlled oscillator NCO of the pseudo-random code must be continuously adjusted to achieve stable signal tracking, otherwise, it will quickly lose lock.

[0006] The purpose of signal tracking is to make the local carrier frequency and phase consistent with the carrier in the received signal, and to make the local pseudo-random code consistent with the phase of the pseudo-random code in the signal, so that the error is kept within a very small range and the lock state is achieved. When the receiver is locked, the correct navigation data information can be demodulated for positioning calculation. Summary of the invention

[0007] The technical problem to be solved by the present invention is to provide a tracking and processing method for the baseband signal of a receiver, which realizes continuous adjustment of the digital controlled oscillator of the local carrier and the pseudo-random code through processing in the carrier synchronization stage, the bit synchronization stage and the pseudo-random code tracking stage, thereby demodulating the correct navigation data information for positioning calculation.

[0008] The technical solution of the present invention is:

[0009] A tracking and processing method for a receiver baseband signal specifically comprises the following steps:

[0010] (1) Carrier synchronization stage: frequency correction is performed through the frequency-locked loop and phase correction is performed through the phase-locked loop to achieve accurate synchronization of carrier frequency and phase;

[0011] (2) Bit synchronization stage: bit synchronization processing and incoherent integration are performed to improve the output signal-to-noise ratio of the loop and the measurement accuracy of the code phase;

[0012] (3) Pseudo-random code tracking stage: The local pseudo-random code generator generates a local pseudo-random code, and then outputs three pseudo-random codes that are delayed by half a code chip: an advance code, an immediate code, and a lag code, and obtains the correlation values ​​of the three pseudo-random codes. The code phase error is obtained by calculating the correlation value difference between the advance code and the lag code. After the code phase error is processed by the loop filter, the adjustment amount of the digital control oscillator is generated, thereby adjusting the phase of the local pseudo-random code.

[0013] The specific steps of the frequency lock loop for frequency correction are: a local carrier generator generates a local carrier signal, mixes it with an intermediate frequency signal received by a receiver, obtains an in-phase branch signal IP and an orthogonal branch signal QP, and uses a local pseudo-random code generated by a local pseudo-random code generator to despread the signal, integrates and clears the despread signal, and then sends it to a phase detector for error detection to obtain a carrier frequency error, and the carrier frequency error signal is processed by a loop filter to obtain an adjustment amount of a digitally controlled oscillator, thereby adjusting the frequency of the local carrier.

[0014] The specific steps of the phase correction of the phase-locked loop are as follows: the frequency-locked loop outputs a carrier frequency value, integrates the carrier frequency value to obtain a carrier phase value, uses the carrier phase value as the initial value of the carrier phase of the phase-locked loop, calculates the carrier phase error through a phase detector, and processes the signal of the carrier phase error through a loop filter to generate an adjustment amount of a digitally controlled oscillator, thereby adjusting the phase of the local carrier.

[0015] The loop filter of the phase-locked loop is a second-order loop filter. The transfer function F(s) of the second-order loop filter is shown in the following formula (1):

[0016]

[0017] In formula (1), τ1 represents the first time constant of the loop filter, which is associated with the first-stage integrator of the loop filter and is used to control the dynamic response speed of the loop; τ2 represents the second time constant of the loop filter, which is associated with the second-stage integrator of the loop filter and is used to control the stability and noise suppression capability of the loop; s represents the complex frequency domain variable in the Laplace transform and is used to describe the frequency response characteristics of the system;

[0018] The system function H(s) of the phase-locked loop is shown in the following equation (2):

[0019]

[0020] In formula (2), K represents the total loop gain, ζ is the damping coefficient, which is calculated by the following formula (3); ω n is the characteristic frequency of the phase-locked loop system, which is calculated by the following formula (4):

[0021]

[0022] The noise bandwidth B corresponding to the second-order loop filter n The calculation formula is shown in the following formula (5):

[0023]

[0024] The system function H(s) of the phase-locked loop in equation (2) is simplified to the following equation (6):

[0025]

[0026] In formula (6), a2 represents a parameter related to the damping coefficient ζ in the second-order loop filter, a2=2ζ;

[0027] A simple frequency jump, a ramp-up of the phase, or a phase change of the excitation signal according to the first power of time represents a frequency step excitation. The frequency step excitation signal at time zero is defined as θ i (n) = Δ θ u(n), the corresponding z-domain transformation is as follows (7):

[0028]

[0029] In formula (3), Δω is the frequency crossing value;

[0030] The transformation formula from s domain to z domain is as follows (8):

[0031]

[0032] In formula (8), t s represents the sampling period, and substitute equation (8) into equation (1) to obtain the Laplace transform F(z) in the Z domain, which is specifically the following equation (9):

[0033]

[0034] In formula (9), C1 represents the first-order coefficient of the loop filter, which is used to control the transient response of the loop filter. C1 determines the loop filter's ability to respond quickly to changes in the input signal. C2 represents the second-order coefficient of the loop filter, which is used to control the steady-state response of the loop filter. C2 determines the loop filter's ability to suppress high-frequency noise and the stability of the system.

[0035] The transfer function N(z) of the digital controlled oscillator NCO is as follows (10):

[0036]

[0037] In formula (10), θ f (z) represents the Z-domain representation of the output phase of the digitally controlled oscillator NCO; V o (z) represents the Z-domain representation of the phase control voltage output by the digitally controlled oscillator NCO; K o stands for loop gain, which is a proportional factor used to control the rate of change of the output frequency of the digitally controlled oscillator NCO;

[0038] Therefore, the system function H(z) in the Z domain corresponding to the phase-locked loop is as follows (11):

[0039]

[0040] In formula (11), K d Represents the gain of the phase detector, which is the proportional relationship between the control voltage output by the phase detector and the phase difference of the input signal;

[0041] Then, a bilinear transformation is performed to transform equation (2) from the S domain to the Z domain, and the following equation (12) is obtained:

[0042]

[0043] In formula (12), γ represents the time constant ratio of the loop filter;

[0044] By equating equation (11) with equation (12), we can get the expressions of C1 and C2 as follows (13):

[0045]

[0046] During the bit synchronization processing, the edge of the bit jump is found by detecting the integral output of the in-phase branch to complete the bit synchronization processing; after the bit synchronization processing, 10 1ms non-coherent integrations are performed from the bit jump moment through non-coherent integration, and the loop update frequency is reduced to 100Hz.

[0047] The specific processing steps of the pseudo-random code tracking stage are:

[0048] S31, generating a local pseudo-random code by a pseudo-random code generator, and using a set pseudo-random code sequence and a phase selector to generate a required code sequence to achieve synchronization with a received satellite signal;

[0049] S32, the pseudo-random code generator outputs three pseudo-random codes that are delayed by half a chip, which are called advanced code, prompt code, and delayed code respectively. The code phase is expressed as the following formula (14):

[0050]

[0051] In formula (14), Indicates the phase of the instant code, t c It represents the time width corresponding to 1 chip, and C represents the carrier-to-noise ratio;

[0052] The signal after carrier stripping received by the receiver is expressed as the following equation (15):

[0053] x(t)=AC(t-τ)+n(t) (15);

[0054] In formula (15), A represents the signal amplitude, C(t-τ) represents the pseudo-random code phase in the signal, n(t) represents the noise, and represents the phase difference between the instantaneous code and the pseudo-random code in the signal; then the relationship between Δτ and the integrator output corresponding to the leading code and the lagging code is as follows (16):

[0055]

[0056] In formula (16), T represents the integration time, R E (Δτ), R P (Δτ) and R L (Δτ) are the integrator outputs corresponding to the advanced code, prompt code and delayed code respectively;

[0057] Calculate R E (Δτ)-R L (Δτ) at -t c <Δτ <t c The correlation value difference between the two is used to obtain the S curve of the delay locked loop, and the straight line crossing the zero point is within ±0.5t c The pseudo code phase discriminator outputs a linear relationship with the pseudo code phase difference. The error signal is fed back to the pseudo code generator through the loop filter, gradually making R E (Δτ)-R L (Δτ) tends to zero, achieving tracking and locking of the pseudo-random code;

[0058] S33. Then, in order to improve the signal-to-noise ratio of the detected signal, especially in a weak signal environment, the receiver further performs non-coherent integration:

[0059] First, define the coherent integration time of the receiver instantaneous branch as T coh , the coherent integration result is I p (n) and Q p (n), and then calculate the corresponding autocorrelation amplitude P(n). The incoherent integral P is the sum of N nc The autocorrelation amplitudes P(n) are accumulated, as shown in the following formula (17):

[0060]

[0061] Incoherent integration gain G nc The calculation formula is as follows (18):

[0062] G nc =10lg N nc -L SQ (18);

[0063] In formula (18), L SQ is the square loss;

[0064] Define the coherent integration result I P and Q PThe noise in is n I and n Q , so the relevant signal power V 2 It is the following formula (19):

[0065]

[0066] In formula (19), n I and n Q is Gaussian noise with zero mean and equal variance, so (I p +n1) 2 is the autocorrelation power of the detected signal.

[0067] Advantages of the present invention:

[0068] The present invention uses a frequency locked loop (FLL) to quickly narrow the carrier frequency error to a smaller range, so that the carrier frequency error is reduced to a level that can be switched to a phase locked loop (PLL) mode. The phase locked loop (PLL) mode further reduces the carrier phase error and realizes accurate carrier synchronization. Before tracking the pseudo random code, the present invention improves the measurement accuracy of the code phase by performing bit synchronization processing and non-coherent integration. The present invention calculates the difference between the leading code correlation value and the lagging code correlation value to calculate the code phase error. The code phase error is processed by a loop filter to generate an adjustment amount of a digital control oscillator, which is used to adjust the phase of the local pseudo random code to smooth the error signal and avoid instability caused by excessive adjustment.

[0069] The present invention is a tracking algorithm with fast convergence, high precision, strong robustness and high resource utilization. The algorithm has high precision and can improve the measurement accuracy of carrier phase and code phase while ensuring fast response by dynamically adjusting the parameters of the loop filter and the incoherent integration. The algorithm can be widely used in various scenarios such as high-precision positioning, low-power applications, high dynamic environments and complex electromagnetic environments, and provides important technical support for the baseband processing of Beidou navigation receivers. BRIEF DESCRIPTION OF THE DRAWINGS

[0070] Figure 1 It is the structure diagram corresponding to the Laplace transform.

[0071] Figure 2 This is the basic structure diagram of the pseudo-code tracking loop.

[0072] Figure 3 This is the relationship diagram of the integrator output of the leading code, the prompt code, and the lagging code in three cases.

[0073] Figure 4 It is the overall design diagram of the present invention. DETAILED DESCRIPTION

[0074] The following will be combined with the drawings in the embodiments of the present invention to clearly and completely describe the technical solutions in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0075] See Figure 4 , a tracking and processing method for a receiver baseband signal, specifically comprising the following steps:

[0076] (1) Carrier synchronization stage: frequency correction is performed through the frequency-locked loop and phase correction is performed through the phase-locked loop to achieve accurate synchronization of carrier frequency and phase. The specific steps are as follows:

[0077] S11, the local carrier generator generates a local carrier signal, mixes it with the intermediate frequency signal received by the receiver, obtains the in-phase branch signal IP and the orthogonal branch signal QP, and despreads it with the local pseudo-random code generated by the local pseudo-random code generator, integrates and clears the despread signal, and then sends it to the phase detector for error detection to obtain the carrier frequency error, and the signal of the carrier frequency error is processed by the loop filter to obtain the adjustment amount of the digital control oscillator, thereby adjusting the frequency of the local carrier;

[0078] S12, the frequency-locked loop outputs a carrier frequency value, integrates the carrier frequency value to obtain a carrier phase value, and uses the carrier phase value as the initial value of the carrier phase of the phase-locked loop. The carrier phase error is calculated by the phase detector, and the signal of the carrier phase error is processed by the loop filter to generate an adjustment amount of the digital control oscillator, thereby adjusting the phase of the local carrier;

[0079] Among them, the loop filter of the phase-locked loop selects a second-order loop filter, and the transfer function F(s) of the second-order loop filter is shown in the following formula (1):

[0080]

[0081] In formula (1), τ1 represents the first time constant of the loop filter, which is associated with the first-stage integrator of the loop filter and is used to control the dynamic response speed of the loop; τ2 represents the second time constant of the loop filter, which is associated with the second-stage integrator of the loop filter and is used to control the stability and noise suppression capability of the loop; s represents the complex frequency domain variable in the Laplace transform and is used to describe the frequency response characteristics of the system;

[0082] The system function H(s) of the phase-locked loop is shown in the following equation (2):

[0083]

[0084] In formula (2), K represents the total loop gain, ζ is the damping coefficient, which is calculated by the following formula (3); ω n is the characteristic frequency of the phase-locked loop system, which is calculated by the following formula (4):

[0085]

[0086] The noise bandwidth B corresponding to the second-order loop filter n The calculation formula is shown in the following formula (5):

[0087]

[0088] The system function H(s) of the phase-locked loop in equation (2) is simplified to the following equation (6):

[0089]

[0090] In formula (6), a2 represents a parameter related to the damping coefficient ζ in the second-order loop filter, a2=2ζ;

[0091] A simple frequency jump, a ramp-up of the phase, or a phase change of the excitation signal according to the first power of time represents a frequency step excitation. The frequency step excitation signal at time zero is defined as θ i (n) = Δ θ u(n), the corresponding z-domain transformation is as follows (7):

[0092]

[0093] In formula (3), Δω is the frequency crossing value;

[0094] The transformation formula from s domain to z domain is as follows (8):

[0095]

[0096] In formula (8), t s represents the sampling period, and substitute equation (8) into equation (1) to obtain the Laplace transform F(z) in the Z domain, which is specifically the following equation (9):

[0097]

[0098] In formula (9), C1 represents the first-order coefficient of the loop filter, which is used to control the transient response of the loop filter. C1 determines the loop filter's ability to respond quickly to changes in the input signal. C2 represents the second-order coefficient of the loop filter, which is used to control the steady-state response of the loop filter. C2 determines the loop filter's ability to suppress high-frequency noise and the stability of the system.

[0099] Figure 1 It is the structure diagram of Laplace transform;

[0100] The transfer function N(z) of the digital controlled oscillator NCO is as follows (10):

[0101]

[0102] In formula (10), θ f (z) represents the Z-domain representation of the output phase of the digitally controlled oscillator NCO; V o (z) represents the Z-domain representation of the phase control voltage output by the digitally controlled oscillator NCO; K o stands for loop gain, which is a proportional factor used to control the rate of change of the output frequency of the digitally controlled oscillator NCO;

[0103] Therefore, the system function H(z) in the Z domain corresponding to the phase-locked loop is as follows (11):

[0104]

[0105] In formula (11), K d Represents the gain of the phase detector, which is the proportional relationship between the control voltage output by the phase detector and the phase difference of the input signal;

[0106] Then, a bilinear transformation is performed to transform equation (2) from the S domain to the Z domain, and the following equation (12) is obtained:

[0107]

[0108] In formula (12), γ represents the time constant ratio of the loop filter;

[0109] By equating equation (11) with equation (12), we can get the expressions of C1 and C2 as follows (13):

[0110]

[0111] (2) Bit synchronization stage: Bit synchronization processing and incoherent integration are performed to improve the output signal-to-noise ratio of the loop and the measurement accuracy of the code phase. During the bit synchronization processing, the edge of the bit jump is found by detecting the integral output of the in-phase branch to complete the bit synchronization processing. After the bit synchronization processing, 10 incoherent integrations of 1 ms are performed starting from the bit jump moment, and the loop update frequency is reduced to 100 Hz.

[0112] (3) Pseudo-random code tracking stage: The local pseudo-random code generator generates a local pseudo-random code, and then outputs three pseudo-random codes that are delayed by half a chip: an advance code, an immediate code, and a lag code, and obtains the correlation values ​​of the three pseudo-random codes. The code phase error is obtained by calculating the correlation value difference between the advance code and the lag code. After the code phase error is processed by a loop filter, an adjustment amount of a digitally controlled oscillator is generated, thereby adjusting the phase of the local pseudo-random code.

[0113] In order to ensure that the phase of the pseudo-random code reproduced locally is consistent with that in the received signal, and to verify that the receiver can demodulate correct and stable data information from the tracking loop, the pseudo-random code tracking loop is implemented through a delay locked loop (DLL); the pseudo-random code tracking loop first copies the pseudo-code of a satellite signal with a certain phase, and performs correlation operations with the received signal, extracts the code phase error, and feeds it back to the pseudo-code generation controller to adjust the frequency and code phase of the pseudo-code in real time, and then repeats the above process until the copied pseudo-code is consistent with the pseudo-code of the received signal.

[0114] Figure 2 It is the basic structure of the pseudo-code tracking loop, which specifically includes the following processing steps:

[0115] S31, generating a local pseudo-random code by a pseudo-random code generator, and using a set pseudo-random code sequence and a phase selector to generate a required code sequence to achieve synchronization with a received satellite signal;

[0116] S32, the pseudo-random code generator outputs three pseudo-random codes that are delayed by half a chip, which are called advanced code, prompt code, and delayed code respectively. The code phase is expressed as the following formula (14):

[0117]

[0118] In formula (14), Indicates the phase of the instant code, t c It represents the time width corresponding to 1 chip, and C represents the carrier-to-noise ratio;

[0119] The signal after carrier stripping received by the receiver is expressed as the following equation (15):

[0120] x(t)=AC(t-τ)+n(t) (15);

[0121] In formula (15), A represents the signal amplitude, C(t-τ) represents the pseudo-random code phase in the signal, n(t) represents the noise, and represents the phase difference between the instantaneous code and the pseudo-random code in the signal; then the relationship between Δτ and the integrator output corresponding to the leading code and the lagging code is as follows (16):

[0122]

[0123] In formula (16), T represents the integration time, R E (Δτ), R P (Δτ) and R L (Δτ) are the integrator outputs corresponding to the advanced code, prompt code and delayed code respectively;

[0124] Let the time t corresponding to a chip be c , so when Δτ=+0.5t c , Δτ=0, Δτ=+0.5t c The integral values ​​are obtained in three cases, such as Figure 3 shown.

[0125] Calculate R E (Δτ)-R L (Δτ) at -t c <Δτ <t c The correlation value difference between the two is used to obtain the S curve of the delay locked loop, and the straight line crossing the zero point is within ±0.5t c The pseudo code phase discriminator outputs a linear relationship with the pseudo code phase difference. The error signal is fed back to the pseudo code generator through the loop filter, gradually making R E (Δτ)-R L (Δτ) tends to zero, achieving tracking and locking of the pseudo-random code;

[0126] S33. Then, in order to improve the signal-to-noise ratio of the detected signal, especially in a weak signal environment, the receiver further performs non-coherent integration:

[0127] First, define the coherent integration time of the receiver instantaneous branch as T coh , the coherent integration result is I p (n) and Q p (n), and then calculate the corresponding autocorrelation amplitude P(n). The incoherent integral P is the sum of N nc The autocorrelation amplitudes P(n) are accumulated, as shown in the following formula (17):

[0128]

[0129] Incoherent integration gain G nc The calculation formula is as follows (18):

[0130] G nc =10lg N nc -L SQ (18);

[0131] In formula (18), L SQ is the square loss;

[0132] Define the coherent integration result I P and Q P The noise in is n I and n Q , so the relevant signal power V 2 It is the following formula (19):

[0133]

[0134] In formula (19), n I and n Q is Gaussian noise with zero mean and equal variance, so (I p +n1) 2 is the autocorrelation power of the detected signal.

[0135] Then, the carrier frequency and the pseudo-random code frequency are compared. The accuracy of the carrier phase measurement is obviously better than the measurement value of the pseudo-random code phase. However, due to the integer ambiguity in the carrier phase measurement, the measurement value has ambiguity, while the pseudo-random code phase measurement value has no ambiguity.

[0136] The carrier loop phase measurement value is several orders of magnitude more accurate than the pseudo-random code tracking loop measurement value. The Doppler measurement value from the carrier loop is highly accurate and can promptly reflect the relative motion of the receiver on the line connecting to the satellite. Therefore, if the carrier velocity information is used to assist the pseudo-random code tracking loop and control the frequency of the local reproducing pseudo-code, the dynamic stress on the pseudo-random code tracking loop can be basically completely eliminated. Under this condition, the receiver is allowed to use a smaller loop bandwidth B L , reducing loop noise and improving the measurement accuracy of code phase. Under the carrier assistance, the pseudo-random code tracking loop does not need a high-order loop, and a second-order loop filter can be used. Because the Doppler frequency shift is proportional to the signal transmission frequency, the carrier loop assistance frequency factor K o It can be expressed as:

[0137]

[0138] f corry Represents the carrier frequency, which is the center frequency of the carrier part of the received signal. It is used to modulate and demodulate the signal and is the basis for the receiver to perform carrier synchronization and demodulation. code Represents the pseudo-random code frequency, which indicates the frequency of the pseudo-random code part in the received signal. It is used to generate a local pseudo-random code so as to perform correlation operations with the received signal to achieve code phase tracking and synchronization.

[0139] The carrier auxiliary amount plus the phase error output by the pseudo-random code tracking loop is used as the adjustment amount of the NCO to adjust the code rate of the pseudo-code.

[0140] Although embodiments of the present invention have been shown and described, it will be appreciated by those skilled in the art that various changes, modifications, substitutions and variations may be made to the embodiments without departing from the principles and spirit of the present invention, and that the scope of the present invention is defined by the appended claims and their equivalents.

Claims

1. A tracking and processing method for a receiver baseband signal, characterized in that: The specific steps include: (1) Carrier synchronization stage: frequency correction is performed through the frequency-locked loop and phase correction is performed through the phase-locked loop to achieve accurate synchronization of carrier frequency and phase; (2) Bit synchronization stage: bit synchronization processing and incoherent integration are performed to improve the output signal-to-noise ratio of the loop and the measurement accuracy of the code phase; (3) Pseudo-random code tracking stage: The local pseudo-random code generator generates a local pseudo-random code, and then outputs three pseudo-random codes that are delayed by half a code chip: an advance code, an immediate code, and a lag code, and obtains the correlation values ​​of the three pseudo-random codes. The code phase error is obtained by calculating the correlation value difference between the advance code and the lag code. After the code phase error is processed by the loop filter, the adjustment amount of the digital control oscillator is generated, thereby adjusting the phase of the local pseudo-random code.

2. The tracking and processing method of a receiver baseband signal according to claim 1, characterized in that: The specific steps of the frequency lock loop for frequency correction are: a local carrier generator generates a local carrier signal, mixes it with an intermediate frequency signal received by a receiver, obtains an in-phase branch signal IP and an orthogonal branch signal QP, and uses a local pseudo-random code generated by a local pseudo-random code generator to despread the signal, integrates and clears the despread signal, and then sends it to a phase detector for error detection to obtain a carrier frequency error, and the carrier frequency error signal is processed by a loop filter to obtain an adjustment amount of a digitally controlled oscillator, thereby adjusting the frequency of the local carrier.

3. The tracking and processing method of a receiver baseband signal according to claim 2, characterized in that: The specific steps of the phase correction of the phase-locked loop are as follows: the frequency-locked loop outputs a carrier frequency value, integrates the carrier frequency value to obtain a carrier phase value, uses the carrier phase value as the initial value of the carrier phase of the phase-locked loop, calculates the carrier phase error through a phase detector, and processes the signal of the carrier phase error through a loop filter to generate an adjustment amount of a digitally controlled oscillator, thereby adjusting the phase of the local carrier.

4. The tracking and processing method of a receiver baseband signal according to claim 3, characterized in that: The loop filter of the phase-locked loop is a second-order loop filter. The transfer function F(s) of the second-order loop filter is shown in the following formula (1): In formula (1), τ1 represents the first time constant of the loop filter, which is associated with the first-stage integrator of the loop filter and is used to control the dynamic response speed of the loop; τ2 represents the second time constant of the loop filter, which is associated with the second-stage integrator of the loop filter and is used to control the stability and noise suppression capability of the loop; s represents the complex frequency domain variable in the Laplace transform and is used to describe the frequency response characteristics of the system; The system function H(s) of the phase-locked loop is shown in the following equation (2): In formula (2), K represents the total loop gain, ζ is the damping coefficient, which is calculated by the following formula (3); ω n is the characteristic frequency of the phase-locked loop system, which is calculated by the following formula (4): The noise bandwidth B corresponding to the second-order loop filter n The calculation formula is shown in the following formula (5): The system function H(s) of the phase-locked loop in equation (2) is simplified to the following equation (6): In formula (6), a2 represents a parameter related to the damping coefficient ζ in the second-order loop filter, a2=2ζ; A simple frequency jump, a ramp-up of the phase, or a phase change of the excitation signal according to the first power of time represents a frequency step excitation. The frequency step excitation signal at time zero is defined as θ i (n) = Δ θ u(n), the corresponding z-domain transformation is as follows (7): In formula (3), Δω is the frequency crossing value; The transformation formula from s domain to z domain is as follows (8): In formula (8), t s represents the sampling period, and substitute equation (8) into equation (1) to obtain the Laplace transform F(z) in the Z domain, which is specifically the following equation (9): In formula (9), C1 represents the first-order coefficient of the loop filter, which is used to control the transient response of the loop filter. C1 determines the loop filter's ability to respond quickly to changes in the input signal. C2 represents the second-order coefficient of the loop filter, which is used to control the steady-state response of the loop filter. C2 determines the loop filter's ability to suppress high-frequency noise and the stability of the system. The transfer function N(z) of the digital controlled oscillator NCO is as follows (10): In formula (10), θ f (z) represents the Z-domain representation of the output phase of the digitally controlled oscillator NCO; V o (z) represents the Z-domain representation of the phase control voltage output by the digitally controlled oscillator NCO; K o stands for loop gain, which is a proportional factor used to control the rate of change of the output frequency of the digitally controlled oscillator NCO; Therefore, the system function H(z) in the Z domain corresponding to the phase-locked loop is as follows (11): In formula (11), K d Represents the gain of the phase detector, which is the proportional relationship between the control voltage output by the phase detector and the phase difference of the input signal; Then, a bilinear transformation is performed to transform equation (2) from the S domain to the Z domain, and the following equation (12) is obtained: In formula (12), γ represents the time constant ratio of the loop filter; By equating equation (11) with equation (12), we can get the expressions of C1 and C2 as follows (13):

5. The tracking and processing method of a receiver baseband signal according to claim 3, characterized in that: During the bit synchronization processing, the edge of the bit jump is found by detecting the integral output of the in-phase branch to complete the bit synchronization processing; after the bit synchronization processing, 10 1ms non-coherent integrations are performed from the bit jump moment through non-coherent integration, and the loop update frequency is reduced to 100Hz.

6. The tracking and processing method of a receiver baseband signal according to claim 5, characterized in that: The specific processing steps of the pseudo-random code tracking stage are: S31, generating a local pseudo-random code by a pseudo-random code generator, and using a set pseudo-random code sequence and a phase selector to generate a required code sequence to achieve synchronization with a received satellite signal; S32, the pseudo-random code generator outputs three pseudo-random codes that are delayed by half a chip, which are called advanced code, prompt code, and delayed code respectively. The code phase is expressed as the following formula (14): In formula (14), Indicates the phase of the instant code, t c It represents the time width corresponding to 1 chip, and C represents the carrier-to-noise ratio; The signal after carrier stripping received by the receiver is expressed as the following equation (15): x(t)=AC(t-τ)+n(t) (15); In formula (15), A represents the signal amplitude, C(t-τ) represents the pseudo-random code phase in the signal, n(t) represents the noise, and represents the phase difference between the instantaneous code and the pseudo-random code in the signal; then the relationship between Δτ and the integrator output corresponding to the leading code and the lagging code is as follows (16): In formula (16), T represents the integration time, R E (Δτ), R P (Δτ) and R L (Δτ) are the integrator outputs corresponding to the advanced code, prompt code and delayed code respectively; Calculate R E (Δτ)-R L (Δτ) at -t c <Δτ <t c The correlation value difference between the two is used to obtain the S curve of the delay locked loop, and the straight line crossing the zero point is within ±0.5t c The pseudo code phase discriminator outputs a linear relationship with the pseudo code phase difference. The error signal is fed back to the pseudo code generator through the loop filter, gradually making R E (Δτ)-R L (Δτ) tends to be close to zero, achieving tracking and locking of the pseudo-random code; S33. Then, in order to improve the signal-to-noise ratio of the detected signal, especially in a weak signal environment, the receiver further performs non-coherent integration: First, define the coherent integration time of the receiver instantaneous branch as T co , the coherent integration result is I p (n) and Q p (n), and then calculate the corresponding autocorrelation amplitude P(n). The incoherent integral P is the sum of N nc The autocorrelation amplitudes P(n) are accumulated, as shown in the following formula (17): Incoherent integration gain G nc The calculation formula is as follows (18): G nc =10lg N nc -L SQ (18); In formula (18), L SQ is the square loss; Define the coherent integration result I P and W P The noise in is n I and n Q , so the relevant signal power V 2 It is the following formula (19): In formula (19), n I and n Q is Gaussian noise with zero mean and equal variance, so (I p +n1) 2 is the autocorrelation power of the detected signal.

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