LLC resonant converter and control method thereof

Abstract

The embodiment of the invention provides an LLC resonant converter and a control method thereof, and the LLC resonant converter comprises a transformer which comprises a primary side coil and a secondary side coil which are coupled with each other, the primary side coil is connected with an input circuit through a switch unit, the input circuit is used for inputting voltage, the secondary side coil is connected with an output circuit, and the output circuit is used for outputting voltage; the sampling circuit is used for sampling the output voltage to obtain a sampling voltage; a driving unit for driving the switching unit; the controller is used for calculating a basic switching frequency according to the sampling voltage; calculating the slope of the sampling voltage; calculating a final switching frequency according to the basic switching frequency and the slope; and generating a driving signal according to the final switching frequency, and sending the driving signal to a driving unit. According to the LLC resonant converter provided by the embodiment of the invention, the response speed to load change can be improved, the output voltage overshoot / undershoot amplitude during load transient change is reduced, the time for recovering a circuit to a steady state is shortened, and the dynamic response performance is improved.

Description

TECHNICAL FIELD

[0001] The present application relates to the technical field of power supply, in particular to an LLC resonant converter and a control method thereof. BACKGROUND

[0002] LLC resonant converter is a kind of isolated DC-DC converter topology with high efficiency and high power density, which is widely used in high-power scenarios such as server power supply, charger, photovoltaic inverter, etc. However, its dynamic response (especially load transient response) is relatively slow, because the traditional voltage mode control (VMC) or current mode control (CMC) mainly depends on the feedback loop, and the loop bandwidth is limited by stability, which is usually much lower than the switching frequency, resulting in delay and overshoot / undershoot in response to load step change.

[0003] Based on this, the prior art proposes some solutions, but there are still deficiencies, for example, the following solutions: Solution one: increase the loop bandwidth. Due to the limitation of stability, its effect is limited, and it may cause the noise sensitivity to increase.

[0004] Solution two: output voltage feedforward, that is, directly adding the output voltage Vout or its error signal to the control quantity (duty ratio or frequency). However, when the load changes rapidly, the Vout feedforward compensation is not timely and sufficient.

[0005] Solution three: load current feedforward. This solution needs to add an additional current sensor, which increases the cost, complexity, and power consumption, and the current detection has precision and delay problems.

[0006] In summary, there is an urgent need for a control method that is relatively simple in structure, low in cost, does not increase additional sensors, and can significantly improve the dynamic response (especially the load transient response) speed of the LLC converter. SUMMARY

[0007] The embodiments of the present application provide an LLC resonant converter and a control method thereof, which can improve the response speed of the LLC resonant converter to load changes, significantly reduce the overshoot / undershoot amplitude of the output voltage during load transient, shorten the time for the circuit to recover to steady state, and improve the dynamic response performance.

[0008] The embodiments of the present application provide an LLC resonant converter, which comprises: A transformer comprising a primary coil and a secondary coil coupled to each other, the primary coil being connected to an input circuit through a switching unit, the input circuit being used for input voltage, the secondary coil being connected to an output circuit, the output circuit being used for output voltage; a sampling circuit, connected with the output circuit, for sampling the output voltage to obtain a sampling voltage; a driving unit, connected with the switching unit, for driving the switching unit; a controller, connected with the sampling circuit and the driving unit, the controller being configured to: calculate a base switching frequency according to the sampling voltage; calculate a slope of the sampling voltage; calculate a final switching frequency according to the base switching frequency and the slope; generate a driving signal according to the final switching frequency and send the driving signal to the driving unit.

[0009] In some embodiments, the controller comprises: an analog-to-digital converter, connected with the sampling circuit, for converting the sampling voltage into a digital voltage; a closed-loop controller, connected with the analog-to-digital converter, for calculating the base switching frequency according to the digital voltage; a calculation unit, connected with the sampling circuit and the closed-loop controller, for calculating the slope according to the digital voltage and calculating the final switching frequency according to the base switching frequency and the slope.

[0010] In some embodiments, the calculation unit calculates the final switching frequency according to the following formula: Fs_cmd=Fs_loop+K_ff*(dVout / dt) wherein Fs_cmd is the final switching frequency, Fs_loop is the base switching frequency, K_ff is a preset gain coefficient, and dVout / dt is the slope.

[0011] In some embodiments, the closed-loop controller is a voltage-mode controller, and the gain coefficient K_ff>0; or the closed-loop controller is a current-mode controller, and the gain coefficient K_ff<0.

[0012] In some embodiments, the absolute value of the gain coefficient K_ff ranges from 2<|K_ff|<4.

[0013] In some embodiments, if the calculated K_ff*(dVout / dt)>10k, it is set as K_ff*(dVout / dt)=10k; if the calculated K_ff*(dVout / dt)<-10k, it is set as K_ff*(dVout / dt)=-10k.

[0014] In some embodiments, the controller further comprises a driving module connected with the computing unit and the driving unit, configured to generate a PWM driving signal according to the final switching frequency and send the PWM driving signal to the driving unit.

[0015] The embodiments of the present application further provide a control method of an LLC resonant converter, applied to the LLC resonant converter of any of the above embodiments, and the control method comprises: calculating a basic switching frequency according to the sampling voltage; calculating a slope of the sampling voltage; calculating a final switching frequency according to the basic switching frequency and the slope; generating a driving signal according to the final switching frequency and sending the driving signal to the driving unit.

[0016] In some embodiments, when the final switching frequency is calculated according to the basic switching frequency and the slope, the calculation is performed according to the following formula: Fs_cmd=Fs_loop+K_ff*(dVout / dt) wherein Fs_cmd is the final switching frequency, Fs_loop is the basic switching frequency, K_ff is a preset gain coefficient, and dVout / dt is the slope.

[0017] In some embodiments, if K_ff*(dVout / dt) calculated is greater than 10k, it is set as K_ff*(dVout / dt)=10k; if K_ff*(dVout / dt) calculated is less than -10k, it is set as K_ff*(dVout / dt)=-10k.

[0018] The LLC resonant converter 100 of the embodiments of the present application introduces the instantaneous change rate (i.e. the slope dVout / dt) of the output voltage Vout as an independent feedforward signal into the control loop, and the signal directly participates in the calculation of the final switching frequency Fs_cmd, which can predict the change trend of the load in advance and make corresponding compensation, so as to adjust the output voltage. Compared with the feedback regulation of the input power of the primary coil through the change of the output voltage, the embodiments of the present application directly use the slope dVout / dt of the output voltage Vout as a feedforward signal to regulate the input power of the primary coil, so as to improve the response speed of the LLC resonant converter to the load change, significantly reduce the overshoot / undershoot amplitude of the output voltage during load transient, shorten the time for the circuit to recover to the steady state, and improve the dynamic response performance. BRIEF DESCRIPTION OF DRAWINGS

[0019] In order to more clearly illustrate the technical solutions in the embodiments of the present application, the drawings needed to be used in the embodiments will be briefly introduced. Obviously, the drawings in the following description are only some embodiments of the present application, and other drawings can be obtained by those skilled in the art without creative effort on the basis of these drawings.

[0020] Figure 1 The figure is a circuit structure schematic diagram of the LLC resonant converter 100 in the embodiment of the present application.

[0021] Figure 2 The figure is a circuit structure schematic diagram of the LLC resonant converter 100 in the embodiment of the present application.

[0022] Figure 3 The figure is a flowchart of the control method of the LLC resonant converter in the embodiment of the present application. DETAILED DESCRIPTION

[0023] The technical solutions in the embodiments of the present application will be described clearly and completely in the following with reference to the drawings in the embodiments of the present application. Obviously, the described embodiments are only some embodiments of the present application, not all the embodiments. Based on the embodiments in the present application, all the other embodiments obtained by those skilled in the art without creative effort belong to the scope of protection of the present application.

[0024] The embodiment of the present application provides an LLC resonant converter, which can be applied to the power supply of high-power equipment such as server power supply, charger, photovoltaic inverter, etc.

[0025] Reference Figure 1 , Figure 1 The figure is a circuit structure schematic diagram of the LLC resonant converter 100 in the embodiment of the present application. The LLC resonant converter 100 includes a transformer T, an input circuit 10, a switching unit 20, an output circuit 30, a sampling circuit 40, a controller 50 and a driving unit 60.

[0026] The transformer T includes a primary coil Na and a secondary coil Ns coupled to each other. The primary coil Na is connected with the input circuit 10 through the switching unit 20, and the secondary coil Ns is connected with the output circuit 30. The input circuit 10 is used for input voltage, for example, input 220V AC voltage. The output circuit 30 is used for output voltage, for example, output 12V DC voltage.

[0027] Switching unit 20 is used to connect or disconnect the input circuit 10 and the primary coil Na. When switching unit 20 is on, the input circuit 10 is connected to the primary coil Na, and the input circuit 10 inputs power to the primary coil Na. When switching unit 20 is off, the input circuit 10 is disconnected from the primary coil Na, and the input circuit 10 stops inputting power to the primary coil Na. In practical applications, switching unit 20 can be a transistor such as a MOSFET, IGBT (Insulated Gate Bipolar Transistor), or a bipolar transistor. Switching unit 20 is connected to driving unit 60, which is used to drive switching unit 20 to turn on or off.

[0028] The sampling circuit 40 is connected to the output circuit 30. The sampling circuit 40 is used to sample the output voltage of the output circuit 30 to obtain the sampled voltage.

[0029] The controller 50 is connected to the sampling circuit 40 and the drive unit 60. The controller 50 is used to: calculate the basic switching frequency based on the sampling voltage of the sampling circuit 40; calculate the slope of the sampling voltage; calculate the final switching frequency based on the basic switching frequency and the slope; generate a drive signal based on the final switching frequency, and send the drive signal to the drive unit 60.

[0030] Understandably, the slope of the sampling voltage is consistent with the slope of the output voltage of the output circuit 30. If the slope is positive, it indicates that the sampling voltage is increasing, and the output voltage is also increasing; the larger the slope, the faster the sampling voltage increases, and the faster the output voltage increases. If the slope is negative, it indicates that the sampling voltage is decreasing, and the output voltage is also decreasing; the larger the absolute value of the slope, the faster the sampling voltage decreases, and the faster the output voltage decreases.

[0031] Understandably, the sampled voltage (or output voltage) is based on a gradually changing slope, for example, a gradually increasing positive slope or a gradually decreasing negative slope. Therefore, when the output voltage changes, the slope can reflect the trend of the output voltage change more quickly (or earlier), and the change in output voltage has a certain lag compared to the slope parameter.

[0032] In this embodiment, the controller 50 calculates the slope of the sampled voltage. The slope allows for a faster (earlier) determination of the output voltage's changing trend. The final switching frequency calculated based on the slope enables the switching unit 20 to adjust the input power of the primary coil more quickly (earlier), thereby adjusting the output voltage. Compared to adjusting the input power of the primary coil through feedback from changes in the output voltage, this embodiment uses the slope as a feedforward signal to directly adjust the input power of the primary coil. Therefore, it improves the LLC resonant converter's response speed to load changes, significantly reduces the output voltage overshoot / undershoot during load transients, shortens the time for the circuit to recover to steady state, and enhances dynamic response performance.

[0033] In some embodiments, reference Figure 2 , Figure 2 This is an example circuit structure diagram of an LLC resonant converter 100 according to an embodiment of this application. The input voltage is represented by Vin, and the output voltage is represented by Vout.

[0034] In some embodiments, the switching unit 20 includes MOSFET Q1 and MOSFET Q2. The gates of both MOSFET Q1 and MOSFET Q2 are connected to the driving unit 60.

[0035] In some embodiments, the LLC resonant converter 100 further includes an input filter circuit. For example, in one example, the input filter circuit includes capacitors C1, C2, and C3, and inductors L1 and L2. The drain of MOSFET Q1 and one end of capacitor C1 are connected to one end of a power supply, for example, to the positive terminal of the power supply. The source of MOSFET Q2 and one end of capacitor C2 are connected to the other end of the power supply, for example, to the negative terminal of the power supply. The source of MOSFET Q1, the drain of MOSFET Q2, the other end of capacitor C1, and the other end of capacitor C2 are connected to one end of inductor L1, and the other end of inductor L1 is connected to one end of the primary winding Na. One end of capacitor C3 is connected to the negative terminal of the power supply, and the other end is connected to the other end of the primary winding Na. One end of inductor L2 is connected to one end of the primary winding Na, and the other end is connected to the other end of the primary winding Na.

[0036] In some embodiments, the LLC resonant converter 100 further includes an output rectifier circuit and an output filter circuit. For example, in one example, the output rectifier circuit includes diodes D1 and D2, and the output filter circuit includes capacitor C4 and resistor R1. The cathode of diode D1 is connected to one end of the secondary coil Ns, and the cathode of diode D2 is connected to the other end of the secondary coil Ns. The anodes of diodes D1 and D2 are connected together and connected to one end of capacitor C4 and one end of resistor R1. The other ends of capacitor C4 and resistor R1 are connected together to the center tap of the secondary coil Ns, which is used for the output voltage Vout.

[0037] In some embodiments, the sampling circuit 40 includes a scaling-down module 41 and a sampling module 42. The scaling-down module 41 is connected to the output circuit, for example, to the center tap of the secondary coil Ns. The scaling-down module 41 is used to scale down the output voltage Vout to obtain a scaled-down voltage. The sampling module 42 is connected to the scaling-down module 41 and the controller 50. The sampling module 42 is used to sample the scaled-down voltage to obtain a sampled voltage.

[0038] In some embodiments, the controller 50 may be a digital control chip (DSP). The controller 50 includes an analog-to-digital converter 51, a closed-loop controller 52, and a computing unit 53.

[0039] The analog-to-digital converter 51 can be represented as an ADC. The analog-to-digital converter 51 is connected to the sampling circuit 40, for example, to the sampling module 42, and is used to convert the sampled voltage obtained by the sampling module 42 into a digital voltage.

[0040] The closed-loop controller 52 can be represented as a 2P2Z, which is a digital loop compensator with two poles and two zeros, ensuring steady-state accuracy and anti-interference capability. The closed-loop controller 52 is connected to the analog-to-digital converter 51. The closed-loop controller 52 is used to calculate the aforementioned basic switching frequency based on the digital voltage converted by the sampling module 42, which can be represented as Fs_loop.

[0041] The calculation unit 53 is connected to the sampling circuit 51 and the closed-loop controller 52. The calculation unit 53 is used to calculate the slope mentioned above based on the digital voltage converted by the sampling module 42, and the slope can be expressed as dVout / dt. The calculation unit 53 is also used to calculate the final switching frequency mentioned above based on the basic switching frequency Fs_loop and the slope dVout / dt, and the final switching frequency can be expressed as Fs_cmd.

[0042] In some embodiments, the calculation unit 53 calculates the final switching frequency according to the following formula: Fs_cmd=Fs_loop+K_ff*(dVout / dt) Where Fs_cmd is the final switching frequency, Fs_loop is the basic switching frequency, K_ff is the preset gain coefficient, and dVout / dt is the slope.

[0043] Understandably, the slope dVout / dt is the derivative of the output voltage Vout with respect to time t; a positive value indicates an increase, and a negative value indicates a decrease. The slope dVout / dt can be calculated using a digital differential method: dVout[k] = (Vout[k] - Vout[k-1]) / Ts. Here, Ts is the sampling period. In one example, the sampling frequency can be set to 200kHz, corresponding to a sampling period Ts of 5µs. Vout[k] is the sampled voltage of the current period, and Vout[k-1] is the sampled voltage of the previous period.

[0044] In practical applications, the slope dVout / dt can be calculated using a low-pass filter to suppress high-frequency noise and prevent noise amplification that could cause frequency jitter. For example, a first-order IIR filter can be added: dVout_filt[k] = α*dVout[k] + (1-α)*dVout_filt[k-1], where α can be adjusted based on the balance between dynamic response speed and stability. With the addition of a low-pass filter, the formula for calculating the final switching frequency Fs_cmd can be adjusted to: Fs_cmd = Fs_loop + K_ff*dVout_filt[k]. In practical applications, the bandwidth of the low-pass filter needs to be balanced between noise suppression and response speed.

[0045] In practical applications, the gain coefficient K_ff is a pre-set fixed value, and the specific value can be determined through experimental debugging. In some embodiments, the gain coefficient K_ff can be set according to the control mode of the closed-loop controller 52. For example, if the closed-loop controller 52 is a voltage-mode controller (VMC), the gain coefficient K_ff is set to > 0; or, if the closed-loop controller 52 is a current-mode controller (CMC), the gain coefficient K_ff is set to < 0.

[0046] In some embodiments, the absolute value of the gain coefficient K_ff ranges from 2 < |K_ff| < 4. For example, in one example, assuming the final switching frequency Fs_cmd is in the range of 60kHz to 200kHz, then K_ff can be set to 1.4285e -5 *Fs_cmd+1.14286, where e is the natural constant, with a value of approximately 2.71828.

[0047] In practical applications, to prevent excessive frequency deviation in the calculated final switching frequency Fs_cmd under extreme conditions, which could exceed the safe range or cause system instability, K_ff*(dVout / dt) can be limited. For example, in some embodiments, K_ff*(dVout / dt) can be limited to within ±10k, that is: if the calculated K_ff*(dVout / dt) > 10k, then K_ff*(dVout / dt) = 10k; if the calculated K_ff*(dVout / dt) < -10k, then K_ff*(dVout / dt) = -10k.

[0048] Accordingly, the final switching frequency Fs_cmd is limited to the range between the minimum frequency Fmin (to prevent entering the capacitive region) and the maximum value Fmax, i.e., Fs_cmd = clamp(Fs_cmd, Fmin, Fmax). The minimum frequency Fmin corresponds to the minimum value of Fs_cmd calculated when K_ff*(dVout / dt) = -10k (since the base switching frequency Fs_loop is dynamic, the Fs_cmd calculated when K_ff*(dVout / dt) = -10k also has a range), and the maximum frequency Fmax corresponds to the maximum value of Fs_cmd calculated when K_ff*(dVout / dt) = 10k (since the base switching frequency Fs_loop is dynamic, the Fs_cmd calculated when K_ff*(dVout / dt) = 10k also has a range).

[0049] The following description uses a closed-loop controller 52 as a voltage-mode controller (VMC) as an example to illustrate the application scenario of this application embodiment. When the closed-loop controller 52 is a voltage-mode controller (VMC), the gain coefficient K_ff > 0.

[0050] When the load suddenly increases, the output voltage Vout begins to drop rapidly, and the slope dVout / dt of the output voltage is negative and has a large absolute value. Since the gain coefficient K_ff>0, the calculated final switching frequency Fs_cmd is less than the basic switching frequency Fs_loop, meaning Fs_cmd is significantly reduced. Therefore, the switching frequencies of MOSFETs Q1 and Q2 immediately decrease, increasing the gain of the LLC resonant converter, enhancing the converter's output power capability, and providing the required current to the load more quickly, thereby effectively suppressing the magnitude and duration of the drop in output voltage Vout. At the same time, the closed-loop controller 52 also responds to the drop in Vout by starting to reduce the basic switching frequency Fs_loop, but the slope (dVout / dt) feedforward action is faster and more direct because the slope (dVout / dt) occurs before the error of Vout is significantly established; it senses the rate of change of the output voltage Vout.

[0051] When the load suddenly decreases, the output voltage Vout begins to rise rapidly, and the slope dVout / dt of the output voltage is positive and has a large absolute value. Since the gain coefficient K_ff>0, the calculated final switching frequency Fs_cmd is greater than the basic switching frequency Fs_loop, meaning Fs_cmd increases significantly. Therefore, the switching frequencies of MOSFETs Q1 and Q2 immediately increase, reducing the gain of the LLC resonant converter, weakening the converter's output power capability, and reducing the current supplied to the load, thereby effectively suppressing the rise of the output voltage Vout. At the same time, the closed-loop controller 52 also responds to the rise of Vout by increasing the basic switching frequency Fs_loop, but similarly, the slope (dVout / dt) feedforward action is faster and more direct.

[0052] In some embodiments, continue to refer to Figure 2 The controller 50 also includes a drive module 54. The drive module 54 can be a PWM drive module. The drive module 54 is connected to the calculation unit 53 and the drive unit 60. The drive module 54 is used to generate a PWM drive signal based on the final switching frequency Fs_cmd calculated by the calculation unit 53, and send the PWM drive signal to the drive unit 60, so that the drive unit 60 drives the MOSFETs Q1 and Q2.

[0053] The LLC resonant converter 100 of this application introduces the instantaneous rate of change of the output voltage Vout (i.e., the slope dVout / dt) as an independent feedforward signal into the control loop. This signal directly participates in the calculation of the final switching frequency Fs_cmd, enabling it to predict the load change trend in advance and perform corresponding compensation, thereby adjusting the output voltage. Compared to adjusting the input power of the primary coil by feedback through changes in the output voltage, this application uses the slope dVout / dt of the output voltage Vout as a feedforward signal to directly adjust the input power of the primary coil. Therefore, it can improve the response speed of the LLC resonant converter to load changes, significantly reduce the output voltage overshoot / undershoot amplitude during load transients, shorten the time for the circuit to recover to steady state, and improve dynamic response performance.

[0054] This application also provides a control method for an LLC resonant converter, applicable to the LLC resonant converter in any of the above embodiments. (Reference) Figure 3 , Figure 3 This is a flowchart illustrating the control method for an LLC resonant converter according to an embodiment of this application. The control method includes the following steps: 210. The basic switching frequency is calculated based on the sampled voltage. 220, calculate the slope of the sampled voltage; 230, the final switching frequency is calculated based on the basic switching frequency and slope; 240. Generate a drive signal based on the final switching frequency and send the drive signal to the drive unit.

[0055] In some embodiments, when calculating the final switching frequency based on the basic switching frequency and the slope in step 230, the calculation is performed according to the following formula: Fs_cmd=Fs_loop+K_ff*(dVout / dt) Where Fs_cmd is the final switching frequency, Fs_loop is the basic switching frequency, K_ff is the preset gain coefficient, and dVout / dt is the slope.

[0056] In some embodiments, if the calculated K_ff*(dVout / dt) > 10k, then K_ff*(dVout / dt) = 10k is set; if the calculated K_ff*(dVout / dt) < -10k, then K_ff*(dVout / dt) = -10k is set.

[0057] The specific implementation methods of each step of the above control method can be found in the descriptions of the various embodiments of the LLC resonant converter 100, and will not be repeated here.

[0058] It should be noted that in the embodiments of this application, "connection" can be understood as electrical connection. The connection between two electrical components can be a direct or indirect connection between the two electrical components. For example, the connection between A and B can be a direct connection between A and B, or an indirect connection between A and B through one or more other electrical components.

[0059] The LLC resonant converter and its control method provided in the embodiments of this application have been described in detail above. Specific examples have been used to illustrate the principles and implementation methods of this application. The descriptions of the embodiments above are only for the purpose of helping to understand this application. Furthermore, those skilled in the art will recognize that, based on the ideas of this application, there will be changes in the specific implementation methods and application scope. Therefore, the content of this specification should not be construed as a limitation of this application.

Claims

1. An LLC resonant converter, characterized in that, include: A transformer includes a primary winding and a secondary winding coupled to each other. The primary winding is connected to an input circuit via a switching unit. The input circuit is used to input voltage. The secondary winding is connected to an output circuit, which is used to output voltage. A sampling circuit, connected to the output circuit, is used to sample the output voltage to obtain a sampling voltage. A driving unit, connected to the switching unit, is used to drive the switching unit; The controller, connected to the sampling circuit and the driving unit, is used for: The basic switching frequency is calculated based on the sampled voltage. Calculate the slope of the sampled voltage; The final switching frequency is calculated based on the basic switching frequency and the slope. A drive signal is generated based on the final switching frequency, and the drive signal is sent to the drive unit.

2. The LLC resonant converter according to claim 1, characterized in that, The controller includes: An analog-to-digital converter, connected to the sampling circuit, is used to convert the sampled voltage into a digital voltage; A closed-loop controller, connected to the analog-to-digital converter, is used to calculate the basic switching frequency based on the digital voltage; The calculation unit, connected to the sampling circuit and the closed-loop controller, is used to calculate the slope based on the digital voltage and to calculate the final switching frequency based on the basic switching frequency and the slope.

3. The LLC resonant converter according to claim 2, characterized in that, The calculation unit calculates the final switching frequency according to the following formula: Fs_cmd=Fs_loop+K_ff*(dVout / dt) Where Fs_cmd is the final switching frequency, Fs_loop is the basic switching frequency, K_ff is the preset gain coefficient, and dVout / dt is the slope.

4. The LLC resonant converter according to claim 3, characterized in that: The closed-loop controller is a voltage-mode controller, and the gain coefficient K_ff > 0; or The closed-loop controller is a current-mode controller, and the gain coefficient K_ff < 0.

5. The LLC resonant converter according to claim 4, characterized in that, The absolute value of the gain coefficient K_ff is in the range of 2 < |K_ff| < 4.

6. The LLC resonant converter according to claim 5, characterized in that: If the calculated K_ff*(dVout / dt) > 10k, then set it to K_ff*(dVout / dt) = 10k; If the calculated K_ff*(dVout / dt) < -10k, then set it to K_ff*(dVout / dt) = -10k.

7. The LLC resonant converter according to claim 2, characterized in that, The controller further includes a drive module, which is connected to the computing unit and the drive unit, and is used to generate a PWM drive signal according to the final switching frequency and send the PWM drive signal to the drive unit.

8. A control method for an LLC resonant converter, characterized in that, The control method, applied to the LLC resonant converter according to any one of claims 1 to 7, comprises: The basic switching frequency is calculated based on the sampled voltage. Calculate the slope of the sampled voltage; The final switching frequency is calculated based on the basic switching frequency and the slope. A drive signal is generated based on the final switching frequency, and the drive signal is sent to the drive unit.

9. The control method according to claim 8, characterized in that, When calculating the final switching frequency based on the basic switching frequency and the slope, the calculation is performed according to the following formula: Fs_cmd=Fs_loop+K_ff*(dVout / dt) Where Fs_cmd is the final switching frequency, Fs_loop is the basic switching frequency, K_ff is the preset gain coefficient, and dVout / dt is the slope.

10. The control method according to claim 9, characterized in that: If the calculated K_ff*(dVout / dt) > 10k, then set it to K_ff*(dVout / dt) = 10k; If the calculated K_ff*(dVout / dt) < -10k, then set it to K_ff*(dVout / dt) = -10k.

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