Active Clamp Flyback Converter

Abstract

The present invention relates to controlling of an active clamp fly-back converter comprising a control unit 2 that turns on / off a clamp switch two times by two pulses during off-period of a main switch. The control unit 2 sets the on-period of the first pulse to half the resonant frequency of a clamp capacitor and a leakage inductor to the resonant frequency, inclusive, turns on the clamp switch when current does not flow through the body diode of the clamp switch and flows the resonant current of the clamp capacitor and the leakage inductor, flows current through the body diode after turning off the clamp switch, turns on the clamp switch after the magnetization current of a magnetizing inductor becomes zero by the second pulse and flows negative direction current through the magnetizing inductor, and performs zero-voltage switching of the main switch QL, thereby providing a high-efficiency and low noise converter.

Description

[Technical field]

[0001] The present invention relates to an active clamp flyback converter. [Background technology]

[0002] An active clamp flyback converter is described in, for example, US Pat. No. 6,399,361.

[0003] In this active clamp flyback converter, a series circuit consisting of a transformer primary winding and main switch is connected to a DC power supply, a series circuit consisting of a clamp switch and clamp capacitor is connected to both ends of the transformer primary winding, and a series circuit of a rectifier element and output capacitor is connected to the transformer secondary winding.To make it easier to explain the current path that flows through the leakage inductance and excitation inductance, the transformer is described as an equivalent circuit in which the excitation inductance of the primary and secondary windings is LM, the leakage inductance LK is an ideal transformer with NP:NS, a series circuit of LK and LM is connected between the primary windings, Np of the ideal transformer is connected to LM, and Ns of the ideal transformer is connected between the secondary windings.

[0004] The main switch switches on and off while adjusting its on-time based on the output voltage of the output capacitor. When the main switch is on, a positive current flows through the excitation inductance, storing positive magnetic energy in the transformer core, and during the off-time, the magnetic energy stored in the transformer core is output as a current from the secondary winding to the output capacitor, transmitting power from the primary side to the secondary side of the transformer.

[0005] However, when the main switch is ON, magnetic energy is also stored in the leakage inductance. After the main switch is turned OFF, this magnetic energy differs from the magnetic energy stored in the magnetizing inductance in that the magnetic energy is stored in the direction that reverse biases the secondary diode, and therefore cannot be output directly to the secondary side, so the magnetic energy stored in the leakage inductance is temporarily transferred to the clamp capacitor.

[0006] In Patent Document 1, the first ON of the clamp switch is set after the main switch is turned OFF, and is set to the period t1 to t2 during which the magnetic energy stored in the leakage inductance immediately after the main switch is turned OFF flows as current to the clamp capacitor via the clamp switch.

[0007] The first ON of the clamp switch does not need to be turned ON if the main switch is an element with a body diode such as a MOS-FET, but by making the ON resistance of the clamp switch an element with a sufficiently small value, the resistance loss due to the ON resistance will be sufficiently smaller than the forward voltage drop loss of the body diode of the clamp switch. Therefore, by setting the first ON, the loss in the clamp switch is reduced, and a highly efficient power supply can be configured.

[0008] The second ON of the clamp switch is turned on at time t3, when the voltage across the clamp switch resonates and becomes zero again after all the magnetic energy stored in the excitation inductance while the main switch is ON has been discharged to the secondary side. The ON period of the second ON of the clamp switch is set to flow the excitation current in the negative direction, and this ON period is set so that the energy stored in the excitation inductance is equal to or greater than the energy stored in the parasitic capacitance and stray capacitance equivalently connected to both ends of the main switch.

[0009] When the second on of the clamp switch ends and the clamp switch turns off, the energy stored in the excitation inductance discharges the capacitance equivalently connected across the main switch, causing the voltage across the main switch to drop to zero volts.

[0010] After the voltage of the main switch becomes zero volts, the excitation current flows through the body diode of the main switch. By turning on the main switch while the current is flowing through this body diode, a zero-volt, zero-current switch is realized and the switching loss of the main switch is greatly reduced, making it possible to configure a power supply that is more efficient and has less noise than conventional flyback converters. [Prior art documents] [Patent documents]

[0011] [Patent Document 1] China No. 112510976 Summary of the Invention [Problem to be solved by the invention]

[0012] However, in Patent Document 1, the rectifying element on the secondary side is made synchronously rectified, making it difficult to achieve further high efficiency.

[0013] In the control of the clamp switch in Patent Document 1, operation waveforms including the current waveforms of each element are shown in FIGS. 18 and 19, and the problems involved will be described.

[0014] In the control of the clamp switch in Patent Document 1, the first ON is set to the period during which the energy stored in the leakage inductance when the main switch is ON flows as a current through the path leakage inductance → clamp switch → clamp capacitor → leakage inductance after the main switch is turned OFF.

[0015] After the main switch is turned off, the current in the magnetizing inductance flows to the secondary side, generating a counter electromotive force N·Vo (N is the turns ratio of the primary to secondary windings, and Vo is the output voltage) in the primary winding Np. Therefore, when the first on of the clamp switch ends, the voltage of the clamp capacitor is charged with a voltage α that is larger than the counter electromotive force N·Vo by the energy stored in the leakage inductance, and becomes a voltage of N·Vo + α.

[0016] After that, all the energy in the excitation inductance has been discharged from the transformer's secondary winding to the secondary side, and after voltage resonance operation, the second ON of the clamp switch begins at zero volts. Because the voltage of the clamp capacitor remains N·Vo+α, a voltage of N·Vo+α is applied to the transformer's primary winding when the clamp switch is turned on for the second time, and a voltage of Vo+α / N is induced in the secondary winding, so if α / N is greater than the forward voltage Vf of the secondary-side diode, a resonant current with the resonant period Taclk of the clamp capacitor and leakage inductance flows in the following path: clamp capacitor → clamp switch → transformer primary winding → clamp capacitor.

[0017] The resonant current flowing in this primary winding flows as resonant current W2 through the secondary winding → diode → output capacitor → secondary winding.

[0018] At this time, a voltage of N · Vo is applied to the primary winding, so during the period Ton2 that the clamp switch is ON, a current ILM of N · Vo / Lm · Ton2 flows in the negative direction through the magnetizing inductance, and a current W1 flows through the clamp switch.

[0019] Therefore, after the main switch is turned OFF, a triangular wave current like that of a flyback converter flows through the secondary side diode, and then, when the clamp switch is turned ON for the second time, the resonant current W2 flows again.

[0020] Furthermore, when the on-time Ton2 of the second ON of the clamp switch is Taclk / 2 or more, the current in the leakage inductance becomes zero within the Ton2 period, but when Ton2 is Taclk / 2 or less, the resonant current W2 turns off quickly as shown by W2 in Figure 19.

[0021] Therefore, if the secondary side diode is of the synchronous rectification type, the following problems arise, making synchronous rectification difficult.

[0022] As shown in Figs. 18 and 19, current flows through the synchronous rectifier element twice per cycle, and therefore the synchronous rectifier element needs to be turned on and off twice per cycle, resulting in a doubling of the drive loss.

[0023] Since the di / dt at which the resonant current turns off is large, as shown by W2 in Figure 19, a high-performance synchronous rectification controller that can quickly control the turn-off of the synchronous rectification element is required, but it is difficult to realize such a high-performance synchronous rectification controller.

[0024] The present invention provides an active clamp flyback converter that facilitates synchronous rectification of the rectified current on the secondary side. [Means for solving the problem]

[0025] The active clamp flyback converter according to the present invention includes a first series circuit in which a first parallel circuit of a main switch (QL), a first diode (BDL), and a voltage resonant capacitor (Cv) and a primary winding (Np) are connected in series across a DC power supply (Vin), and a second series circuit in which a second parallel circuit of a clamp switch (QH) and a second diode (BDH) and a clamp capacitor (Cac) are connected in series across the primary winding (Np), and a transformer (T) having a secondary winding (Ns) electromagnetically coupled to the primary winding (Np), the transformer (T) is configured such that the primary winding (Np) has an excitation inductance (Lm) and a coupling coefficient between the primary winding and the secondary winding is less than 1 so that the primary winding (Np) has a leakage inductance (Llk), and a third series circuit in which a rectifier element (Ds) and an output capacitor (Co) are connected in series across the secondary winding (Ns).

[0026] The converter includes a control unit (2) that turns on and off the main switch (QL) and turns on and off the clamp switch (QH) twice during an off period of the main switch. The control unit (2) sets an on period of the first on of the clamp switch to Taclk / 2 to Taclk, and sets an on time of Tof, where Toff1 is a time when the excitation current of the transformer decreases to zero after the main switch (QL) is turned off and the energy stored in the leakage inductance (Llk) is transferred to a clamp capacitor, and Taclk is a resonance period between the clamp capacitor and the leakage inductance. The off time is set from f1-Taclk to Toff1-Taclk / 2, the off time is set from Toff1-Taclk / 2 to Toff1, and the energy stored in the leakage inductance (Llk) when the main switch is ON and transferred to the clamp capacitor (Cac) causes a resonant current to flow through the clamp capacitor (Cac) and leakage inductance at the first ON, discharges the clamp capacitor (Cac), and regeneratively outputs the energy of the clamp capacitor to the secondary side. When the resonant current flows in the reverse direction, the resonant current in the reverse direction is limited by the excitation current, thereby limiting the charging to the clamp capacitor (Cac).

[0027] By making the charge of the charge described above half or less of the charge of the discharge described above, no resonant current will flow even if the clamp switch is next turned on, and no resonant current will flow on the secondary side either.

[0028] The control unit (2) turns on the clamp switch (QH) at the second ON at the time when the excitation current is decreasing and becomes zero. Since the energy of the clamp capacitor is regeneratively output to the secondary side at the first ON, no resonant current flows at the second ON, and the excitation current of the excitation inductance flows in the negative direction, thereby reducing the turn-on loss of the main switch. [Brief description of the drawings]

[0029] [Figure 1] FIG. 1 is a configuration diagram of an active clamp flyback converter according to a first embodiment of the present invention. [Diagram 2] FIG. 2 is a diagram showing a first operation waveform of the active clamp flyback converter according to the first embodiment of the present invention. [Diagram 3] FIG. 3 is a diagram showing a second operation waveform of the active clamp flyback converter according to the first embodiment of the present invention. [Figure 4] FIG. 4 is a diagram showing a current path during a period T1 in the active clamp flyback converter according to the first embodiment of the present invention. [Diagram 5] FIG. 5 is a diagram showing a current path during the period T2 in the active clamp flyback converter according to the first embodiment of the present invention. [Figure 6] FIG. 6 is a diagram showing a current path during period T3 in the active clamp flyback converter according to the first embodiment of the present invention. [Figure 7] FIG. 7 is a diagram showing a current path during a period T4 in the active clamp flyback converter according to the first embodiment of the present invention. [Figure 8] FIG. 8 is a diagram showing a current path during a period T5 in the active clamp flyback converter according to the first embodiment of the present invention. [Figure 9]FIG. 9 is a diagram showing a current path during a period T6 in the active clamp flyback converter according to the first embodiment of the present invention. [Figure 10] FIG. 10 is a diagram showing a current path during a period T7 in the active clamp flyback converter according to the first embodiment of the present invention. [Figure 11] FIG. 11 is a diagram showing a current path during a period T8 in the active clamp flyback converter according to the first embodiment of the present invention. [Figure 12] FIG. 12 is a diagram showing a specific example of the active clamp flyback converter according to the first embodiment of the present invention. [Figure 13] FIG. 13 is a diagram showing a first operation waveform of the active clamp flyback converter shown in FIG. [Figure 14] FIG. 14 is a diagram showing a second operation waveform of the active clamp flyback converter shown in FIG. [Figure 15] FIG. 15 is a diagram showing operation waveforms of the on-timing detection unit 32 and the second on-signal unit in the specific example of the active clamp flyback converter according to the first embodiment of the present invention shown in FIG. [Figure 16] FIG. 16 is a configuration diagram of an active clamp flyback converter according to a second embodiment of the present invention. [Figure 17] FIG. 17 is a diagram showing operation waveforms of the active clamp flyback converter according to the second embodiment of the present invention. [Figure 18] FIG. 18 is a diagram showing operational waveforms of an active clamp flyback converter according to the prior art. [Figure 19] FIG. 19 is a diagram showing operational waveforms of an active clamp flyback converter according to a conventional technique, the operational waveforms having different timings from those in FIG. [Figure 20] FIG. 20 is a diagram showing a modified configuration of the active clamp flyback converter according to the embodiment of the present invention. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0030] Hereinafter, active clamp flyback converters according to several embodiments of the present invention will be described in detail with reference to the drawings. The same or corresponding parts in the drawings of the active clamp flyback converters according to the embodiments are denoted by the same reference numerals and their description will be omitted.

[0031] (First embodiment) 1 is a configuration diagram of an active clamp flyback converter according to a first embodiment of the present invention. The active clamp flyback converter according to the first embodiment includes a first series circuit in which a main switch QL and a primary winding Np of a transformer T are connected in series across a DC power supply Vin, and a second series circuit in which a clamp switch QH and a clamp capacitor Cac are connected in series across the primary winding Np.

[0032] The main switch QL and clamp switch QH are composed of MOSFETs (Metal Oxide Semiconductor Field Effect Transistors), each with a body diode BDL, BDH between the drain and source. The MOSFET has an output capacitance Coss between the drain and source, and this capacitance appears in the resonant operation. To simplify the explanation of the operation, in Fig. 1 it is connected as a voltage resonant capacitor (Cv) across the main switch QL, but it is not necessary to attach a capacitor other than the output capacitance Coss of the MOSFET.

[0033] The converter further includes a transformer T having a secondary winding Ns electromagnetically coupled to the primary winding Np. The transformer T is configured so that the primary winding Np has an excitation inductance Lm and a coupling coefficient between the primary winding Np and the secondary winding Ns is less than 1 so that the primary winding Np has a leakage inductance Llk. A third series circuit in which a rectifier element Ds and an output capacitor Co are connected in series is connected to both ends of the secondary winding Ns.

[0034] The converter includes an output voltage detection unit 1 that detects the output voltage of the output capacitor Co, and a control unit 2 that turns a main switch QL on and off based on the output voltage detected by the output voltage detection unit 1, and turns a clamp switch QH on and off twice with a first pulse and a second pulse during the off period of the main switch QL.

[0035] The control unit 2 is equipped with a first on signal that turns the main switch QL on and off, a second on signal (the first pulse) that turns the clamp switch QH on and off during the period in which the main switch QL is off by the first on signal, and a third on signal (the second pulse) that turns the clamp switch QH on and off again after the second on signal.

[0036] The control unit 2 sets the on timing of the second on signal so that the resonant current that reverses after 1 / 2 the resonant period of the resonant circuit of the lamp capacitor Cac and leakage inductance Llk formed when the clamp switch QH is turned on with the second on signal and flows in a direction to charge the clamp capacitor Cac is limited by the excitation current of the excitation inductance of the primary winding, sets the on period of the second on signal to be more than 1 / 2 the resonant period so that the clamp capacitor Cac is charged and discharged only once by the resonant current that discharges the clamp capacitor Cac when the clamp switch QH is turned on and the current that charges the clamp capacitor Cac is limited by the inverted resonant current and the excitation current, turns the clamp switch QH on and off with the second on signal, outputs a third on signal after the excitation current of the excitation inductance Lm becomes zero, and turns on the clamp switch QH with the third on signal to pass a negative current through the excitation inductance.

[0037] After the main switch QL is turned off and the energy stored in the leakage inductance Llk is transferred to the clamp capacitor Cac, the time when the excitation current of the transformer T decreases to zero is Toff1, and the resonance period of the clamp capacitor and leakage inductance Llk is Taclk.

[0038] In the present invention, the on-timing of the first pulse of the clamp switch QH is set so that the on-time period occurs when the current in the exciting inductance becomes sufficiently small compared to the resonant current, that is, before the current in the exciting inductance becomes zero.

[0039] If the on-period of the clamp switch QH is Taclk / 2 and the end time of the on-period is set to the time when the excitation current of the excitation inductance Lm becomes zero, then no negative excitation current will flow and only the positive resonant current will flow. Therefore, the voltage of the clamp capacitor Cac will be discharged to the voltage of N·Vo-α. Therefore, when the clamp switch QH is turned on for the second time, a voltage lower than the output voltage Vo will be applied to the secondary winding Ns, and no resonant current will be generated between the clamp capacitor Cac and the leakage inductance Llk.

[0040] Furthermore, if the voltage of the clamp switch QH is below N·Vo when it is turned on for the second time, no resonant current will flow when it is turned on for the second time. Therefore, if the positive charge of the resonant current that flows when it is turned on for the first time is Qp, and the negative charge Qn that is subsequently limited by the exciting inductance Lm is Qp / 2 ≧ Qn, no resonant current will flow when it is turned on for the second time.

[0041] Furthermore, if the resonance period of the clamp capacitor Cac and leakage inductance Llk is Taclk, the first on-period is longer than the period during which a positive current flows, and the off-timing may be while a negative current flows. Therefore, the period during which a negative current flows is Taclk / 2 if it is not limited by the excitation inductance Lm, and is a period of Taclk / 2 or more if the negative current is limited. Therefore, it is best to set at least the first on-period of the clamp switch QH to Taclk / 2 to Taclk, but it is sufficient to turn off the first pulse before the negative current ends.

[0042] In the present invention, the control unit 2 changes the on-time of the second on-signal from Toff1-Taclk to Toff1-Taclk / 2, sets the on-period to be equal to or longer than Taclk / 2, and sets the off-time to the time until the excitation current becomes zero.

[0043] When the main switch QL is ON, the energy is stored in the leakage inductance Llk. After the main switch QL is turned OFF, the energy transferred to the clamp capacitor Cac is turned ON by a second ON signal and regeneratively output to the secondary side.

[0044] Furthermore, after the excitation current decreases and becomes zero in response to the third ON signal, the control unit 2 turns on the clamp switch QH to cause the excitation current of the excitation inductance Lm to flow in the negative direction.

[0045] When the clamp switch QH is turned off by the third on signal, the energy of the voltage resonant capacitor Cv is regenerated to Vin by the negative excitation current of the excitation inductance Lm, thereby reducing the turn-on loss of the main switch QL.

[0046] Figure 20 is a modified example of the configuration diagram of the active clamp flyback converter according to the first embodiment of the present invention in Figure 1. In controlling the clamp switch of the present invention, it can be easily determined that the same effect can be obtained even if the second series circuit of the clamp switch QH and the clamp capacitor Cac is connected as shown in Figure 20, so this explanation will be given using the configuration diagram of Figure 1, but the scope of application of the present invention also applies to the active clamp flyback converter modified as in Figure 20.

[0047] FIG. 2 is a diagram showing a first operation waveform of the active clamp flyback converter according to the first embodiment of the present invention. In FIG. 2, LG is a gate signal of the main switch QL. HG is a gate signal of the clamp switch QH. ILM is an excitation current flowing through the excitation inductance Lm. Vds(L / S) is a voltage between the drain and source of the main switch QL. Ids(L / S) is a drain current of the main switch QL. Vds(H / S) is a voltage between the drain and source of the clamp switch QH. Ids(H / S) is a drain current of the clamp switch QH. Vrm is a voltage across the diode Ds. If is a current through the diode Ds. The resonance period of the clamp capacitor Cac and the leakage inductance Llk is Taclk.

[0048] As shown in FIG. 2, the first first pulse P1 is turned on after a time before Taclk / 2 from time t5 when the excitation current ILM of the transformer T becomes zero and before Taclk from time t5, and the on period is set between Taclk / 2 and Taclk so that the first pulse P1 is turned off before time t5 when the excitation current ILM of the transformer T becomes zero. Therefore, the first pulse P1 is turned off between time t5 and before Taclk / 2 from time t5. The second second pulse P2 is turned on when the excitation current becomes zero once, and the drain-source voltage Vds(H / S) of the clamp switch QH oscillates due to the resonant operation of the excitation inductance Lm and the voltage resonant capacitor Cv, the clamp switch QH is the bottom switch of the drain-source voltage Vds(H / S) of the clamp switch QH (the top of the drain-source voltage Vds(L / S) of the main switch QL), and the resonant current flowing in the excitation current ILM is decreasing and becomes zero.

[0049] Next, the operation of periods T1 to T8 of the first operation waveform shown in FIG. 2 will be described with reference to FIG. 4 to FIG. 11. First, the operation of period T1 will be described with reference to FIG. 4. In period T1, at time t0, the main switch QL is turned on, and a current flows through the path of Vin positive pole → Llk → Lm → QL → Vin negative pole. At this time, an excitation current ILM flows through the excitation inductance Lm of the transformer T, and the excitation current ILM increases linearly, and energy is stored in the excitation inductance Lm. Since the same current as the excitation current flowing through the excitation inductance Lm also flows through the leakage inductance Llk, energy is also stored in the leakage inductance Llk. At time t1, the main switch QL turns off.

[0050] Next, the operation of period T2 will be described with reference to Figure 5. In period T2, at time t1, the main switch QL turns off, and the energy stored in the leakage inductance Llk of the transformer T cannot flow to the secondary side. For this reason, the energy stored in the leakage inductance Llk flows as a current through the path Llk → Lm → BDH → Cac → Llk. In other words, the clamp capacitor Cac is charged through the body diode BDH of the clamp switch QH. At this time, because the voltage of the clamp capacitor Cac is equal to or lower than N·Vo, the energy stored in the excitation inductance Lm of the transformer T charges the clamp capacitor Cac.

[0051] Next, when the voltage of the clamp capacitor Cac reaches or exceeds N·Vo, the energy stored in the magnetizing inductance Lm of the transformer T begins to flow from Lm to Np, so that current begins to flow from the secondary winding of the transformer T through the path Ns → Ds → Co → Ns. At time t2, the energy stored in the leakage inductance Llk finishes discharging. At this point, the voltage of the clamp capacitor Cac becomes higher than N·Vo by the amount of the energy stored in the leakage inductance Llk, and becomes N·Vo+α, where N is the turns ratio of Np / Ns.

[0052] Next, the operation during period T3 will be described with reference to Figure 6. The operation during period T3 is the same as that of a normal flyback converter, with the energy stored in the magnetizing inductance Lm flowing from Lm to Np to Lm, and on the secondary side from Ns to Ds to Co to Ns, and current W2 flows through the diode current If. During period T3, at time t2, the energy stored in the leakage inductance Llk is fully charged in the clamp capacitor Cac, and the voltage of the clamp capacitor Cac remains at N·Vo+α during period T3. Also, the voltage across the primary winding Np is N·Vo if we ignore the forward voltage of the diode Ds, so the voltage of the main switch QL is Vin+N·Vo.

[0053] Next, the operation during period T4 will be described with reference to FIG.

[0054] Because time t3 is set before the excitation current ILM becomes zero, the excitation current ILM is discharging current to the secondary side. Therefore, the back electromotive force of the primary winding Np is N·Vo. At time t3, the clamp switch QH is turned on by the first pulse P1. The voltage of the clamp capacitor Cac is N·Vo+α, and a potential difference of α is applied to Llk.

[0055] Therefore, during period T4, the clamp capacitor Cac and leakage inductance Llk resonate, and a resonant current W3 flows from Cac to QH to Np to Llk to Cac. The excitation current ILM of the excitation inductance Lm flows from Lm to Np to Lm. Therefore, the excitation current ILM of the excitation inductance Lm and the resonant current flow in the same direction in Np, so a current W4, which is the excitation current of the excitation inductance Lm superimposed on the resonant current, flows from Ns to Ds to Co to Ns on the secondary winding Ns side. At time t4, the resonant current W3 becomes zero after 1 / 2 the resonant period Taclk has elapsed. At this time, the clamp capacitor Cac is resonating, so it becomes N·Vo-α.

[0056] Next, the operation during period T5 will be described with reference to FIG.

[0057] In period T5, at time t4, the resonant current goes from 0 to negative, and the resonant current flows in the opposite direction to period T4, Llk → Np → QH → Cac → Llk. The current in the excitation inductance Lm flows in the same direction as in period T4, Lm → Np → Lm. Since the resonant current and the excitation current ILM flow in the opposite direction, when the excitation current is greater than the resonant current, a current equivalent to the difference between the excitation current and the resonant current flows on the secondary side. When the resonant current becomes equal to the excitation current, a current limited by the excitation current ILM flows on the primary side, as shown in waveform W5, Llk → Lm → QH → Cac → Llk. Therefore, no current flows on the secondary side.

[0058] During this period T5, current flows from the source to the drain of the clamp switch QH, so if the clamp switch QH is turned off during this period, the current is commutated to the body diode BDH. Therefore, the operating mode does not change.

[0059] When the main switch QL was ON, the current value flowing through the leakage inductance Llk was equal to the peak value of the resonant current, so if the first pulse P1 is set as above, the charge of waveform W5 will be less than half the charge of W3. Therefore, the voltage of the clamp capacitor Cac will be less than N·Vo, and the difference voltage with respect to the N·Vo voltage is expressed as α'.

[0060] Next, the operation during period T6 will be described with reference to FIG.

[0061] In period T6, at time t5, the voltage of the clamp capacitor Cac becomes equal to or lower than N·Vo (N·Vo-α´), and the energy stored in the transformer is completely discharged.

[0062] The diode Ds becomes non-conductive due to the discharge caused by the resonant operation in the period T4, and both the clamp switch QH and the main switch QL are turned off.

[0063] Since the voltage of the voltage resonant capacitor Cv is Vin+N·Vo-α', there is a potential difference of N·Vo-α' between the voltage resonant capacitor Cv and the DC power supply Vin, Lm+Llk. As a result, the voltage resonant capacitor Cv and the inductance Lm+Llk of the transformer primary winding resonate in series, and current flows through the path Cv→Lm→Llk→Vin→Cv, causing the voltage of the voltage resonant capacitor Cv to drop. If the resonance period between the voltage resonant capacitor Cv and the inductance Lm+Llk of the transformer primary winding is Tcvlm, the voltage of the voltage resonant capacitor Cv drops to Vin-N·Vo+α' when Tcvlm / 2 has elapsed since time t5, and then a current flows in the opposite direction to the above path, and the voltage of the voltage resonant capacitor Cv becomes Vin+N·Vo-α' again when Tcvlm has elapsed since time t5.

[0064] In addition, since the phase of the resonant current advances by π / 2 with respect to the resonant voltage, the time when the voltage of the voltage resonant capacitor Cv becomes Vin+N·Vo-α' again is the same as the time when the resonant current flowing through the magnetizing inductance becomes zero and is decreasing.

[0065] This resonant operation continues as long as the main switch QL and clamp switch QH remain OFF, and the voltage of the voltage resonant capacitor repeatedly becomes N·Vo after n·Tcvlm has elapsed from time t5 (n is a natural number). (In reality, this series resonance becomes an LCR series resonance due to the presence of parasitic resistance, and the amplitudes of the resonant voltage and resonant current attenuate, but the voltage of the voltage resonant capacitor becomes closest to N·Vo after n·Tcvlm has elapsed from time t5 (n is a natural number).)

[0066] Next, the operation during period T7 will be described with reference to FIG.

[0067] During period T7, at time t6 when the voltage of voltage resonant capacitor Cv returns to Vin+N·Vo-α´ again, that is, the time when the excitation current ILM of excitation inductance Lm becomes zero while decreasing, the clamp switch QH is turned on by the second pulse P2.

[0068] This causes the potential difference of the clamp switch QH to turn at a minimum value. In other words, bottom switching is performed, resulting in less switching loss.

[0069] Because the voltage of the clamp capacitor Cac is less than N·Vo, a voltage greater than Vo is not generated in the secondary winding Ns, and the diode Ds is non-conducting, so no resonant current due to the clamp capacitor Cac and leakage inductance Llk flows on either the primary or secondary side.

[0070] Therefore, in period T7, current flows in the order of Cac→QH→Lm→Llk→Cac, a current flows in the negative direction through the magnetizing inductance Lm, and at time t8, the clamp switch QH is turned off.

[0071] Here, time t6, that is, the time when the clamp switch is turned on by the second pulse, is the time when the voltage of the voltage resonant capacitor Cv returns to Vin+N·Vo-α' again, in other words, the time when the excitation current ILM of the excitation inductance Lm becomes zero while decreasing. In the explanation above, period T6 was described as Tcvlm, but the same effect can be obtained even if period T6 is n·Tcvlm (n is a natural number).

[0072] Next, the operation in period T8 will be described with reference to Fig. 11. In period T8, the clamp switch QH is turned off at time t7, causing a current to flow from Lm → Llk → Vin → Cv → Lm due to the energy stored in the excitation inductance Lm, discharging the charge in the voltage resonant capacitor Cv, and at time t8 the applied voltage of the main switch QL becomes zero volts.

[0073] FIG. 3 is a diagram showing a second operation waveform of the active clamp flyback converter according to the first embodiment of the present invention.

[0074] 2 in that there is no period T6, and the corresponding periods are the same as those described for the first operation waveform shown in FIG. 2, so detailed description of each period will be omitted.

[0075] As shown in FIG. 3, during period T1, when the main switch QL is turned on at time t0 and turned off at time t1, during period T2, the current stored in the leakage inductance Llk flows from the body diode BDH of the clamp switch QH to the clamp capacitor Cac, and a current W1 flows in the clamp switch QH.

[0076] Due to this current, the voltage across the clamp capacitor Cac becomes N·Vo+α, and the voltage Vds(L / S) of the main switch QL becomes Vin+N·Vo+α.

[0077] During period T3, the voltage of the main switch QL is Vin + N · Vo, and the excitation current flows from the secondary winding Ns to the output, so the primary winding becomes N · Vo, and the clamp switch QH is off, so the energy stored in the excitation inductance Lm of the transformer T is output from the secondary winding Ns to the secondary side, just like a normal flyback converter.

[0078] The first pulse P1 of the clamp switch QH is turned on at time t3. Time t3 corresponds to the period during which the energy stored in the excitation inductance Lm of the transformer T in a normal flyback converter is output from the secondary winding Ns to the secondary side, and corresponds to the time from Taclk / 2 to Taclk before time t5 when the excitation current ILM of the transformer T becomes zero.

[0079] During period T4, the primary winding of the transformer has a voltage of N·Vo, and the voltage of the clamp capacitor Cac is N·Vo+α, so when the clamp switch QH is turned on at time t3, a voltage of +α is applied to the leakage inductance Llk, causing the leakage inductance Llk and clamp capacitor Cac to resonate, and a resonant current W3 flows through the clamp switch QH.

[0080] The resonant current W3 becomes zero after Taclk / 2 has elapsed since time t3, and the voltage of the clamp capacitor Cv, which was N·Vo+α at time t3, becomes N·Vo-α' at time t4. The resonant current W3 that flows during period T3 flows in the same direction as the current accumulated in the exciting inductance Lm is discharged to the secondary side, so that a resonant current W4 superimposed on the exciting current flows on the secondary side.

[0081] In period T5, time t4 is the time when 1 / 2 the resonance period of resonant current W3 has elapsed from time t3. At time t4, resonant current W3 becomes zero, and from time t4, the resonant current flows from the source to the drain of the clamp switch (in the negative direction).

[0082] However, since this resonant current flows in the direction opposite to the direction in which the current stored in the excitation inductance Lm is discharged to the secondary side, the current waveform of W5 is limited by the excitation current.

[0083] The clamp switch QH turns off while the resonant current limited by this reverse-flowing excitation current is flowing. When the clamp switch QH is turned off, the resonant current limited by the excitation current flows to the body diode BDH. The charge Qp of the resonant current flowing in period T4 discharges the voltage α, which is the voltage difference between the clamp capacitor Cac and the back electromotive force generated in the primary winding, to voltage -α', so the charge Qn of the current flowing in period T5 is Qp / 2 or less, and the voltage of the clamp capacitor at current W5 in period T5 does not exceed N·Vo.

[0084] In period T7, at time t5 the excitation current ILM decreases to zero, and the clamp switch QH turns on at time t5, causing a negative current W6 to flow through the excitation inductance Lm. At time t5, the voltage of the clamp capacitor is below N·Vo, so no resonant current flows even when the clamp switch QH is turned on. Therefore, no resonant current flows on the secondary side either.

[0085] In period T8, when the clamp switch QH is turned off at time t7, the negative current W6 of the magnetizing inductance discharges the voltage resonant capacitor Cv, and the voltage of the main switch QL reaches zero volts at time t8.

[0086] At time t8, when the voltage of the main switch QL becomes zero, the main switch QL is turned on, and the main switch QL performs zero-voltage switching, thereby forming a highly efficient, low-noise power supply.

[0087] Furthermore, the current in the secondary-side diode Ds becomes a continuous current like W2+W4, and no current flows in periods T7 and T8, so that synchronous rectification can be easily achieved and a more efficient power supply can be constructed.

[0088] In the second operating waveform of the active clamp flyback converter according to the first embodiment of the present invention shown in Fig. 3, the first first pulse P1 of the clamp switch QH may be turned off at time t5 as long as it is within period T5. Also, the second second pulse P2 of the clamp switch QH is at time t5, which is the end of period T5. Therefore, in the operating mode shown in Fig. 3, the first pulse P1 and the second pulse P2 may be coupled together for driving.

[0089] (Specific example of the first embodiment) Fig. 12 is a diagram showing a specific example of an active clamp flyback converter AFC according to a first embodiment of the present invention. Fig. 13 is a diagram showing a first operation waveform of the AFC shown in Fig. 12. Here, the configuration of Fig. 12 will be explained with reference to Fig. 13.

[0090] 12 differs from the configuration shown in FIG. 1 in that an auxiliary winding Na is provided in the transformer T and in that a control unit 2a is provided.

[0091] The auxiliary winding Na is electromagnetically coupled to the primary winding Np and the secondary winding Ns. One end of the auxiliary winding Na is connected to one end of a resistor Ra1 and the anode of a diode Da, the cathode of the diode Da is connected to one end of a capacitor Ca, and the other end of the auxiliary winding Na is connected to one end of a resistor Ra2 and the other end of the capacitor Ca, and the other end of the capacitor Ca is grounded.

[0092] The other end of resistor Ra1 and the other end of resistor Ra2 detect a voltage Vna obtained by resistively dividing the voltage of the auxiliary winding Na, and the detected voltage Vna is output to the excitation current detection unit 30, threshold generation unit 31, and bottom detection unit 25 of the control unit 2a.

[0093] The control unit 2a includes an FB control unit 20, a first on signal unit 21, a first drive circuit 22, a bottom detection unit 25, an excitation current detection unit 30, a threshold generation unit 31, an on timing detection unit 32, a second on signal unit 33, a third on signal unit 34, and a second drive circuit 35.

[0094] The FB control unit 20 generates a feed control signal based on the output voltage detected by the output voltage detection unit 1 and a predetermined value, and outputs the feed control signal to the first on signal unit 21. The first on signal unit 21 generates a first on signal for turning on the main switch QL based on the feed control signal. The first drive circuit 22 turns on the main switch QL by the first on signal from the first on signal unit 21.

[0095] The excitation current detection unit 30 detects an excitation voltage VLm proportional to the excitation current based on the voltage Vna. The threshold generation unit 31 generates a threshold Vth proportional to the output voltage Vo based on the voltage Vna. The on-timing detection unit 32 detects the on-timing at which the second drive circuit 35 turns on the first pulse.

[0096] The zero excitation current detection unit 26 detects the time when the excitation voltage VLm detected by the excitation current detection unit 30 becomes zero with a negative gradient.

[0097] The bottom detection unit 25 detects the timing when the voltage of the clamp switch QH reaches the bottom (the top of the voltage of the main switch QL) based on the voltage Vna.

[0098] The second ON signal section 33 generates a second ON signal P1 at the timing detected by the ON timing detection section 32. The third ON signal section 34 generates a third ON signal P2 at the timing detected by the bottom detection section 25.

[0099] The second drive circuit 35 turns on the clamp switch QH at the on timing when the second on signal P1 from the second on signal unit 33 turns on the first pulse for the first time.

[0100] Moreover, the second drive circuit 35 turns on the clamp switch QH at the on timing when the third on signal P2 from the third on signal unit 34 turns on the second pulse for the second time.

[0101] Next, Fig. 13 shows a first operating waveform. The operation of the active clamp flyback converter shown in Fig. 12 will be described with reference to Fig. 13 and the first operating waveform of the active clamp flyback converter according to the first embodiment of the present invention shown in Fig. 2. In Fig. 13, Vna is a voltage obtained by dividing the voltage of the auxiliary winding Na by resistors Ra1 and Ra2. VLm is a voltage proportional to the excitation current detected by the excitation current detection unit 30. Vth is a voltage proportional to the output voltage detected by the threshold generation unit 31.

[0102] The other reference characters are the same as those shown in FIG. 2 and FIG. 3, and therefore their explanation will be omitted.

[0103] In FIG. 13, when the main switch QL is turned on, a voltage of Vin is applied to the primary winding Np, and when the main switch QL is turned off, a voltage of -N·Vo is generated in the primary winding Np.

[0104] The excitation current detection unit 30 detects the excitation current of the excitation inductance Lm of the primary winding Np based on the following operation. The current I flowing through the inductance increases at I=V / L×t (t is time) where V is the voltage across the inductance and L is the inductance value. The voltage Vna is proportional to the voltage across the auxiliary winding Na, and since the auxiliary winding Na is electromagnetically coupled to the primary winding Np and the secondary winding Ns, the voltage across the auxiliary winding Na is proportional to the voltage across the primary winding Np and is also proportional to the voltage across the excitation inductance Lm. Therefore, the voltage Vna is used by an integrator to detect an excitation voltage VLm proportional to the excitation current ILM.

[0105] The threshold generating unit 31 generates a voltage Vth proportional to the output voltage based on the positive voltage of the auxiliary winding Na. That is, the auxiliary winding Na is electromagnetically coupled to the primary winding Np and the secondary winding Ns, and the auxiliary winding Na and the secondary winding Ns have the same winding direction, so that the positive voltage of the auxiliary winding Na generates a voltage proportional to the positive voltage of the secondary winding Ns, i.e., the output voltage, and therefore the voltage Vth proportional to the output voltage can be generated based on the positive voltage of the auxiliary winding voltage Vna.

[0106] The on-timing detection unit 32 detects the on-timing t11 when the gradient of the excitation voltage VLm detected by the excitation current detection unit 30 is negative (when the first on signal is off) and the excitation voltage VLm becomes the threshold value Vth generated by the threshold value generation unit 31.

[0107] The second ON signal section 33 generates a preset first pulse P1 at the timing t11 detected by the ON timing detection section 32.

[0108] The bottom detection unit 25 detects the zero voltage at which the voltage Vna goes from negative to positive after the first pulse P1 generated by the second on signal unit 33, with the resonant period of the voltage resonant capacitor Cv and the transformer excitation inductance Lm being Tcvlm, and detects the on timing t13 that is delayed by Tcvlm / 4 from the time at which the zero voltage was detected.

[0109] The third ON signal section 34 generates a second pulse P2 that is set in advance at the timing t13 detected by the bottom detection section 25.

[0110] The second drive circuit 35 generates a drive signal HG from the first pulse P1 of the second on signal section 33 and the second pulse P2 of the third on signal section 34, and turns on the clamp switch QH a first time with the first pulse P1 and a second time with the second pulse P2.

[0111] The threshold generation unit 31 adjusts Vth so that the on-timing t11 detected by the on-timing detection unit 32 is detected before Taclk / 2 to Taclk from the time tz at which the excitation voltage VLm becomes zero, and generates a voltage proportional to the output voltage based on the positive voltage of the voltage Vna.

[0112] A setting example is explained in FIG.

[0113] 15, VLm(20) is the detection voltage of the excitation current detection unit 30 when the output voltage is 20 V. VLm(10) is the detection voltage of the excitation current detection unit 30 when the output voltage is 10 V. Since the excitation current has a relational expression of N·Vo / Lm·t, when the output voltage is halved, the slope also becomes halved.

[0114] Vth(20) is adjusted so that it is detected 0.8 Taclk before the time tz between Taclk / 2 and Taclk when the excitation voltage VLm(20) becomes zero when the output voltage is 20V, for example, as shown in Fig. 15. Then, in the second on signal section 33, the on width is set in the range of Taclk / 2 to Taclk and the off timing is set until the excitation current becomes zero. For example, in Fig. 15, if it is set to 0.6 Taclk, the off timing is 0.2 Taclk before the excitation current ILM becomes zero.

[0115] Since the threshold generation unit 31 adjusts Vth in proportion to the output voltage based on the positive voltage of the voltage Vna, when the output voltage becomes 10 V as shown by the dashed line VLm(10) in Fig. 15, the slope of the excitation voltage VLm detected by the excitation current detection unit 30 becomes 1 / 2, but the threshold voltage generated by the threshold generation unit 31 becomes Vth(10) shown by the dashed line. The on-timing t11 detected by the on-timing detection unit 32 becomes t11', which is 0.8 Taclk before the excitation current becomes zero, and the off-timing t12 becomes t12', which is 0.2 Taclk before the excitation current ILM becomes zero.

[0116] In addition, the negative slope of the excitation current does not change even if the input voltage Vin changes. Therefore, when the output voltage Vo is a certain voltage, for example 20 V, by adjusting the threshold value Vth as described above, the same effect as that of the active clamp flyback converter according to the first embodiment can be obtained even if the input voltage Vin or the output voltage Vo changes.

[0117] Next, Fig. 14 shows a second operating waveform. The operation of the active clamp flyback converter shown in Fig. 12 will be described with reference to Fig. 14 and the second operating waveform of the active clamp flyback converter according to the first embodiment of the present invention shown in Fig. 3.

[0118] The second operation waveform shown in Fig. 3 differs from the first operation waveform shown in Fig. 2 in that it does not have period T6. Because the first operation waveform has period T6, the bottom detection unit 25 detects the bottom of the clamp switch QH and detects the on timing t13, but the second operation waveform does not have period T6. Therefore, the third on signal unit 34 generates a pulse P2 at the timing when the excitation current detected by the excitation current zero detection unit 26 becomes zero.

[0119] Since the other corresponding periods have the same operation as that of the first operational waveform shown in Fig. 2, detailed description will be omitted, but the same effect as that of the active clamp flyback converter according to the first embodiment can be obtained. In the second operational waveform shown in Fig. 14, the first pulse P1 of HG and the second pulse P2 of HG that drive the clamp switch QH may be connected.

[0120] Second embodiment FIG. 16 is a configuration diagram of an active clamp flyback converter according to a second embodiment of the present invention.

[0121] The active clamp flyback converter shown in FIG. 16 includes a current detector 3 and a controller 2b.

[0122] During the off period of the main switch QL, the control unit 2b sets the on timing of the first pulse P1 of the clamp switch QH to a timing at which the amount of charge required for the excitation current that charges the clamp capacitor Cac to reach zero becomes smaller than half the amount of charge of the resonant current that discharges the clamp capacitor Cac due to the first pulse on.

[0123] This makes it possible to prevent a resonant current from flowing when the second pulse is turned on.

[0124] The control unit 2b includes an FB control unit 20, a first on signal unit 21, a first drive circuit 22, an excitation current detection unit 30, a zero excitation current detection unit 26, a bottom detection unit 25, a second on signal unit 33, a third on signal unit 34, a second drive circuit 35, an integration circuit 37, and a delay circuit 38.

[0125] The FB control unit 20, the first on signal unit 21, the first drive circuit 22, the excitation current detection unit 30, the zero excitation current detection unit 26, the bottom detection unit 25, the second on signal unit 33, and the third on signal unit 34 have been explained in FIG. 12, so their explanation will be omitted.

[0126] The second ON signal has an ON width that is equal to or greater than half the resonance period of the clamp capacitor Cac and the leakage inductance Llk and is equal to or less than the resonance period. The third ON signal is turned ON in the same manner as in the first embodiment, and therefore a description thereof will be omitted.

[0127] In the current detection unit 3, one end of a capacitor Cs is connected to the connection point between the clamp capacitor Cac and the clamp switch QH, and the other end of the capacitor Cs is grounded via a resistor Rs.

[0128] The current detector 3 detects the positive and negative currents flowing through the clamp capacitor Cac when the first pulse is turned on. The detected currents are proportional to the capacitance ratio between the clamp capacitor Cac and the capacitor CS.

[0129] Integration circuit 37 integrates the current detected by current detection unit 3 in the direction discharging the clamp capacitor to determine the amount of discharge, and integrates the current detected by current detection unit 3 in the direction charging the clamp capacitor to determine the amount of charge.

[0130] Specifically, when an integration coefficient KP in the direction in which the clamp capacitor Cac is discharged and an integration coefficient KN in the direction in which the clamp capacitor Cac is charged, the integration circuit 37 has a ratio of KP:KN=1:2.

[0131] The delay circuit 38 delays the ON timing of the first pulse P1 with respect to the first ON signal from the first ON signal unit 21 so that the charge amount obtained by the integration circuit 37 becomes half or less of the discharge amount.

[0132] The second ON signal section 33 outputs to the first drive circuit 22 a second ON signal in which the ON timing of the first first pulse P1 is delayed by the delay circuit 38 with respect to the first ON signal.

[0133] FIG. 17 is a diagram showing operation waveforms of the active clamp flyback converter according to the second embodiment of the present invention.

[0134] 17, the ON period of the first first pulse P1 is equal to or longer than Taclk / 2. The turn-on timing of the first first pulse P1 is when the excitation current is equal to or greater than zero.

[0135] The delay circuit 38 delays the on-timing of the first first pulse P1 until time t3 so that the charge amount of the negative-going waveform W5 that flows after Taclk / 2 has elapsed since the first first pulse P1 is turned on is smaller than half the discharge amount of the positive-going waveform W3 of the current Id(H / S) of the clamp switch QH that flows during the period of Taclk / 2 from the turn-on of the first first pulse P1.

[0136] By delaying and controlling the on-timing of the first first pulse P1 in this way, the voltage of the clamp capacitor Cac is N·Vo+α at time t3, drops to N·Vo-α´ at time t4, and is charged to N·Vo at time t5. For this reason, no resonant current flows on the secondary side at the second second pulse P2. [Industrial Applicability]

[0137] The active clamp flyback converter of the present invention is applicable to switching devices. [Explanation of symbols]

[0138] Vin DC power supply QL Main Switch QH Clamp Switch T transformer Np primary winding Ns Secondary winding Na Auxiliary Winding Lm Excitation inductance Llk Leakage inductance Cac Clamp Capacitor Cv Voltage Resonant Capacitor BDL, BDH Body diode Ds Synchronous rectification element Co Output capacitor Ca, Cs capacitors Ra1, Ra2, Rs resistance 1 Output voltage detection section 2,2a,2b control section 3 Current detection section 20 FB control section 21 First ON signal section 22 First drive circuit 25 Bottom detection section 26 Excitation current zero detection section 30 Excitation current detection section 31 Threshold Generation Unit 32 On-timing detection section 33 Second ON signal section 34 3rd ON signal section 35 Second drive circuit 37 Integral circuit 38 Delay Circuit

Claims

1. An active clamp flyback converter including: a first series circuit having a main switch and a primary winding connected in series to both ends of a DC power supply; a clamp switch and a clamp capacitor connected in series to a connection point of the first series circuit and one end of the DC power supply; and a control unit that turns on and off the main switch and the clamp switch, the control unit includes a first on signal for turning on and off the main switch, a second on signal for turning on and off the clamp switch while the main switch is off, and a third on signal for turning on and off the clamp switch again after the second on signal, an on-timing is set so that a resonant current which is formed when the clamp switch is turned on by the second on signal and which reverses after 1 / 2 a resonant period of a resonant circuit of the clamp capacitor and leakage inductance and flows in a direction to charge the clamp capacitor is limited by an excitation current of an excitation inductance of the primary winding, an on-period of the second on signal is set to be equal to or more than 1 / 2 the resonant period so that the resonant current which discharges the clamp capacitor when the clamp switch is turned on and the current limited by the resonant current reversed and the excitation current charge and discharge the clamp capacitor only once, the clamp switch is turned on and off by the second on signal, the third on signal is output after the excitation current of the excitation inductance becomes zero, and the clamp switch is turned on by the third on signal to pass a negative current through the excitation inductance.

2. the control unit includes a bottom detection means for detecting a bottom of voltage oscillation of the clamp switch after the excitation current of the excitation inductance becomes zero, 2. The active clamp flyback converter according to claim 1, wherein the third on signal is output at a time when the voltage of the clamp switch is minimum detected by the bottom detection means, to turn on the clamp switch and cause a current in a negative direction to flow through the excitation inductance.

3. The control unit determines an on-timing of the second on-signal as follows: From 1 / 2 of the resonance period to just before the resonance period from the time when the excitation current becomes zero, 3. The active clamp flyback converter according to claim 1, wherein the off timing is set to a time until the excitation current becomes zero, and the on period of the second on signal is set to be equal to or greater than ½ of a resonance period between the clamp capacitor and a leakage inductance of the primary winding.

4. the control unit includes an excitation current detection unit that detects the excitation current and a threshold generation unit that generates a threshold, 4. The active clamp flyback converter according to claim 1, wherein the second on signal is sent as an on timing of the second on signal when the excitation current detected by the excitation current detection unit becomes the threshold value generated by the threshold value generation unit.

5. The active clamp flyback converter according to claim 4 , wherein the threshold generating unit changes the threshold in response to an output voltage.

6. 2. The active clamp flyback converter according to claim 1, wherein the control unit sets an on-period of the second on signal to be equal to or greater than half and equal to or less than a resonance period between the clamp capacitor and a leakage inductance of the primary winding, and sets an on-timing of the second on signal to a timing at which, when the clamp switch is turned on by the second on signal, the resonance current is limited to the excitation current and an amount of charge of a current flowing in a direction to charge the clamp capacitor becomes smaller than half the amount of charge of a resonance current flowing in a direction to discharge the clamp capacitor.

7. a current detection unit that detects a current in a discharging direction and a current in a charging direction flowing through the clamp capacitor in response to the second on signal; The control unit is an integrating circuit that integrates the current in the discharging direction detected by the current detection unit to obtain a discharging integral amount, and that integrates the current in the charging direction to obtain a charging integral amount; a delay circuit that delays an on-timing of the second on-signal so that the charge integral amount obtained by the integrating circuit becomes equal to or less than half of the discharge integral amount; 3. The active clamp flyback converter of claim 2, comprising:

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