Switching amplifier

Abstract

This switching amplifier having a push-pull configuration and comprising a narrow band filter on an output side also includes: a partial resonance circuit for suppressing a first surge voltage caused by a square wave; and an LCR series resonator for suppressing a second surge voltage caused by a harmonic frequency component. The waveform shape of a drain-source voltage Vds of the switching element is a trapezoidal waveform. As a result, the surge voltages of the first surge voltage caused by the square wave and the second surge voltage caused by the harmonic frequency component are suppressed, the frequency characteristics of the output of the switching amplifier are expanded, and the output frequency is made variable.

Description

Switching Amplifier

[0001] The present invention relates to a switching amplifier using a switching mode system such as class D, class E, class F, inverse class F, or class EF, and more particularly to a switching amplifier suitable for a variable frequency high frequency power supply capable of varying the output frequency over a wide band.

[0002] A switching amplifier generates a square wave through switching operation and amplifies the output by modulating the pulse width of the generated square wave according to the input signal level.One circuit method for switching amplifiers is to operate two switching elements in opposite phases, alternately source and sink current to increase the output current, and by applying a push-pull circuit, a large-capacity output can be obtained.

[0003] In a push-pull switching amplifier, the drain-source voltage generated by switching operation is a square waveform. The drain-source voltage waveform is distorted (surge) by leakage inductance. It is known that in order to make the drain-source voltage waveform square, the degree of coupling of the output transformer is increased to reduce the leakage inductance (Non-Patent Document 1).

[0004] Microsemi PPG Application Note1812 / 13.56 MHz, Class D Push-Pull, 2KW RF Generator with Microsemi DRF1300 Power MOSFET Hybrid / By Gui Choi 2011 September

[0005] In a switching amplifier, a problem occurs in that a surge voltage (hereinafter referred to as a first surge voltage) is generated due to leakage inductance of an output transformer or the like during a switching operation that generates a square wave.

[0006] Furthermore, the output waveform of the output transformer of a switching amplifier contains harmonic frequency components in addition to the fundamental frequency component. One possible harmonic countermeasure to remove these harmonic frequency components from the output is to provide a narrow-band filter, such as an LPF, on the secondary side of the output transformer. The narrow-band filter passes the fundamental frequency component while blocking the harmonic frequency components. These harmonic frequency components are reflected by the narrow-band filter and returned to the switching element. The returned harmonic frequency components act as a surge voltage (hereinafter referred to as a second surge voltage) on the switching element. In a VHF switching amplifier using switching elements with a wide frequency band, such as LDMOS, the frequencies of the harmonic frequency components reflected by the narrow-band filter and returned are also within the frequency band of the switching element, and are therefore affected by the surge voltage, making the problem of the second surge voltage even more pronounced.

[0007] 14A and 14B are diagrams illustrating a second surge voltage due to harmonic frequency components, in which, in an example circuit of a push-pull switching amplifier of a class D amplifier, Fig. 14A shows an example circuit without a narrow-band filter, and Fig. 14B shows an example circuit with a narrow-band filter.

[0008] 14A includes a switching unit 111 having two switching elements 111A and 111B, an output transformer 112 connected between the drains of the switching elements 111A and 111B, and a DC power supply 113 connected to a center tap of a primary winding 112a of the output transformer 112. The switching amplifier 101a forms a push-pull circuit 114 by alternately switching the switching elements 111A and 111B in response to gate signals Pg of opposite phases from signal sources 110A and 110B. A current is transmitted to a secondary winding 112b of the output transformer 112 by the push-pull circuit 114. The transmitted current is supplied to a load 117 connected to an output terminal 116.

[0009] 14A, the drain-source voltages VdsA and VdsB of switching elements 111A and 111B are switched to produce a square wave If0 and a current Inf0 of harmonic frequency components, which are transmitted to and consumed by load 117. In this configuration, the output waveform also becomes a square wave, and therefore is not suitable as a high-frequency power supply for semiconductor manufacturing equipment that must output a sine wave waveform.

[0010] 14B, a switching amplifier 101b has a sinusoidal output waveform due to the provision of narrow-band filter 115 on the secondary side of output transformer 112. While a sinusoidal output voltage can be obtained with this configuration that includes narrow-band filter 115, the waveforms of drain-source voltages VdsA and VdsB cannot be maintained as square waves. The inventors of the present application discovered that the reason the square waves are not maintained is that current Inf0, which is a harmonic frequency component contained in drain-source voltages VdsA and VdsB, is reflected by narrow-band filter 115 and returns to switching elements 111A and 111B.

[0011] In VHF band amplifiers, switching elements with wide frequency bands, such as LDMOS, are used to cover a wide frequency band. Therefore, in switching amplifiers with wide output frequency characteristics, it was not possible to proactively install a narrow-band filter on the secondary side of the output transformer to prevent harmonic frequency components from being reflected back. This resulted in the problem of being unable to achieve both wide bandwidth and a sinusoidal output waveform.

[0012] Therefore, in a VHF band amplifier, two surge voltages are generated during switching operation: a first surge voltage due to leakage inductance of the output transformer, etc., and a second surge voltage due to harmonic frequency components reflected by the narrow-band filter. These surge voltages are factors that prevent the output frequency of a high-frequency power supply using a switching amplifier from being broadened.

[0013] In order to widen the frequency characteristics of the output of a switching amplifier and obtain a stable output even when the output frequency is variable, it is necessary for the waveform shape of the drain-source voltage Vds of the switching element to be a trapezoidal waveform in which the first surge voltage caused by a square wave and the second surge voltage caused by harmonic frequency components are suppressed.

[0014] The present invention aims to solve the above-mentioned problems of the conventional technology by suppressing the first surge voltage caused by a square wave and the second surge voltage caused by harmonic frequency components, widening the frequency characteristic of the output of a switching amplifier, and making the output frequency variable.

[0015] The switching amplifier according to the present invention is a push-pull switching amplifier equipped with a narrow-band filter on the output side, and is equipped with a partial resonant circuit that suppresses a first surge voltage caused by a square wave and an LCR series resonant element that suppresses a second surge voltage caused by harmonic frequency components, and the waveform shape of the drain-source voltage Vds of the switching element is made trapezoidal, thereby widening the frequency characteristic of the output of the switching amplifier and making the output frequency variable.

[0016] (Drain-source voltage Vds having a trapezoidal waveform) A trapezoidal waveform has a slope in the section where the voltage rises and takes a predetermined time during the rise, and a constant peak value in the section where the voltage falls after rising.

[0017] The ramp waveform suppresses surge voltages and harmonic frequency components in the drain-source voltage Vds of the switching element that occur during voltage rise and fall. A waveform that maintains a constant peak value keeps the drain-source voltage Vds constant even when the output frequency is varied and swept over a wide band, so the output gain can be kept constant.

[0018] (Configuration of Switching Amplifier) ​​A switching amplifier according to the present invention comprises the following components: (a) a transformer-coupled push-pull circuit composed of a pair of switching elements that alternately perform on / off switching operations in response to input signals of opposite phases, a primary winding to both ends of which an output signal of the switching elements is input, a secondary winding from which the output signal is taken out, and a transformer having an external power supply connected to a center tap of the primary winding, (b) a narrow-band filter connected to the output end of the secondary winding of the transformer, (c) a pair of partial resonant circuits composed of an inductance component on the drain side of each switching element and a capacitance component between the drain and source of each switching element, and (d) an LCR series resonator composed of a series circuit of inductance, capacitance, and resistance connected between the drains of each switching element. The switching elements included in the switching amplifier according to the present invention can be MOSFETs, and in the case of power MOSFETs, LDMOS or VDMOS with a DMOS structure can be used.

[0019] (Push-pull circuit) A push-pull circuit alternately switches a pair of switching elements on and off using input signals of opposite phases. This alternate switching operation alternately connects each switching element to an external power supply connected to a center tap of the transformer's primary winding, causing an inductance current to flow in alternate opposite directions through the primary winding. This inductance current causes a load current to flow through the transformer's secondary winding due to transformer coupling. The inductance current flowing through the primary winding alternately flows in the opposite direction. This inductance current on the primary winding side corresponds to the secondary winding alternating between sourcing and drawing in current, so the transmitted load current flows through the secondary winding.

[0020] The load current induced in the secondary winding contains fundamental frequency components and harmonic frequency components that correspond to the switching frequency of the switching element. Because the even-order harmonic frequency components cancel each other out due to the characteristics of the push-pull circuit, the load current contains only odd-order harmonic frequency components.

[0021] (Narrow-band filter) The narrow-band filter passes the fundamental frequency component contained in the load current of the secondary winding and blocks the harmonic frequency component. The harmonic frequency component is reflected by the narrow-band filter and returned to the switching element. The harmonic frequency component returned to the switching element becomes a surge voltage (second surge voltage) against the switching element.

[0022] (LCR series resonator) The LCR series resonator is composed of a series circuit of inductance, capacitance, and resistance, and is connected between the drains of a pair of switching elements. The resonant frequency of the series circuit is set to match the frequency of the harmonic frequency components of the load current.

[0023] (Parallel Connection of LCR Series Resonators) The LCR series resonators can widen the frequency band of harmonic frequency components of the load current by configuring a plurality of series circuits with different resonant frequencies connected in parallel.

[0024] The LCR series resonator, which is formed by connecting two series circuits in parallel, includes a first series circuit having a resonant frequency lower than the harmonic frequency n×f0 of the high frequency output, where f0 is the fundamental frequency of the high frequency output output from the secondary side of the transformer, and a second series circuit having a resonant frequency higher than the harmonic frequency n×f0 of the high frequency output.

[0025] When the LCR series resonator is configured with a plurality of series circuits connected in parallel, the number of series circuits connected in parallel is not limited to two and may be any number.

[0026] (Suppression of the second surge voltage by the LCR series resonator) The harmonic frequency components reflected by the narrowband filter flow to ground via the LCR series resonator and the switching element in the on state. At this time, the harmonic frequency components are consumed and attenuated by the resistance of the series circuit. This attenuation of the harmonic frequency components suppresses the second surge voltage caused by the harmonic frequency components applied to the switching element. Suppression of the second surge voltage suppresses fluctuations in the peak value of the drain-source voltage Vds of the switching element and keeps it approximately constant.

[0027] (Partial Resonant Circuit) The partial resonant circuit is composed of an inductance component on the drain side of the switching element and a capacitance component between the drain and source of the switching element. The partial resonant circuit is composed of each switching element of a pair of switching elements.

[0028] The inductance component (Ls) of each partial resonant circuit is either a series inductance (Ls1) connected in series between the switching element and one end of the transformer, a leakage inductance (Lleak) of the output transformer, or a wiring inductance (Lline) between the switching element and one end of the transformer, or a combination of any of these.

[0029] The capacitance component (Ctotal) of each partial resonant circuit is the output parasitic capacitance (Coss) of the switching element, or a combined capacitance of the output parasitic capacitance (Coss) of the switching element and an additional capacitance (Cp) connected in parallel to the switching element.

[0030] When the switching element changes from the on state to the off state, the charge of the capacitance component (Ctotal) that was discharged in the on state is charged by the first resonance due to the partial resonance of the primary current of the transformer. During this charging, the drain-source voltage Vds of the switching element increases with a time constant τr determined by the inductance component (Ls) and the capacitance component (Ctotal).

[0031] On the other hand, when the switching element changes from the OFF state to the ON state, the charge of the capacitance component (Ctotal) that was charged in the OFF state is discharged by the second resonance due to the partial resonance of the primary current of the transformer. During this discharge, the drain-source voltage Vds of the switching element decreases with a time constant τf determined by the inductance component (Ls) and the capacitance component (Ctotal).

[0032] During the dead time period, the capacitance component (Ctotal) of the first resonance is the capacitance component (Ctotal) of two switching elements connected in series. On the other hand, the capacitance component (Ctotal) of the second resonance is the capacitance component (Ctotal) of only one switching element, so the time constant τr of the first resonance is smaller than the time constant τf of the second resonance, and the voltage rises faster. Due to the first and second resonances, the drain-source voltage Vds has a trapezoidal waveform.

[0033] (Suppression of first surge voltage by partial resonant circuit and LCR series resonant body) Due to partial resonance of the primary current of the transformer, the drain-source voltage Vds of the switching element increases or decreases with time constants τr and τf during the dead time period, so that the surge voltage generated in the drain-source voltage Vds is suppressed.

[0034] Furthermore, during periods excluding dead time, the charging current of the first resonance and the discharging current of the second resonance due to partial resonance flow between the two drains via the LCR series resonator. The charging and discharging currents that flow through the LCR series resonator are consumed and attenuated by the resistance Rr of the LCR series resonator. During periods excluding dead time, the charging and discharging currents due to partial resonance increase and decrease with time constants τr and τf, and are consumed and attenuated by the LCR series resonator, thereby suppressing the occurrence of the first surge voltage in the drain-source voltage Vds.

[0035] Therefore, the first surge voltage caused by the square wave is suppressed by the partial resonant circuit and the LCR series resonant body, and the waveform of the drain-source voltage Vds becomes a trapezoidal wave.

[0036] (Resonant frequency and bandwidth of partial resonant circuit) The resonant frequency of the pair of partial resonant circuits is in the range from the fundamental frequency f0 to the third harmonic frequency 3×f0 of the high frequency output from the secondary side of the transformer, and reduces the impedance in the frequency range where the impedance is high due to the partial resonant circuit, while setting a predetermined time constant for the rise and fall of the drain-source voltage Vds of the LDMOS.

[0037] (Resonant Frequency and Bandwidth of LCR Series Resonator) When the fundamental frequency of the high-frequency output from the secondary side of the transformer is f0, the resonant frequency of the LCR series resonator is a harmonic frequency n×f0 (n is an integer), and the bandwidth of the resonant frequency is set to a frequency range of (n×f0-min to n×f0-max) according to the variable frequency range (f0-min to f0-max) of the fundamental frequency f0.

[0038] (Functions of the Switching Amplifier of the Present Invention) The switching amplifier of the present invention suppresses surge voltages, widens the frequency characteristic of the output of the switching amplifier, and makes the output frequency variable, thereby achieving the following secondary effects.

[0039] Eliminates the need for leakage countermeasures for the output transformer. Prior art documents disclose that in order to obtain a square-wave drain-source voltage waveform, the degree of coupling of the output transformer is increased to reduce leakage inductance. To reduce leakage inductance, a structure is shown in which a magnetic core is prepared as the output transformer material and a twisted-wire winding is wound around this core. This is a common method for increasing coupling and reducing leakage inductance. Another method is to select a material with a high relative permeability μ for the core material. However, because losses due to the core material increase in proportion to the relative permeability μ, it is difficult to design a circuit that can withstand continuous high output voltages.

[0040] According to the present invention, it is possible to eliminate the need for a core material with high relative permeability or a twisted wire structure as a leakage countermeasure for the output transformer, and to realize an amplifier capable of continuously outputting high output power.

[0041] Generally, even when using a coreless, air-core output transformer with large leakage inductance or a parallel-winding planar transformer with poor coupling due to its winding structure, it is possible to insert an LPF (low-pass filter) on the secondary side of the transformer, thereby achieving a class D push-pull amplifier with few output harmonics.

[0042] The peak value of the drain-source voltage Vds can be made substantially the same over the entire wide band, so the power gain can be made constant.

[0043] According to the present invention, the waveform of the drain-source voltage Vds always has a similar peak value, so that output fluctuations can be suppressed, and fluctuations in power gain due to frequency variation, which are prominent in class AB and class C amplifiers, are suppressed, resulting in flat output characteristics.

[0044] Even if the leakage inductance is large, a stable trapezoidal Vds waveform can be obtained, enabling a wide bandwidth.

[0045] - An LPF (low-pass filter) can be inserted on the secondary side of the output transformer, reducing output harmonics.

[0046] The surge voltage due to partial resonance can be suppressed, and the drain-source voltage Vds of the switching can be prevented from exceeding the breakdown voltage of the switching element.

[0047] As described above, according to the present invention, it is possible to suppress the first surge voltage caused by a square wave and the second surge voltage caused by harmonic frequency components, widen the frequency characteristic of the output of the switching amplifier, and make the output frequency variable.

[0048] FIG. 1 is a diagram illustrating a first configuration of a switching amplifier according to the present invention. FIG. 2 is a diagram illustrating a second configuration of a switching amplifier according to the present invention. FIG. 3 is a diagram comparing the pass characteristics of an LCR series resonator and an LC series resonator. FIG. 4 is a diagram illustrating an equivalent circuit of an LCR series resonator and an LC series resonator. FIG. 5 is a diagram illustrating signals at each section of a switching amplifier according to the present invention. FIG. 6 is a diagram illustrating an example of operation in each section of a switching amplifier according to the present invention. FIG. 7 is a diagram illustrating an example of operation in each section of a switching amplifier according to the present invention. FIG. 8 is a diagram illustrating an example of operation in each section of a switching amplifier according to the present invention. FIG. 9 is a diagram illustrating an example of operation in each section of a switching amplifier according to the present invention. FIG. 10 is a diagram illustrating an example of operation in each section of a switching amplifier according to the present invention. FIG. 11 is a diagram illustrating an example of operation in each section of a switching amplifier according to the present invention. FIG. 12 is a diagram illustrating an example of operation in each section of a switching amplifier according to the present invention. 1 is a Smith chart for explaining the relationship between a frequency band of partial resonance and a frequency band of high impedance; FIG. 2 is a frequency characteristics diagram for explaining the relationship between a frequency band of partial resonance and a frequency band of high impedance; FIG. 3 is a diagram for explaining a second surge voltage due to a harmonic frequency component; FIG. 4 is a diagram for explaining a second surge voltage due to a harmonic frequency component;

[0049] (1) Schematic Configuration of Switching Amplifier The schematic configuration of a switching amplifier of the present invention will be described below with reference to Figures 1 and 2. Figure 1 shows a first example configuration including one LCR series resonator, and Figure 2 shows a second example configuration including two LCR series resonators connected in parallel.

[0050] (1-1: First Configuration Example) The first configuration example of the switching amplifier 1 includes a switching unit 11, an output transformer 12, an external power supply consisting of a DC power supply 13, and a switching amplifier including a narrow-band filter 15, as well as an LCR series resonator 20 and a partial resonant circuit 21 (21A, 21B).

[0051] The switching unit 11 includes two switching elements. MOSFETs are used as the switching elements, and power MOSFETs are used for large current applications. DMOSs such as LDMOSs and VDMOSs are used as power MOSFETs. Here, an LDMOS is used as an example of a switching element, and the two switching elements included in the switching unit 11 are represented as LDMOS (11A) and LDMOS (11B).

[0052] The gate terminal of the LDMOS (11A) receives a gate signal Pg from the signal source 10A, and switches the on / off state between the drain terminal and the grounded source terminal. Similarly, the gate terminal of the LDMOS (11B) receives a gate signal Pg from the signal source 10B, and switches the on / off state between the drain terminal and the grounded source terminal. The gate signals Pg of the signal sources 10A and 10B are in opposite phase to each other, so that the outputs of the LDMOS (11A) and LDMOS (11B) are in opposite phase.

[0053] An output transformer 12 is connected between the drain terminal of the LDMOS (11A) and the drain terminal of the LDMOS (11B), a DC power supply 13 is connected to the center tap of the output transformer 12, and the outputs of the LDMOS (11A) and LDMOS (11B) are in opposite phase to form a push-pull circuit 14. The push-pull circuit 14 alternately causes the secondary winding 12b of the output transformer 12 to source and sink current, thereby transmitting the load current, which is the output current.

[0054] A narrow-band filter 15 such as an LPF is provided between the secondary winding 12b of the output transformer 12 and the output terminal 16. The narrow-band filter 15 blocks the harmonic frequency n×f0 contained in the load current of the output current and passes the fundamental frequency f0 to the output terminal 16.

[0055] In the push-pull circuit 14, even-order harmonic frequency components cancel each other out, so the harmonic frequency n×f0 flowing through the secondary winding 12b becomes an odd-order harmonic frequency. Furthermore, the peak value of the harmonic frequency decreases as the order increases. Therefore, in designing the narrow-band filter 15, the harmonic frequency n×f0 can be sufficiently blocked by providing frequency characteristics that block the third-order harmonic frequency 3×f0.

[0056] The switching amplifier according to the present invention is characterized by comprising an LCR series resonator 20 as a configuration for suppressing the second surge voltage generated due to harmonic frequency components contained in the load current of the output current, and is characterized by comprising an LCR series resonator 20 and a partial resonant circuit 21 (21A, 21B) as a configuration for suppressing the first surge voltage generated due to high frequency components contained in the drain-source voltage Vds, which is the output voltage of the LDMOS, which is a switching element.

[0057] (LCR Series Resonator) The LCR series resonator 20 is composed of a series circuit of an inductance Lr, a capacitance Cr, and a resistance Rr, and is connected between the drain terminals of the LDMOS (11A) and the LDMOS (11B), and is also connected in parallel to the primary winding 12a of the output transformer 12.

[0058] The resonant frequency of the series circuit of the LCR series resonator 20 is set to match the frequency of the harmonic frequency components of the load current.

[0059] In a push-pull circuit, the harmonic frequency components contained in the load current are odd-order harmonic frequency components, and the peak values ​​of the fifth-order and higher harmonic frequency components are smaller than the peak value of the third-order harmonic frequency, so the resonant frequency of the series circuit is set to a frequency approximately three times the fundamental frequency f0 and its bandwidth [f0-min, f0-max], and further set to approximately two to four times the frequency taking into account the deviation of the resonant constant. Note that f0-min is the lower limit frequency of the load current, and f0-max is the upper limit frequency of the load current.

[0060] Resistor Rr, which is the resistance of the series circuit, is set to, for example, 0.5 Ω [ohm] to 15 Ω [ohm] so that sufficient attenuation can be obtained at the center frequency of the third harmonic frequency and attenuation can be obtained over a wide band within the bandwidth. The setting range of resistor Rr is such that the resistance value as seen from the primary side of the output transformer is approximately the same as the load impedance of the fundamental wave.

[0061] When the turn ratio of the output transformer is m:n, the primary resistance is R1, and the secondary resistance is R2, R1 = (m / n) ２ × R2. In this relationship, for example, when m = 1, n = 2, and R2 = 50 ohms, the primary resistance R1 is 12.5 ohms, so 15 ohms is set as the upper limit. Furthermore, as the fundamental frequency increases, the load impedance suitable for the LDMOS characteristics decreases, so 0.5 ohms is set as the lower limit.

[0062] The LCR series resonator 20 attenuates harmonic frequency components reflected by the narrowband filter and returned to the LDMOS, thereby maintaining a constant peak value of the drain-source voltage of the LDMOS, thereby maintaining a constant output gain characteristic even when the load current frequency fluctuates over a wide band, enabling stable output over a wide band.

[0063] (Partial Resonant Circuit) The partial resonant circuit 21 includes a partial resonant circuit 21A on the LDMOS (11A) side and a partial resonant circuit 21B on the LDMOS (11B) side.

[0064] The partial resonant circuit 21A is composed of a series circuit of an inductance component Ls-A on the drain end side of the LDMOS (11A) and a capacitance component CA of the LDMOS (11A).

[0065] The inductance component Ls-A is any one of a series inductance Ls1-A connected in series between the LDMOS (11A) and one end of the output transformer 12, a leakage inductance Lleak of the output transformer 12, and a wiring inductance Lline-A between the LDMOS (11A) and one end of the output transformer 12, or a combination of any of these. The series inductance Ls1-A is connected as necessary depending on the resonant frequency of the partial resonant circuit 21A.

[0066] The capacitance component CA of the LDMOS (11A) can be the parasitic capacitance Coss-A of the LDMOS (11A), or a combined capacitance Ctotal-A of the parasitic capacitance Coss-A and additional capacitance Cp-A connected in parallel between the drain and source terminals of the LDMOS (11A).

[0067] Similar to the partial resonant circuit 21A, the partial resonant circuit 21B is configured by a series circuit of a series inductance component Ls-B on the drain end side of the LDMOS (11B) and a capacitance component CB of the LDMOS (11B).

[0068] The inductance component Ls-B is any one of a series inductance Ls1-B connected in series between the LDMOS (11B) and the other end of the output transformer 12, a leakage inductance Lleak of the output transformer 12, and a wiring inductance Lline-B between the LDMOS (11B) and the other end of the output transformer 12, or a combination of any of these. The series inductance Ls1-B is connected as necessary depending on the resonant frequency of the partial resonant circuit 21B.

[0069] The capacitance component C-B of the LDMOS (11B) can be the parasitic capacitance Coss-B of the LDMOS (11B), or it can be a composite capacitance Ctotal-B of the additional capacitance Cp-B connected in parallel between the drain and source terminals of the LDMOS (11B) and the parasitic capacitance Coss-B. The additional capacitances Cp-A and Cp-B may be either lumped constants or distributed constants.

[0070] The resonant frequencies of the partial resonant circuits 21A and 21B are in the range from the fundamental frequency f0 to the third harmonic frequency 3×f0 of the high frequency output from the secondary side of the output transformer 12, and reduce the impedance in the frequency range where the impedance becomes high due to the partial resonant circuits, while setting a predetermined time constant for the rise and fall of the drain-source voltage Vds of the LDMOS.

[0071] The partial resonant circuit slopes the drain-source voltage Vds so that it forms a trapezoidal waveform with a predetermined time width at the rise and fall, thereby suppressing high-frequency components that occur at the rise and fall. At this time, the partial resonant circuits 21A and 21B partially resonate in the vicinity of a frequency range where the impedance between the drains becomes high, thereby reducing the impedance.

[0072] (Series resonator and partial resonant circuit) The series resonator and partial resonant circuit have different resonant frequencies that are suppressed by resonance. The resonant frequency of the series resonator is a frequency for reducing the harmonic frequency components contained in the high-frequency output to suppress surge voltages. On the other hand, the resonant frequency of the partial resonant circuit is a frequency for reducing impedance at frequencies that result in high impedance and suppressing surge voltages by making the drain-source voltage Vds a trapezoidal waveform with sloping rise and fall characteristics.

[0073] When the series resonator is resonating, the only impedance is the resistance Rr, and a parallel damping circuit is formed by this resistance Rr, the inductance component Ls of the partial resonant circuit, and the capacitance component C. The damping factor ζ of this parallel damping circuit is ζ = (1 / (2 × Rr)) × (Ls / C), and the smaller the resistance Rr, the greater the damping factor ζ and the greater the damping effect.

[0074] The resonant frequency of the series resonator is set to a harmonic frequency bandwidth [n×f0min to n×f0max] that corresponds to the variable frequency range [f0min to f0max] when the fundamental frequency f0 is varied. Therefore, when the frequency of the high-frequency output is changed, multiple resonant frequencies are set according to the changed frequency.

[0075] On the other hand, the resonant frequency of the partial resonant circuit is set so that the drain-source voltage Vds has a trapezoidal waveform with a slope in a frequency range near the frequency at which the impedance becomes high. The inductance component Ls and the capacitance component C are set according to the set resonant frequency. The frequency range in which the resonant frequency is set does not fluctuate significantly even if the frequency of the high-frequency output is changed. Therefore, even if the frequency of the high-frequency output is changed, there is no need to change the resonant frequency according to the changed frequency, and it is possible to set the resonant frequency at only one point.

[0076] (1-2: Second Configuration Example) The second configuration example of the switching amplifier 1, like the first configuration example, includes a switching unit 11, an output transformer 12, a DC power supply 13, a narrow-band filter 15, an LCR series resonator 20, and a partial resonant circuit 21 (21A, 21B).

[0077] In the switching amplifier 2 of the second configuration example, two LCR series resonators 20A and 20B are connected in parallel between the drain terminal of the LDMOS (11A) and the drain terminal of the LDMOS (11B) and to the primary winding 12a of the output transformer 12.

[0078] The first LCR series resonator 20A and the second LCR series resonator 20B have a series circuit similar to that of the LCR series resonator 20 of the first configuration example, and the first LCR series resonator 20A is composed of a series circuit of inductance Lr1, capacitance Cr1, and resistance Rr1, while the second LCR series resonator 20B is composed of a series circuit of inductance Lr2, capacitance Cr2, and resistance Rr2.

[0079] The first LCR series resonator 20A and the second LCR series resonator 20B are set to have different resonant frequencies of the series circuits, thereby widening the frequency band in which harmonic frequency components of the load current are attenuated.

[0080] The first LCR series resonator 20A has a resonant frequency lower than the harmonic frequency n×f0 of the high-frequency output, where f0 is the fundamental frequency of the high-frequency output from the secondary side of the transformer, and the second LCR series resonator 20B has a resonant frequency higher than the harmonic frequency n×f0 of the high-frequency output. By shifting the resonant frequencies of the LCR series resonators connected in parallel, the band of harmonic frequencies to be attenuated is widened.

[0081] Although two LCR series resonators are connected in parallel in this example, the number of LCR series resonators connected in parallel is not limited to two and may be any number, and the number of LCR series resonators connected in parallel and the frequency values ​​of each resonant frequency are set according to the bandwidth of the harmonic frequencies to be output. Suppression of harmonic frequencies by the LCR series resonator 20 will be described later.

[0082] (1-3: Comparison between LCR series resonators and LC series resonators) The surge voltage suppression effect of an LCR series resonator will be explained below in comparison with an LC series resonator. Fig. 3A shows the attenuation characteristics of a configuration in which two LC series resonators are connected in parallel. The attenuation characteristics of Fig. 3A show the characteristics of a configuration in which two LCR series resonators are connected in parallel, as represented by the equivalent circuit shown in Fig. 3B.

[0083] In Figure 3A, the solid line indicates the pass characteristic of the LCR series resonator, and the dashed line indicates the pass characteristic of the LC series resonator, both of which are set to resonate at the same frequency. In Figure 3A, the pass characteristic is shown as the attenuation rate versus frequency, with the attenuation rate increasing downward on the vertical axis.

[0084] Because the drain-source voltage Vds contains odd-order harmonics, the resonant frequencies of the LCR series resonator and the LC series resonator are designed to resonate at frequencies two to four times the fundamental frequency f0, including frequencies approximately three times its bandwidth (f0-min to f0-max). In Figure 3A, the attenuation frequency range is the frequency range in which the impedance of the series resonator appears to be 50 Ω [ohm] or less, and the attenuation rate at which the amplitude is halved is -6 dB or less.

[0085] By using two series resonators with different frequency ranges, the frequency range of the resonant frequency can be broadened. For this purpose, an LC series resonator is inappropriate, whereas an LCR series resonator is suitable.

[0086] According to the pass characteristics of the LC series resonator without resistor Rr, shown by the dashed line in Figure 3A, there is no attenuation effect at frequencies including third harmonics ((2 to 4) x f0). If there is a frequency range in which no attenuation effect is achieved, frequency components in that frequency range will appear in the drain-source voltage Vds, causing surge voltages. If the surge voltage exceeds the breakdown voltage of an element such as an LDMOS, it may result in the destruction of the element.

[0087] On the other hand, the pass characteristics of the LCR series resonator with resistor Rr shown by the solid line in Figure 3A show sufficient attenuation even at frequencies including the third harmonic ((2 to 4) x f0), and can provide attenuation effects over a wide frequency range.

[0088] (2) Example of Operation of Switching Amplifier An example of operation of the switching amplifier according to the present invention will be described with reference to Figures 4 to 8. Figure 4 shows signals at each part of the switching amplifier, and Figures 5 to 8 show examples of operation in each section of the switching amplifier.

[0089] Figure 4A shows the gate-source voltage VgsA of the LDMOS (11A), and Figure 4B shows the gate-source voltage VgsB of the LDMOS (11B). Figure 4C shows the drain-source voltage VdsA of the LDMOS (11A), and Figure 4D shows the drain-source voltage VdsB of the LDMOS (11B). Figure 4E shows the current IdsA flowing from the LDMOS (11A) to the primary winding 12a of the output transformer 12, and the current IdsB flowing from the LDMOS (11B) to the primary winding 12a of the output transformer 12. The current IdsA is shown by a solid line, and the current IdsB is shown by a dashed line.

[0090] The LDMOS (11A) and LDMOS (11B) switch between on and off with the gate-source voltages VgsA and VgsB at the threshold voltage as a boundary, and the gate-source voltages VgsA and VgsB switch in opposite phases to each other, thereby performing push-pull operation. In order to avoid a period in which the two LDMOSs (11A) and (11B) that make up the push-pull circuit are simultaneously in the on state, a dead time period is provided in which both gate-source voltages become 0 V when each gate-source voltage switches, and the two LDMOSs (11A) and (11B) are both in the off state.

[0091] Below, an example of the operation of the LDMOS (11A) and LDMOS (11B) will be explained by dividing them into eight sections, sections T1 to T8. Sections T1 and T5 are dead time sections in which both LDMOS (11A) and (11B) are in the off state. Sections T2, T3, and T4 are sections in which the LDMOS (11A) is in the off state and the LDMOS (11B) is in the on state. Meanwhile, sections T6, T7, and T8 are sections in which the LDMOS (11A) is in the on state and the LDMOS (11B) is in the off state.

[0092] Sections T1 and T2 are transition sections in which the drain-source voltage VdsA of LDMOS 11A increases as LDMOS 11A is turned off, and sections T5 and T6 are transition sections in which the drain-source voltage VdsB of LDMOS 11B increases as LDMOS 11B is turned off.

[0093] Sections T4 and T5 are transition sections in which the drain-source voltage VdsA of LDMOS 11A drops due to resonance of LDMOS 11A and turn-off of LDMOS 11B. Sections T8 and T1 are transition sections in which the drain-source voltage VdsB of LDMOS 11B drops due to resonance of LDMOS 11B and turn-off of LDMOS 11A.

[0094] (Section T1) Section T1 is a dead time section in which both the LDMOS 11A and the LDMOS 11B are in the OFF state. Note that section T1 is the state in which the LDMOS 11A is turned off.

[0095] The operating state of section T1 will be explained using Figure 5A. When the LDMOS (11A) is turned off, a current IdsA flows toward the LDMOS (11A) side in the partial resonant circuit 21A on the LDMOS (11A) side due to the energy of the primary winding 12a. The capacitance component CA of the partial resonant circuit 21A is charged by the current IdsA. As the capacitance component CA is charged, the drain-source voltage VdsA rises with a time constant τr determined by the LC constant of the capacitance component CA and the inductance component Ls-A.

[0096] The rise of the drain-source voltage VdsA can be delayed by causing the drain-source voltage VdsA to rise with the time constant τr. By delaying the rise of the drain-source voltage VdsA, high-frequency components generated by the steep rise of the drain-source voltage VdsA are reduced, and the surge voltage at the time of rise is suppressed.

[0097] At this time, the capacitance component CA of the partial resonant circuit 21A is connected in series with the capacitance component CB of the partial resonant circuit 21B. This series connection reduces the LC constant, and the rise time tr of the drain-source voltage VdsA becomes slightly shorter than the fall time tf.

[0098] On the other hand, the current IdsB flows from the LDMOS (11B) side to the partial resonant circuit 21B on the LDMOS (11B) side toward the DC power supply 13, due to the current IdsA of the partial resonant circuit 21A. The flow of this current IdsB discharges the charge stored in the capacitance component C-B of the partial resonant circuit 21B.

[0099] As the capacitance component CB is discharged, the drain-source voltage VdsB drops toward 0 V with a time constant τf determined by the LC constant of the capacitance component CA and the series inductance Ls1-B.

[0100] (Section T2) Section T2 is a section in which the LDMOS (11A) is in the OFF state and the LDMOS (11B) is in the ON state.

[0101] 5B, the operation state of section T2 will be described. The LDMOS (11A) is in the off state, and due to the resonance of the partial resonant circuit 21A on the LDMOS (11A) side, the charging voltage of the drain-source voltage VdsA increases with a time constant τr, and the current IdsA decreases.

[0102] As in section T1, in section T2, the drain-source voltage VdsA rises with a time constant τr, delaying the rise of the drain-source voltage VdsA. This reduces the high-frequency components that occur when the drain-source voltage VdsA rises sharply, suppressing surge voltages at the time of rise. Sections T1 and T2 cause the rising waveform of the drain-source voltage VdsA to become a trapezoidal waveform.

[0103] Since the LDMOS (11B) is on, the drain-source voltage VdsB is maintained at 0 V. The current IdsB of the partial resonant circuit 21B switches from the current direction in the section T1 and flows from the DC power supply 13 toward the ground side.

[0104] (Section T3) Section T3 is a section in which the LDMOS (11A) is in the OFF state and the LDMOS (11B) is in the ON state.

[0105] The operation state of section T3 will be described with reference to Figure 6A. Because LDMOS (11B) is in the ON state, current IdsB begins to flow toward LDMOS (11B), but because LDMOS (11A) is in the OFF state, the charge of capacitance component CA of LDMOS (11A) remains charged, and the peak value of drain-source voltage VdsA remains almost constant.

[0106] Thereafter, on the LDMOS (11A) side, the current IdsA is attracted by the current IdsB and flows from the LDMOS (11A) toward the DC power supply 13, and the drain-source voltage VdsA oscillates due to resonance by the partial resonant circuit 21A as well as resonance including the inductance component of the primary winding 12a of the output transformer 12. In the middle period of the section T3 accompanied by resonant oscillation, the peak value of the drain-source voltage VdsA is almost constant, and the surge voltage generated by a sudden change in the load current is suppressed by the partial resonance.

[0107] In the latter half of section T3, the charge on capacitance component CA of LDMOS (11A) changes to discharge, and the peak value of drain-source voltage VdsA tends to decrease. As current IdsB flows through primary winding 12a, load current Iload flows through secondary winding 12b of output transformer 12, accompanied by harmonic frequency components. Load current Iload passes through narrowband filter 15 and flows from output terminal 16 to load 17. Meanwhile, harmonic frequency components are reflected by narrowband filter 15, flow through output transformer 12 to LCR series resonator 20, and then flow to LDMOS (11B) in the ON state. At this time, the harmonic frequency components are reduced by resistance Rr of LCR series resonator 20, and surge voltages generated by the harmonic frequency components are suppressed.

[0108] (Section T4) Section T4 is a section in which the LDMOS (11A) is in the OFF state and the LDMOS (11B) is in the ON state, similar to section T3.

[0109] 6B, the operating state of section T4 will be described. In section T4, continuing from the latter half of section T3, the drain-source voltage VdsA shows a downward trend due to resonance including the inductance component of the primary winding 12a of the output transformer 12, and the surge voltage is suppressed by resonance of the partial resonant circuit 21A.

[0110] (Section T5) Section T5 is a dead time section in which both the LDMOS (11A) and the LDMOS (11B) are in the OFF state, and the LDMOS (11B) is turned off from the ON state.

[0111] 7A, the operation state of section T5 will be described. When the LDMOS (11B) is turned off, a current IdsB flows in the partial resonant circuit 21B on the LDMOS (11B) side due to the energy of the primary winding 12a toward the LDMOS (11B) side.

[0112] A current IdsA flows from the LDMOS (11A) side to the DC power supply 13 side in the partial resonant circuit 21A on the LDMOS (11A) side due to the current IdsB of the partial resonant circuit 21B. The flow of this current IdsA discharges the charge stored in the capacitance component C-A of the partial resonant circuit 21A. As the capacitance component C-A is discharged, the drain-source voltage VdsA drops toward 0 V with a time constant τf.

[0113] On the other hand, the capacitance component C-B of the partial resonant circuit 21B is charged by the current IdsB. As the capacitance component C-B is charged, the drain-source voltage VdsB rises with a time constant τr determined by the LC constant of the capacitance component C-B and the inductance component Ls-B.

[0114] The drain-source voltage VdsB rises with a time constant τr, which delays the rise of the drain-source voltage VdsB. By delaying the rise of the drain-source voltage VdsB, high-frequency components generated by the steep rise of the drain-source voltage VdsB are reduced, and the surge voltage at the rise time is suppressed.

[0115] At this time, the capacitance component C-B of the partial resonant circuit 21B is connected in series with the capacitance component C-A of the partial resonant circuit 21A. This series connection reduces the LC constant, and the rise time tr of the drain-source voltage VdsB becomes slightly shorter than the fall time tf.

[0116] (Section T6) Section T6 is a section in which the LDMOS (11A) is in the ON state and the LDMOS (11B) is in the OFF state.

[0117] The operating state in section T6 will be described using Figure 7B. The drain-source voltage VdsA is maintained at 0 V because the LDMOS (11A) is on. The current IdsA of the partial resonant circuit 21A switches from the current direction in section T5 and flows from the DC power supply 13 to the ground.

[0118] The LDMOS (11B) is in an OFF state, and due to the resonance of the partial resonant circuit 21B on the LDMOS (11B) side, the charging voltage of the drain-source voltage VdsB rises with a time constant τr, and the current IdsB decreases.

[0119] As in section T5, in section T6, the drain-source voltage VdsB rises with a time constant τr, delaying the rise of the drain-source voltage VdsB. This reduces the high-frequency components that occur when the drain-source voltage VdsB rises sharply, suppressing surge voltages at the time of rise. Sections T5 and T6 cause the rising waveform of the drain-source voltage VdsB to become a trapezoidal waveform.

[0120] (Section T7) Section T7 is a section in which the LDMOS (11A) is in the ON state and the LDMOS (11B) is in the OFF state.

[0121] The operation state of section T7 will be explained using Figure 8A. Because LDMOS 11A is in the ON state, current IdsA begins to flow toward LDMOS 11A, but because LDMOS 11B is in the OFF state, the charge of capacitance component C-B of LDMOS 11B remains charged, and the peak value of drain-source voltage VdsB remains almost constant.

[0122] Thereafter, on the LDMOS (11B) side, current IdsB is attracted by current IdsA and flows from LDMOS (11AB) toward DC power supply 13, and drain-source voltage VdsB oscillates due to resonance by partial resonant circuit 21B as well as resonance including the inductance component of primary winding 12a of output transformer 12. In the middle period of section T7 accompanied by resonant oscillation, the peak value of drain-source voltage VdsB is almost constant, and surge voltages generated in conjunction with sudden changes in load current are suppressed by partial resonance.

[0123] In the latter half of section T7, the charge on capacitance component C-B of LDMOS (11B) begins to discharge, and the peak value of drain-source voltage VdsB tends to decrease. As current IdsA flows through primary winding 12a, load current Iload flows through secondary winding 12b of output transformer 12, accompanied by harmonic frequency components. Load current Iload passes through narrowband filter 15 and flows from output terminal 16 to load 17. Meanwhile, harmonic frequency components are reflected by narrowband filter 15, flow through output transformer 12 to LCR series resonator 20, and then flow to LDMOS (11A) in the ON state. At this time, the harmonic frequency components are reduced by resistance Rr of LCR series resonator 20, and surge voltages generated by the harmonic frequency components are suppressed.

[0124] (Section T8) Section T8 is a section in which the LDMOS (11A) is in the ON state and the LDMOS (11B) is in the OFF state, similar to section T7.

[0125] 8B, the operating state of section T8 will be described. In section T8, continuing from the latter half of section T7, the drain-source voltage VdsB shows a downward trend due to resonance including the inductance component of the primary winding 12a of the output transformer 12, and the surge voltage is suppressed by resonance of the partial resonant circuit 21B.

[0126] (3) Trapezoidal Waveform The trapezoidal waveform of the drain-source voltage of the switching element according to the present invention will now be described. The trapezoidal waveform shown in Fig. 9 is the drain-source voltage VdsA of the LDMOS (11A) between sections T1 and T5, and shows two resonances due to the partial resonant circuit 21 and the LCR series resonator 20, and surge voltage suppression due to the two resonances.

[0127] (Suppression of First Surge Voltage) The partial resonant circuit 21 reduces high-frequency components contained in the load current in the sections T1 to T5 and the sections T6 to T8, and suppresses the first surge voltage generated by the high-frequency components.

[0128] The load current flows as a current IdsA in the partial resonant circuit 21A, and as a current IdsB in the partial resonant circuit 21B.

[0129] In sections T1 and T2, the current IdsA charges the capacitance component CA of the partial resonant circuit 21A in both sections T1 and T2, and the current IdsB discharges the capacitance component CB of the partial resonant circuit 21B in section T1.

[0130] In the section T3, the current IdsA changes the capacitance component CA of the partial resonant circuit 21A from a charged state to a discharged state.

[0131] In sections T4 and T5, the current IdsA discharges the capacitance component CA of the partial resonant circuit 21A in both sections T1 and T2, and the current IdsB charges the capacitance component CB of the partial resonant circuit 21B in section T5.

[0132] In the partial resonant circuit 21, the drain-source voltage Vds rises and falls in a trapezoidal shape with a rise time tr and a fall time tf due to charging and discharging in the sections T1, T2, T4, and T5, thereby reducing the high-frequency components contained in the current Ids and suppressing the first surge voltage generated by the high-frequency components. In the partial resonant circuit 21, the high-frequency components contained in the current Ids are reduced by the resistance of the load impedance in conjunction with the resonant operation in the section T3, thereby suppressing the first surge voltage generated by the high-frequency components.

[0133] (Suppression of Second Surge Voltage) In the section T3, the partial resonant circuit 21 reduces the harmonic frequency components contained in the load current, and suppresses the second surge voltage generated by the harmonic frequency components.

[0134] In section T3, a load current flows through the second side of the output transformer 12. The load current includes a frequency component of the fundamental frequency f0 and a frequency component of the harmonic frequency n×f0. By setting the resonant frequency of the LCR series resonator 20 to correspond to the harmonic frequency n×f0, the LCR series resonator 20 captures the frequency component of the harmonic frequency n×f0 reflected by the narrowband filter and dissipates and attenuates it through the resistor Rr. In this way, the LCR series resonator 20 suppresses the second surge voltage generated by the harmonic frequency component.

[0135] (4) Wide Bandwidth The following describes the wideband output of the switching amplifier according to the present invention. The wideband output of the switching amplifier in partial resonance and the wideband output of the LCR series resonator are shown below.

[0136] (Wide Band of Partial Resonance) Broadening of the output band by partial resonance will be described with reference to Fig. 10 and Fig. 11. Fig. 10 shows an example of a waveform when the fundamental frequency f0 is 45 MHz, and Fig. 11 shows an example of a waveform when the fundamental frequency f0 is 35 MHz.

[0137] 10A and 11A show currents IdsA and IdsB, FIGS. 10B and 11B show drain-source voltages VdsA and VdsB, FIGS. 10C and 11C show gate-source voltages VgsA and VgsB, and FIGS. 10D and 11D show voltages VR1 and VR2 across resistor Rr of the LCR DC resonator.

[0138] The resonant frequency of the partial resonant circuit is set in the range of 1 to 3 times the fundamental frequency f0, and is preferably designed to be about 1.5 to 2.5 times. When the resonant frequency of the partial resonant circuit is set in the above frequency range, even when the fundamental frequency f0 is varied, it is desirable that the overall loss, taking into account switching loss and the increased loss opportunity due to higher frequencies, be suitable for the output of the switching amplifier.

[0139] The waveform examples of FIG. 10 and FIG. 11 show examples in which the fundamental frequency f0 is set to 40 MHz and the frequency is varied between 35 MHz and 45 MHz.

[0140] When the frequency is 35 MHz, there is a slight voltage rise in the waveform of the residual drain-source voltage Vds just before the LDMOS switches from the off state to the on state due to partial resonance. However, although this voltage rise increases the switching loss in one cycle, the effect on the total loss is small because the frequency is low.

[0141] On the other hand, when the frequency is 45 MHz, the LDMOS gate-source voltage Vgs is applied when the drain-source voltage Vds is near its lowest point, so switching is performed by ZVS, reducing the switching loss per cycle. Although the frequency increases, the impact on overall loss is small because the switching loss is reduced. When the partial resonance frequency exceeds three times the fundamental frequency f0, the residual drain-source voltage Vds increases further, resulting in a decrease in overall efficiency.

[0142] (Wide Band of LCR Series Resonator) Broadening of the output band by an LCR series resonator will be described with reference to Fig. 12. Fig. 12 shows a resonance curve expressed by admittance or current.

[0143] FIG. 12A shows the resonance curve when the LCR series resonator is configured as one series circuit, and FIG. 12B shows the resonance curve when the LCR series resonator is configured as two series circuits connected in parallel.

[0144] The curve on the left in FIG. 12A represents the passband of the variable frequency band, with the lower limit frequency f0-min and upper limit frequency f0-max for the fundamental frequency f0 being set by frequencies close to the maximum value of 1 on the resonance curve.

[0145] 12A is a resonance curve of a series circuit in the harmonic frequency band when the LCR series resonator is configured as a single series circuit, with respect to the pass band of the variable frequency band. The resonance curve of this series circuit in the harmonic frequency band is set so that the lower limit frequency n×f0-min and the upper limit frequency n×f0-max for the harmonic frequency n×f0 (n is an integer) are set within a frequency range in which the series resonator of the series circuit in the harmonic frequency band is 50 Ω [ohm] or less (standard 1) and the current flowing through the series resonator is 1 or more.

[0146] The lower limit frequency n×f0-min and upper limit frequency n×f0-max set the frequency range of the harmonic frequency band. This frequency range is set, for example, to ±1.5% or more of the harmonic frequency n×f0, for a total of 3% or more. By setting this harmonic frequency band, even if the frequency changes within the variable frequency band, the admittance or current will be more than 1 times the standard of 50 Ω [ohm], and high loss (attenuation) can be expected, allowing for a wide frequency range.

[0147] The curve on the left side of Figure 12B represents the passband of the variable frequency band, similar to the curve on the left side of Figure 12A, and the lower limit frequency f0-min and upper limit frequency f0-max for the fundamental frequency f0 are set by frequencies that are close to the maximum value of 1 on the resonance curve.

[0148] In contrast to the resonance curves for the variable frequency band, the two curves on the right side of Fig. 12B are resonance curves for a series circuit in the harmonic frequency band when two series circuits of LCR series resonators are configured. These resonance curves for the series circuit in the harmonic frequency band are configured by using the two resonance curves for the series circuit in the harmonic frequency band on the right side of Fig. 12A and shifting the center frequencies f1 and f2.

[0149] The resonance curve of the series circuit in one harmonic frequency band is set to a center frequency f1 shifted to the lower frequency side, and the resonance curve of the series circuit in the other harmonic frequency band is set to a center frequency f2 shifted to the higher frequency side, and the two resonance curves are set at a frequency at which the current flowing through the series resonator in the harmonic frequency band, which is a combination of the lower limit frequency f0-min and the upper limit frequency f0-max for the harmonic frequency n×f0 (n is an integer), is equal to or greater than 1.

[0150] The lower limit frequency n×f0-min and upper limit frequency n×f0-max set as the frequency range of the harmonic frequency band are set, for example, to ±3% or more of the harmonic frequency n×f0, with a total of 6% or more. By using two series circuits, the frequency range of the harmonic frequency band can be widened. By setting this harmonic frequency band, even if the frequency changes within the variable frequency band, the admittance or current will be more than 1 times the standard of 50 Ω [ohm], and high loss (attenuation) can be expected, allowing for a wide frequency range.

[0151] (Partial Resonance Frequency Band and High Impedance Frequency Band) The relationship between the partial resonance frequency band and the high impedance frequency band will be described with reference to Fig. 13. Fig. 13A shows a Smith chart, and Fig. 13B shows a frequency characteristic diagram of the attenuation factor.

[0152] The Smith chart in Fig. 13A shows a constant SWR circle S that represents the reflection state with the load. When the frequency is swept, the impedance is represented by a locus on the constant SWR circle S that progresses clockwise from the low frequency side to the high frequency side. In the constant SWR circle S shown in Fig. 13A, in the frequency range from the fundamental frequency f0 to the harmonic frequency nxf0, a high impedance band fb where the impedance is high exists between the fundamental frequency f0 and the harmonic frequency nxf0.

[0153] Therefore, a switching amplifier that has a variable frequency output range that is from the fundamental frequency f0 to the harmonic frequency n×f0 has two frequencies and frequency ranges where the impedance is high: the high impedance band fb and the harmonic frequency n×f0. High impedance can cause surge voltages.

[0154] In order to suppress the surge voltage in the high impedance band fb, the impedance in the frequency range near this high impedance band fb is reduced. In the present invention, the partial resonant frequency band FB2 is set by the partial resonant circuit to reduce the impedance and suppress the surge voltage in the high impedance band fb.

[0155] In order to suppress surge voltages in a frequency range centered on the harmonic frequency n×f0, the impedance in this frequency range is reduced. In the present invention, the LCR series resonator band FB3 is set using an LCR series resonator to reduce impedance and suppress surge voltages in the frequency range centered on the harmonic frequency n×f0.

[0156] 13B shows the attenuation factor vs. frequency characteristics of the Smith chart of FIG. 13A. The partial resonant frequency band FB2 corresponds to the high-impedance band fb, and the LCR series resonator band FB3 corresponds to a frequency range centered on the harmonic frequency n×f0. By setting the attenuation factors in the partial resonant frequency band FB2 and the LCR series resonator band FB3 high, surge voltages can be suppressed.

[0157] The switching amplifier according to the present invention can be applied to a high frequency power source (RF generator) used in semiconductor manufacturing equipment, liquid crystal panel manufacturing equipment, and the like.

[0158] 1, 2 Switching amplifier 10A Signal source 10B Signal source 11 Switching section 12 Output transformer 12a Primary winding 12b Secondary winding 13 DC power supply 14 Push-pull circuit 15 Narrow-band filter 16 Output end 17 Load 20, 20a, 20b LCR series resonator 21, 21A, 21B Partial resonant circuit 101a, 101b Switching amplifier 110A, 110B Signal source 111 Switching section 111A, 111B Switching element 112 Output transformer 112a Primary winding 112b Secondary winding 113 DC power supply 114 Push-pull circuit 115 Narrow-band filter 116 Output end 117 Load C, C-A, C-B Capacitance component Coss-A, Coss-B Parasitic capacitance Cp-A, Cp-B Additional capacitance Cr, Cr1, Cr2 Capacitance Ctotal-A, Ctotal-B Total capacitance FB2 Partial resonance frequency band FB3 LCR series resonance band Ids, IdsA, IdsB Current Iload Load current Lline-A, Lline-B Wiring inductance Lr, Lr1, Lr2 Inductance Lleak Leakage inductance Ls, Ls-A, Ls-B Inductance component Ls1-A, Ls1-B Series inductance Pg Gate signal R1 Primary side resistance component Rr, Rr1, Rr2 Resistance S Constant SWR circle T1, T2, T3, T4, T5, T6, T7, T8 Section Vds, VdsA, VdsB Drain-source voltage Vgs, VgsA, VgsB Gate-source voltage f0 Fundamental frequency f0-max Upper limit frequency f0-min Lower limit frequency f1, f2 Center frequency fb High impedance band τf, τr Time constant

Claims

1. A switching amplifier comprising: a transformer-coupled push-pull circuit composed of a pair of switching elements which alternately perform on / off switching operations in response to input signals of opposite phases, a primary winding to which the output signals of the switching elements are input, a secondary winding to which output signals of opposite phases are taken, and an output transformer having an external power supply connected to a center tap of the primary winding; a narrow-band filter connected to the output end of the secondary winding of the output transformer; a pair of partial resonant circuits composed of an inductance component on the drain side of each of the switching elements and a capacitance component between the drain and source of each of the switching elements; and an LCR series resonator composed of a series circuit of inductance, capacitance and resistance connected between the drain ends of each of the switching elements.

2. A switching amplifier as described in claim 1, wherein the inductance component of each of the partial resonant circuits is either a series inductance connected in series between each switching element and one end of the output transformer, a leakage inductance of the output transformer, a wiring inductance between each switching element and one end of the output transformer, or a combination of any of these, and the capacitance component of each of the partial resonant circuits is an output parasitic capacitance of the switching element, or a combined capacitance of the output parasitic capacitance of the switching element and an additional capacitance connected in parallel to the switching element.

3. A switching amplifier as described in claim 1, wherein the resonant frequency of each of the partial resonant circuits is set so as to reduce the impedance in the frequency range in which the impedance becomes high due to the partial resonant circuit within the range from the fundamental frequency f0 to the third harmonic frequency (3 x f0) of the high frequency output output from the secondary side of the output transformer, and is a frequency at which the time constants of the rise and fall of the drain-source voltage of the switching element are set as setting values.

4. The switching amplifier of claim 1, wherein, relative to a fundamental frequency f0 of the high frequency output output from the secondary side of the output transformer, the resonant frequency of the LCR series resonator is a harmonic frequency n×f0 (n is an integer), and the bandwidth of the resonant frequency is set to a frequency range of (n×f0-min to n×f0-max) according to a variable frequency range (f0-min to f0-max) of the fundamental frequency f0.

5. The switching amplifier according to claim 1, comprising a plurality of the LCR series resonators having different resonant frequencies connected in parallel.

6. The switching amplifier according to claim 5, comprising: a first LCR series resonator having a resonant frequency on the lower frequency side than a harmonic frequency n×f0 of the high frequency output, with respect to a fundamental frequency f0 of the high frequency output output from the secondary side of the output transformer; and a second LCR series resonator having a resonant frequency on the higher frequency side than the harmonic frequency n×f0 of the high frequency output.

7. The switching amplifier according to any one of claims 1 to 6, wherein the switching element is a DMOS.

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