Integrated transformer capable of utilizing leakage inductance, and three-port on-board charger

Abstract

Disclosed in the present invention are an integrated transformer capable of utilizing leakage inductance, and a three-port on-board charger. The integrated transformer comprises a first C-shaped magnetic core and a second C-shaped magnetic core, wherein two ends of the first C-shaped magnetic core are respectively butt-jointed to two ends of the second C-shaped magnetic core to form a closed magnetic ring, a first winding and a third winding are wound round the first C-shaped magnetic core, and a second winding is wound round the second C-shaped magnetic core. The integrated transformer further comprises a bypass magnetic core, wherein the bypass magnetic core is arranged in the middle of the magnetic ring, and divides the interior of the magnetic ring into a first cavity and a second cavity. The present invention reuses a bypass magnetic core, and controls the coupling between transformer windings to obtain leakage inductance as resonant inductance; and four power magnetic elements in the prior art are integrated into a transformer, and the transformer is then applied to an on-board charger. Thus, the number of magnetic devices of a converter is reduced, and thus the costs are reduced; the power density of the converter is significantly improved; the overall loss of the converter is reduced, and the efficiency is improved; and the costs of the magnetic devices of the converter are reduced.

Description

Integrated transformer and three-port on-board charger that can utilize leakage inductance Technical Field

[0001] The present invention relates to a transformer, and in particular to an integrated transformer which can utilize leakage inductance and integrate multiple coils, and a three-port vehicle-mounted charger. Background Art

[0002] In existing on-board charger technology, the DC conversion circuit is generally divided into two stages. The front-stage circuit is a bidirectional high-voltage DC conversion circuit, whose output is connected to the on-board power battery and provides a high-voltage DC charging current. This circuit topology is generally an LLC resonant circuit, in which the resonant inductor and the main transformer are independent of each other and are used in the circuit as two discrete components. The rear-stage circuit is a low-voltage DC conversion circuit, whose input is connected in parallel with the on-board power battery to the output of the front-stage bidirectional high-voltage DC conversion circuit, and the output is connected to the low-voltage battery and other low-voltage loads. Its circuit topology is also often an LLC resonant circuit, in which the transformer and resonant inductor are independent of each other and are used as two discrete components. Referring to the circuit diagram of the existing on-board charger transformer shown in Figure 1, the two-stage DC conversion circuit has a total of two resonant inductors, two transformers, and a total of four power magnetic devices. For on-board integrated charging products, there are the following problems: 1. The total loss of multiple power magnetic devices is large, which is not conducive to improving product efficiency; 2. The multiple power magnetic devices generate severe heat overall, and the heat dissipation cost is high; 3. The total volume of multiple magnetic components is large, which is not conducive to the high power density and miniaturization development of the product; 4. The total cost of multiple magnetic components is high.

[0003] Although some transformers in existing technical solutions mention the use of leakage inductance as a resonant inductor, the following problems still exist: 1. The design and control of leakage inductance are relatively difficult; 2. In multi-level output applications such as in-vehicle charging products, the front- and rear-stage DC conversion circuits still require the use of more than two power magnetic devices, which results in relatively high power consumption and cost. Summary of the Invention

[0004] In order to solve the above-mentioned defects in the prior art, the present invention proposes an integrated transformer and a three-port on-board charger that can utilize leakage inductance.

[0005] The technical solution adopted by the present invention is: designing an integrated transformer that can utilize leakage inductance, which includes a first C-shaped magnetic core and a second C-shaped magnetic core, the two ends of the first C-shaped magnetic core and the two ends of the second C-shaped magnetic core are connected in pairs to form a closed magnetic ring, the first C-shaped magnetic core is wound with a first winding and a third winding, and the second C-shaped magnetic core is wound with a second winding; the integrated transformer also includes a bypass magnetic core, which is arranged in the middle of the magnetic ring and divides the interior of the magnetic ring into a first chamber and a second chamber.

[0006] Optionally, a magnetic flux regulating block is provided at least at one of the two joints between the first C-shaped magnetic core and the second C-shaped magnetic core, and the magnetic flux regulating block is clamped between the ends of the first C-shaped magnetic core and the second C-shaped magnetic core.

[0007] In an optional solution, a flux regulating block is provided at both joints of the first C-shaped magnetic core and the second C-shaped magnetic core, and the flux regulating block is clamped between the ends of the first C-shaped magnetic core and the second C-shaped magnetic core; one end of the bypass magnetic core is connected to one end of the first C-shaped magnetic core, and the other end of the bypass magnetic core is connected to the other end of the first C-shaped magnetic core.

[0008] In another optional solution, a flux regulating block is provided at both joints of the first C-shaped magnetic core and the second C-shaped magnetic core, and the flux regulating block is clamped between the ends of the first C-shaped magnetic core and the second C-shaped magnetic core; the bypass magnetic core is connected between the two flux regulating blocks.

[0009] In another optional scheme, a flux adjustment block is provided at both joints of the first C-shaped magnetic core and the second C-shaped magnetic core, and the flux adjustment block is clamped between the ends of the first C-shaped magnetic core and the second C-shaped magnetic core; wherein, one end of the bypass magnetic core is connected to one end of the second C-shaped magnetic core, and the other end of the bypass magnetic core is connected to the other end of the second C-shaped magnetic core; or, one end of the bypass magnetic core is connected to one end of the first C-shaped magnetic core, and the other end of the bypass magnetic core is connected to the flux adjustment block.

[0010] In an optional solution, the third winding includes a first strand of the third winding and a second strand of the third winding. The first winding is wound tightly against the first C-shaped magnetic core, the first strand of the third winding is wound on the outside of the first winding, and the second strand of the third winding is wound on the outside of the first strand of the third winding.

[0011] In another optional solution, the third winding includes a first strand of the third winding and a second strand of the third winding, and the first winding, the first strand of the third winding and the second strand of the third winding are all wound closely around the first C-shaped magnetic core.

[0012] Optionally, the bypass magnetic core includes multiple magnetic sheets, and air gaps are provided between connected magnetic sheets.

[0013] In one optional solution, the second winding is entirely wound around the second C-shaped magnetic core. The magnetic path length, cross-sectional area, and magnetic permeability of the first C-shaped magnetic core, the second C-shaped magnetic core, the bypass magnetic core, and the magnetic flux regulating block are adjusted to change the magnetic resistance of the magnetic flux regulating block and the bypass magnetic core, thereby controlling the excitation inductance of the first winding and the leakage inductance of the first winding to the second winding (W2).

[0014] The cross-sectional area of ​​the first C-shaped magnetic core is calculated according to Formula A1: Ae1 = V1*T1 / (2*N1*B1), where Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core, V1 is the voltage applied to the connection port of the first winding, T1 is the switch circumference of the power switch in the DC conversion circuit connected to the first winding, N1 is the number of turns of the first winding, and B1 is the magnetic flux density of the first C-shaped magnetic core;

[0015] Calculate the cross-sectional area of ​​the second C-shaped magnetic core according to Formula A2: Ae2 = V2*T2 / (2*N2*B2), where Ae2 is the cross-sectional area of ​​the second C-shaped magnetic core, V2 is the voltage applied to the connection port of the second winding, T2 is the switch circumference of the power switch in the DC conversion circuit connected to the second winding, N2 is the number of turns of the first winding, and B2 is the magnetic flux density of the second C-shaped magnetic core;

[0016] The maximum number of turns of the third winding W3 is calculated according to formula A3, which is: N3=V3*T3*D / (Ae1*B1), where N3 is the maximum number of turns of the third winding W3, V3 is the voltage applied to the connection port of the third winding W3, T3 is the switch circumference of the power switch in the DC conversion circuit connected to the third winding W3, D is the duty cycle of the voltage waveform at the connection port of the third winding W3, Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core 1, and B1 is the magnetic flux density of the first C-shaped magnetic core 1;

[0017] The magnetic resistance of the first C-shaped magnetic core is calculated according to Formula A4, which is: Rm1 = Le1 / (u0*ur1*Ae1), where Rm1 is the magnetic resistance of the first C-shaped magnetic core, Le1 is the magnetic path length of the first C-shaped magnetic core, u0 is the vacuum permeability, ur1 is the relative permeability of the magnetic material of the first C-shaped magnetic core, and Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core;

[0018] The magnetic resistance of the second C-shaped magnetic core is calculated according to Formula A5: Rm2 = Le2 / (u0*ur2*Ae2), where Rm2 is the magnetic resistance of the second C-shaped magnetic core, Le2 is the magnetic path length of the second C-shaped magnetic core, u0 is the vacuum permeability, ur2 is the relative permeability of the magnetic material of the second C-shaped magnetic core, and Ae2 is the cross-sectional area of ​​the second C-shaped magnetic core.

[0019] The magnetic resistance of the two flux regulating blocks is calculated according to formula A6, which is: Rm3 = Le3 / (u0*ur3*Ae3), where Rm3 is the magnetic resistance of the two flux regulating blocks, Le3 is the magnetic path length of the two flux regulating blocks, u0 is the vacuum permeability, ur3 is the relative permeability of the magnetic material of the flux regulating block, and Ae3 is the cross-sectional area of ​​the flux regulating block;

[0020] The magnetic resistance of the bypass core is calculated according to formula A7, which is: Rm4 = Le4 / (u0*ur4*Ae4), where Rm4 is the magnetic resistance of the bypass core, Le4 is the magnetic path length of the bypass core, u0 is the vacuum permeability, ur4 is the relative permeability of the magnetic material of the bypass core, and Ae4 is the cross-sectional area of ​​the bypass core;

[0021] The total magnetic resistance of the first winding is calculated according to Formula A8, which is: Rm = Rm1 + (Rm2 + Rm3*2) * Rm4 / (Rm2 + Rm3*2 + Rm4), where Rm is the total magnetic resistance of the first winding, Rm1 is the magnetic resistance of the first C-shaped magnetic core, Rm2 is the magnetic resistance of the second C-shaped magnetic core, Rm3 is the magnetic resistance of the flux regulating block, and Rm4 is the magnetic resistance of the bypass magnetic core;

[0022] Calculate the magnetizing inductance of the first winding according to Formula A9: Lm1 = N1² / Rm, where Lm1 is the magnetizing inductance of the first winding, N1 is the number of turns of the first winding, and Rm is the total magnetic resistance of the first winding;

[0023] Calculate the coupling coefficient between the first winding and the second winding W2 according to Formula A10: k12 = Rm4 / (Rm2+Rm3*2+Rm4), where k12 is the coupling coefficient between the first winding and the second winding, Rm4 is the magnetic resistance of the bypass magnetic core, Rm2 is the magnetic resistance of the second C-shaped magnetic core, and Rm3 is the magnetic resistance of the flux regulation block;

[0024] The leakage inductance of the first winding with respect to the second winding is calculated according to formula A11. Formula A11 is: L12 = Lm1 * (1–k12²), where L12 is the leakage inductance of the first winding with respect to the second winding, Lm1 is the magnetizing inductance of the first winding, and k12 is the coupling coefficient between the first winding and the second winding.

[0025] In another optional solution, the second winding includes a first strand of the second winding and a second strand of the second winding, the first strand of the second winding being wound around the second C-shaped magnetic core, and the second strand of the second winding being wound around the second C-shaped magnetic core and the bypass magnetic core. The magnetic resistance of the flux regulating block and the bypass magnetic core is changed by adjusting the magnetic path length, cross-sectional area, and magnetic permeability of the first C-shaped magnetic core, the second C-shaped magnetic core, the bypass magnetic core, and the flux regulating block to control the excitation inductance of the first winding and the leakage inductance of the first winding to the second winding.

[0026] Calculate the cross-sectional area of ​​the first C-shaped magnetic core according to formula D1: Ae1 = V1*T1 / (2*N1*B1), where Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core, V1 is the voltage applied to the connection port of the first winding, T1 is the switch circumference of the power switch in the DC conversion circuit connected to the first winding, N1 is the number of turns of the first winding, and B1 is the magnetic flux density of the first C-shaped magnetic core;

[0027] The cross-sectional area of ​​the second C-shaped magnetic core is calculated according to formula D2: Ae2 = K12 * Ae1, where Ae2 is the cross-sectional area of ​​the second C-shaped magnetic core, k12 is the coupling coefficient between the first winding and the second winding, and Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core;

[0028] The maximum number of turns of the third winding is calculated according to formula D3, which is: N3 = V3*T3 / (2*Ae1*B1), where N3 is the maximum number of turns of the third winding, V3 is the voltage applied to the connection port of the third winding, T3 is the switch circumference of the power switch in the DC conversion circuit connected to the third winding, Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core, and B1 is the magnetic flux density of the first C-shaped magnetic core;

[0029] The magnetic reluctance of the first C-shaped magnetic core is calculated according to formula D4: Rm1 = Le1 / (u0*ur1*Ae1), where Rm1 is the magnetic reluctance of the first C-shaped magnetic core, Le1 is the magnetic path length of the first C-shaped magnetic core, u0 is the vacuum permeability, ur1 is the relative permeability of the magnetic material of the first C-shaped magnetic core, and Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core.

[0030] The magnetic resistance of the second C-shaped magnetic core is calculated according to formula D5, where formula D5 is: Rm2 = Le2 / (u0*ur2*Ae2), where Rm2 is the magnetic resistance of the second C-shaped magnetic core, Le2 is the magnetic path length of the second C-shaped magnetic core, u0 is the vacuum permeability, ur2 is the relative permeability of the magnetic material of the second C-shaped magnetic core, and Ae2 is the cross-sectional area of ​​the second C-shaped magnetic core;

[0031] The magnetic resistance of the two magnetic flux regulating blocks is calculated according to formula D6, and the formula D6 is: Rm3 = Le3 / (u0*ur3*Ae3), wherein Rm3 is the magnetic resistance of the two magnetic flux regulating blocks, Le3 is the magnetic path length of the two magnetic flux regulating blocks, u0 is the vacuum permeability, ur3 is the relative permeability of the magnetic material of the magnetic flux regulating block, and Ae3 is the cross-sectional area of ​​the magnetic flux regulating block;

[0032] The magnetic resistance of the bypass core is calculated according to formula D7, which is: Rm4 = Le4 / (u0*ur4*Ae4), where Rm4 is the magnetic resistance of the bypass core, Le4 is the magnetic path length of the bypass core, u0 is the vacuum permeability, ur4 is the relative permeability of the magnetic material of the bypass core, and Ae4 is the cross-sectional area of ​​the bypass core;

[0033] The total magnetic resistance of the first winding is calculated according to Formula D8, which is: Rm = Rm1 + (Rm2 + Rm3*2) * Rm4 / (Rm2 + Rm3*2 + Rm4), where Rm is the total magnetic resistance of the first winding, Rm1 is the magnetic resistance of the first C-shaped magnetic core, Rm2 is the magnetic resistance of the second C-shaped magnetic core, Rm3 is the magnetic resistance of the flux regulation block, and Rm4 is the magnetic resistance of the bypass magnetic core;

[0034] Calculate the magnetizing inductance of the first winding according to Formula D9: Lm1 = N1² / Rm, where Lm1 is the magnetizing inductance of the first winding, N1 is the number of turns of the first winding, and Rm is the total magnetic resistance of the first winding;

[0035] Calculate the coupling coefficient between the first winding and the first strand of the second winding according to Formula D10: k122 = Rm4 / (Rm2+Rm3*2+Rm4), where k122 is the coupling coefficient between the first winding and the first strand of the second winding, Rm4 is the magnetic resistance of the bypass magnetic core, Rm2 is the magnetic resistance of the second C-shaped magnetic core, and Rm3 is the magnetic resistance of the flux regulation block.

[0036] Calculate the inductance of the first winding according to formula D11: AL1 = Lm1 / (N1²), where AL1 is the inductance of the first winding, Lm1 is the magnetizing inductance of the first winding, and N1 is the number of turns of the first winding;

[0037] The leakage inductance of the first winding to the second winding is calculated according to formula D12. The formula D12 is: Lk12 =AL1*(N1*(1–N21 / N2))²*(1-K122²), where Lk12 is the leakage inductance of the first winding to the second winding, AL1 is the inductance of the first winding, N1 is the number of turns of the first winding, N21 is the number of turns of the second strand of the second winding, N2 is the number of turns of the second winding, and k122 is the coupling coefficient between the first winding and the first strand of the second winding.

[0038] The present invention also designs a three-port on-board charger suitable for an on-board charger. In addition to the integrated transformer with the above-mentioned leakage inductance, the three-port on-board charger also includes: a bidirectional high-voltage DC conversion circuit connected to the first winding, a primary-side conversion circuit connected to the second winding, and a low-voltage DC conversion circuit connected to the third winding; wherein the magnetic path length, cross-sectional area, and magnetic permeability of the first C-shaped magnetic core, the second C-shaped magnetic core, the bypass magnetic core, and the magnetic flux adjustment block are used to change the magnetic resistance of the magnetic flux adjustment block and the magnetic resistance of the bypass magnetic core to control the excitation inductance of the first winding and the leakage inductance of the first winding to the second winding.

[0039] The beneficial effects of the technical solution provided by the present invention are:

[0040] The present invention reuses a magnetic core and controls the coupling between the transformer windings to obtain leakage inductance as the resonant inductance, integrating the four power magnetic components in the prior art into an integrated transformer for application in a three-port power supply circuit. It has the following advantages: 1. The number of magnetic components in the converter is reduced, thereby lowering the cost; 2. The power density of the converter is greatly improved; 3. The overall loss of the converter is reduced, thereby improving the efficiency; 4. The cost of the magnetic components of the converter is reduced. BRIEF DESCRIPTION OF THE DRAWINGS

[0041] The present invention is described in detail below with reference to the embodiments and accompanying drawings, in which:

[0042] Figure 1 is a circuit diagram of a conventional vehicle-mounted charger transformer;

[0043] FIG2 is a schematic diagram of the structure of an integrated transformer according to the present invention;

[0044] FIG3 is a schematic diagram of the integrated transformer winding and magnetic circuit of the present invention;

[0045] FIG4 is a schematic diagram of an integrated transformer having only one magnetic sheet according to the present invention;

[0046] FIG5 is a schematic diagram of an integrated transformer in which a bypass core is connected between two magnetic sheets according to the present invention;

[0047] FIG6 is a schematic diagram of an integrated transformer in which a magnetic sheet of the present invention has multiple air gaps;

[0048] FIG7 is a schematic diagram of an integrated transformer in which the second winding of the present invention is divided into two strands;

[0049] FIG8 is a functional block diagram of a vehicle-mounted charger according to the present invention;

[0050] FIG9 is a circuit diagram of a vehicle-mounted charger according to the present invention;

[0051] 10 is a timing diagram of the drive signal when port A of the on-board charger of the present invention transmits energy to ports B and C;

[0052] 11 is a timing diagram of the drive signal when port B of the on-board charger of the present invention transmits energy to ports A and C;

[0053] Explanation of reference numerals: first C-shaped magnetic core (1), second C-shaped magnetic core (2), bypass magnetic core (3), magnetic sheet (3a), air gap (3b), first chamber (4), second chamber (5), magnetic flux regulating block (6), first winding (W1), second winding (W2), first strand of second winding (W2a), second strand of second winding (W2b), third winding (W3), first strand of third winding (W3a), second strand of third winding (W3b). DETAILED DESCRIPTION

[0054] In order to make the purpose, technical solutions and advantages of the present invention more clearly understood, the present invention will be further described in detail below in conjunction with the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are only used to explain the present invention and are not intended to limit the present invention.

[0055] The present invention discloses an integrated transformer that can utilize leakage inductance. The present invention reuses a magnetic core and controls the coupling between the transformer windings to obtain leakage inductance as a resonant inductance, integrating four power magnetic components into a three-port on-board charger. Referring to an embodiment shown in Figure 3, the integrated transformer includes a first C-shaped magnetic core 1 and a second C-shaped magnetic core 2. The two ends of the first C-shaped magnetic core 1 and the two ends of the second C-shaped magnetic core 2 are connected in pairs to form a closed magnetic ring. The first C-shaped magnetic core 1 is wound with a first winding W1 and a third winding W3, and the second C-shaped magnetic core 2 is wound with a second winding W2. The integrated transformer also includes a bypass magnetic core 3, which is arranged in the middle of the magnetic ring and divides the interior of the magnetic ring into a first chamber 4 and a second chamber 5.

[0056] The first winding W1 and the third winding W3 pass through the first cavity 4 on the left, while the second winding W2 passes through the second cavity 5 on the right. The first C-shaped magnetic core 1 and the second C-shaped magnetic core 2 form the main magnetic circuit, and the bypass magnetic core 3 forms the bypass magnetic circuit. By adjusting the magnetic circuit length, cross-sectional area, and magnetic permeability of the first C-shaped magnetic core 1, the second C-shaped magnetic core 2, the bypass magnetic core 3, and the magnetic flux adjustment block 6, the magnetic resistance of the magnetic flux adjustment block 6 and the magnetic resistance of the bypass magnetic core 3 are changed to control the excitation inductance of the first winding W1 and the leakage inductance of the first winding W1 to the second winding W2. This serves as the resonant inductance of the resonant power supply circuit, achieving the purpose of eliminating discrete inductance, as shown in the resonant inductor Lr in Figure 9.

[0057] To achieve the purpose of regulating magnetic flux, a magnetic flux regulating block 6 is provided at least at one of the two joints between the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2. The magnetic flux regulating block 6 is sandwiched between the ends of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2. Figure 4 shows a schematic diagram of an integrated transformer with only one magnetic flux regulating block 6.

[0058] Referring to another optional embodiment shown in Figure 2, a flux adjustment block 6 is provided at both joints of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2, and the flux adjustment block 6 is clamped between the ends of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2; one end of the bypass magnetic core 3 is connected to one end of the first C-shaped magnetic core 1, and the other end of the bypass magnetic core 3 is connected to the other end of the first C-shaped magnetic core 1.

[0059] In an embodiment shown in Figure 5, a flux regulating block 6 is provided at both joints of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2, and the flux regulating block 6 is clamped between the ends of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2; the bypass magnetic core 3 is connected between the two flux regulating blocks 6.

[0060] In other embodiments, a flux regulating block 6 is provided at both joints of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2, and the flux regulating block 6 is sandwiched between the ends of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2. Alternatively, one end of the bypass magnetic core 3 is connected to one end of the second C-shaped magnetic core 2, and the other end of the bypass magnetic core 3 is connected to the other end of the second C-shaped magnetic core 2. Alternatively, one end of the bypass magnetic core 3 is connected to one end of the first C-shaped magnetic core 1, and the other end of the bypass magnetic core 3 is connected to the flux regulating block 6. It should be noted that the above embodiments do not include drawings.

[0061] A complete tertiary winding W3 can be adapted for a full-bridge rectifier circuit. To connect to a full-wave rectifier circuit, the tertiary winding W3 needs to be divided into two winding groups with taps in the middle, and two winding methods are differentiated. Referring to one winding method shown in FIG2 , the tertiary winding W3 includes a first tertiary winding strand W3a and a second tertiary winding strand W3b. The first winding W1 is wound closely to the first C-shaped magnetic core 1, the first tertiary winding strand W3a is wound outside the first winding W1, and the second tertiary winding strand W3b is wound outside the first tertiary winding strand W3a. Referring to another winding method shown in FIG3 , the tertiary winding W3 includes a first tertiary winding strand W3a and a second tertiary winding strand W3b. The first winding W1, the first tertiary winding strand W3a, and the second tertiary winding strand W3b are all wound closely to the first C-shaped magnetic core 1.

[0062] 6 , the bypass core 3 comprises a plurality of magnetic sheets 3 a with air gaps 3 b disposed between the connected magnetic sheets 3 a. The magnetic resistance of the bypass core 3 can be adjusted by adjusting the combination of the magnetic sheets and the air gaps.

[0063] In the embodiments shown in Figures 2, 3, 4, and 6, the second winding W2 is entirely wound around the second C-shaped magnetic core 2. This winding method differs from the winding method shown in Figure 7 in subsequent parameter calculations. Their common points are that flux regulating blocks 6 are provided at both joints between the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2. One end of the bypass magnetic core 3 is connected to one end of the first C-shaped magnetic core 1, and the other end of the bypass magnetic core 3 is connected to one end of the second C-shaped magnetic core 2. The second winding W2 is entirely wound around the second C-shaped magnetic core 2. The cross-sectional area of ​​the first C-shaped magnetic core 1 is calculated based on its magnetic density, which is determined by the material properties and heat dissipation parameters of the first C-shaped magnetic core 1. The cross-sectional area of ​​the second C-shaped magnetic core 2 is calculated based on its magnetic density, which is calculated based on the magnetic density of the first C-shaped magnetic core 1 and the coupling coefficient between the first winding W1 wound on the first C-shaped magnetic core 1 and the second winding W2 wound on the second C-shaped magnetic core 2.

[0064] By adjusting the magnetic path length, cross-sectional area, and magnetic permeability of the first C-shaped magnetic core 1, the second C-shaped magnetic core 2, the bypass magnetic core 3, and the magnetic flux regulating block 6, the magnetic resistance of the magnetic flux regulating block 6 and the magnetic resistance of the bypass magnetic core 3 are changed to control the excitation inductance of the first winding W1 and the leakage inductance of the first winding W1 to the second winding W2;

[0065] The cross-sectional area of ​​the first C-shaped magnetic core 1 is calculated according to formula A1, and the formula A1 is: Ae1 = V1*T1 / (2 * N1 * B1), where Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core 1, V1 is the voltage applied to the connection port of the first winding W1, T1 is the switch circumference of the power switch in the DC conversion circuit connected to the first winding W1 (i.e., the switch circumference of the power switch in the bidirectional high-voltage DC conversion circuit in Figure 8), N1 is the number of turns of the first winding W1, and B1 is the magnetic flux density of the first C-shaped magnetic core 1.

[0066] The cross-sectional area of ​​the second C-shaped magnetic core 2 is calculated according to formula A2, which is: Ae2 = V2*T2 / (2*N2*B2), where Ae2 is the cross-sectional area of ​​the second C-shaped magnetic core 2, V2 is the voltage applied to the connection port of the second winding W2, T2 is the switch circumference of the power switch in the DC conversion circuit connected to the second winding W2 (i.e., the switch circumference of the power switch in the primary conversion circuit in Figure 8), N2 is the number of turns of the first winding W2, and B2 is the magnetic flux density of the second C-shaped magnetic core 2.

[0067] The maximum number of turns of the third winding W3 is calculated according to formula A3, which is: N3=V3*T3*D / (Ae1*B1), where N3 is the maximum number of turns of the third winding W3, V3 is the voltage applied to the connection port of the third winding W3, T3 is the switch circumference of the power switch in the DC conversion circuit connected to the third winding W3 (i.e., the switch circumference of the power switch in the low-voltage DC conversion circuit in Figure 8), D is the duty cycle of the voltage waveform at the connection port of the third winding W3, Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core 1, and B1 is the magnetic flux density of the first C-shaped magnetic core 1.

[0068] Analyzing the magnetic circuit in conjunction with Figures 9, 8, and 3, when excitation is applied to port B, the total magnetic flux φ generated by the first winding W1 is split at bypass core 3 into main magnetic flux φ1 and bypass magnetic flux φ2. The magnetic flux path of the main magnetic flux φ1 is: first C-shaped magnetic core 1 - flux adjustment block 6 - second C-shaped magnetic core 2 - flux adjustment block 6 - first C-shaped magnetic core 1; the magnetic flux path of the bypass magnetic flux φ2 is: first C-shaped magnetic core 1 - bypass core 3 - first C-shaped magnetic core 1.

[0069] The magnetic resistance of the first C-shaped magnetic core 1 is calculated according to formula A4, and the formula A4 is: Rm1 = Le1 / (u0*ur1*Ae1), where Rm1 is the magnetic resistance of the first C-shaped magnetic core 1, Le1 is the magnetic path length of the first C-shaped magnetic core 1, u0 is the vacuum magnetic permeability, ur1 is the relative magnetic permeability of the magnetic material of the first C-shaped magnetic core 1, and Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core 1.

[0070] The magnetic resistance of the second C-shaped magnetic core 2 is calculated according to formula A5, and the formula A5 is: Rm2 = Le2 / (u0*ur2*Ae2), where Rm2 is the magnetic resistance of the second C-shaped magnetic core 2, Le2 is the magnetic path length of the second C-shaped magnetic core 2, u0 is the vacuum magnetic permeability, ur2 is the relative magnetic permeability of the magnetic material of the second C-shaped magnetic core 2, and Ae2 is the cross-sectional area of ​​the second C-shaped magnetic core 2.

[0071] The magnetic resistance of the two flux regulating blocks 6 is calculated according to formula A6: Rm3 = Le3 / (u0*ur3*Ae3), where Rm3 is the magnetic resistance of the two flux regulating blocks 6, Le3 is the magnetic path length of the two flux regulating blocks 6, u0 is the vacuum magnetic permeability, ur3 is the relative magnetic permeability of the magnetic material of the flux regulating blocks 6, and Ae3 is the cross-sectional area of ​​the flux regulating blocks 6. It should be noted that to reduce processing difficulty and computational complexity, the cross-sectional area of ​​the flux regulating blocks 6 can be the same as the cross-sectional area of ​​the first C-shaped magnetic core 1 (or the second C-shaped magnetic core 2).

[0072] The magnetic resistance of the bypass core 3 is calculated according to formula A7: Rm4 = Le4 / (u0*ur4*Ae4), where Rm4 is the magnetic resistance of the bypass core 3, Le4 is the magnetic path length of the bypass core 3, u0 is the vacuum permeability, ur4 is the relative permeability of the magnetic material of the bypass core 3, and Ae4 is the cross-sectional area of ​​the bypass core 3. Ae4 can be preset manually.

[0073] The total magnetic resistance of the first winding W1 is calculated according to formula A8, and the formula A8 is: Rm = Rm1+ (Rm2+Rm3*2)*Rm4 / (Rm2+Rm3*2+Rm4), where Rm is the total magnetic resistance of the first winding W1, Rm1 is the magnetic resistance of the first C-shaped magnetic core 1, Rm2 is the magnetic resistance of the second C-shaped magnetic core 2, Rm3 is the magnetic resistance of the flux adjustment block 6 (it should be noted that when the structure of Figure 4 is used for calculation, because there is only one adjustment block 6, Rm3 does not need to be multiplied by 2), and Rm4 is the magnetic resistance of the bypass magnetic core 3.

[0074] The excitation inductance of the first winding W1 is calculated according to formula A9, and the formula A9 is: Lm1 = N1² / Rm, where Lm1 is the excitation inductance of the first winding W1, N1 is the number of turns of the first winding W1, and Rm is the total magnetic resistance of the first winding W1.

[0075] The coupling coefficient between the first winding W1 and the second winding W2 is calculated according to formula A10, and the formula A10 is: k12 = Rm4 / (Rm2+Rm3*2+Rm4), where k12 is the coupling coefficient between the first winding W1 and the second winding W2, Rm4 is the magnetic resistance of the bypass core 3, Rm2 is the magnetic resistance of the second C-shaped magnetic core 2, and Rm3 is the magnetic resistance of the flux regulation block 6.

[0076] The leakage inductance of the first winding W1 to the second winding W2 is calculated according to formula A11. The formula A11 is: L12 = Lm1 * (1–k12²), where L12 is the leakage inductance of the first winding W1 to the second winding W2, Lm1 is the excitation inductance of the first winding W1, and k12 is the coupling coefficient between the first winding W1 and the second winding W2.

[0077] In another winding method shown in Figure 7, a flux regulating block 6 is provided at both joints of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2; one end of the bypass magnetic core 3 is connected to one end of the first C-shaped magnetic core 1, and the other end of the bypass magnetic core 3 is connected to one end of the second C-shaped magnetic core 2; the second winding W2 includes a first strand W2a of the second winding and a second strand W2b of the second winding, the first strand W2a of the second winding is wound around the second C-shaped magnetic core 2, and the second strand W2b of the second winding is wound around the second C-shaped magnetic core 2. The second strand W2b of the second winding is fully coupled with the first winding W1. The first strand W2a of the second winding is bypassed by the bypass core 3. The magnetic path length, cross-sectional area, and magnetic permeability of the first C-shaped magnetic core 1, the second C-shaped magnetic core 2, the bypass core 3, and the magnetic flux adjustment block 6 are adjusted to change the magnetic resistance of the magnetic flux adjustment block 6 and the magnetic resistance of the bypass core 3, so as to control the excitation inductance of the first winding W1 and the leakage inductance of the first winding W1 to the second winding W2.

[0078] The calculation method involved in FIG7 is different from the aforementioned calculation method indicators such as Ae1, Ae2, Ae3, Ae4, N3, Rm, Lm1, k122, k12, etc. involved in FIG2 to FIG6, and is specifically as follows:

[0079] The cross-sectional area of ​​the first C-shaped magnetic core 1 is calculated according to formula D1, and the formula D1 is: Ae1 = V1*T1 / (2*N1*B1), where Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core 1, V1 is the voltage applied to the connection port of the first winding W1, T1 is the switch circumference of the power switch in the DC conversion circuit connected to the first winding W1 (i.e., the switch circumference of the power switch in the bidirectional high-voltage DC conversion circuit in Figure 8), N1 is the number of turns of the first winding W1, and B1 is the magnetic flux density of the first C-shaped magnetic core 1.

[0080] The cross-sectional area of ​​the second C-shaped magnetic core 2 is calculated according to formula D2, and the formula D2 is: Ae2 = K12 * Ae1, where Ae2 is the cross-sectional area of ​​the second C-shaped magnetic core 2, k12 is the coupling coefficient between the first winding W1 and the second winding W2, and Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core 1.

[0081] The maximum number of turns of the third winding W3 is calculated according to formula D3, which is: N3 = V3*T3 / (2*Ae1*B1), where N3 is the maximum number of turns of the third winding W3, V3 is the voltage applied to the connection port of the third winding W3, T3 is the switch circumference of the power switch in the DC conversion circuit connected to the third winding W3 (i.e., the switch circumference of the power switch in the low-voltage DC conversion circuit in Figure 8), Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core 1, and B1 is the magnetic flux density of the first C-shaped magnetic core 1.

[0082] The magnetic resistance of the first C-shaped magnetic core 1 is calculated according to formula D4, and the formula D4 is: Rm1 = Le1 / (u0*ur1*Ae1), where Rm1 is the magnetic resistance of the first C-shaped magnetic core 1, Le1 is the magnetic path length of the first C-shaped magnetic core 1, u0 is the vacuum magnetic permeability, ur1 is the relative magnetic permeability of the magnetic material of the first C-shaped magnetic core 1, and Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core 1.

[0083] Analyzing the magnetic circuit in conjunction with Figures 9, 8, and 7, when excitation is applied to port B, the total magnetic flux φ generated by the first winding W1 is split at bypass core 3, dividing into main magnetic flux φ1 and bypass magnetic flux φ2. The magnetic flux path of the main magnetic flux φ1 is: first C-shaped magnetic core 1 - flux adjustment block 6 - second C-shaped magnetic core 2 - flux adjustment block 6 - first C-shaped magnetic core 1; the magnetic flux path of the bypass magnetic flux φ2 is: first C-shaped magnetic core 1 - bypass core 3 - first C-shaped magnetic core 1.

[0084] The magnetic resistance of the second C-shaped magnetic core 2 is calculated according to formula D5, and the formula D5 is: Rm2 = Le2 / (u0*ur2*Ae2), where Rm2 is the magnetic resistance of the second C-shaped magnetic core 2, Le2 is the magnetic path length of the second C-shaped magnetic core 2, u0 is the vacuum magnetic permeability, ur2 is the relative magnetic permeability of the magnetic material of the second C-shaped magnetic core 2, and Ae2 is the cross-sectional area of ​​the second C-shaped magnetic core 2.

[0085] The magnetic resistance of the two flux regulating blocks 6 is calculated according to formula D6, and the formula D6 is: Rm3 = Le3 / (u0*ur3*Ae3), wherein Rm3 is the magnetic resistance of the two flux regulating blocks 6, Le3 is the magnetic path length of the two flux regulating blocks 6, u0 is the vacuum magnetic permeability, ur3 is the relative magnetic permeability of the magnetic material of the flux regulating block 6, and Ae3 is the cross-sectional area of ​​the flux regulating block 6.

[0086] The magnetic resistance of the bypass core 3 is calculated according to formula D7, and the formula D7 is: Rm4 = Le4 / (u0*ur4*Ae4), wherein Rm4 is the magnetic resistance of the bypass core 3, Le4 is the magnetic path length of the bypass core 3, u0 is the vacuum permeability, ur4 is the relative permeability of the magnetic material of the bypass core 3, and Ae4 is the cross-sectional area of ​​the bypass core 3; Ae4 can be preset manually.

[0087] The total magnetic resistance of the first winding W1 is calculated according to formula D8, and the formula D8 is: Rm = Rm1+ (Rm2+Rm3*2)*Rm4 / (Rm2+Rm3*2+Rm4), where Rm is the total magnetic resistance of the first winding W1, Rm1 is the magnetic resistance of the first C-shaped magnetic core 1, Rm2 is the magnetic resistance of the second C-shaped magnetic core 2, Rm3 is the magnetic resistance of the flux regulation block 6, and Rm4 is the magnetic resistance of the bypass magnetic core 3.

[0088] The excitation inductance of the first winding W1 is calculated according to formula D9, and the formula D9 is: Lm1 = N1² / Rm, where Lm1 is the excitation inductance of the first winding W1, N1 is the number of turns of the first winding W1, and Rm is the total magnetic resistance of the first winding W1.

[0089] In this structure, the second winding W2 has N2 turns and is divided into two parts. The second strand W2b of the second winding is wound on the left side of the bypass core 3 and has N21 turns. The first strand W2a of the second winding is wound on the right side of the bypass core 3 and has N22 turns, where N2 = N21 + N22. In the above analysis, the total magnetic flux passes through the second strand W2b of the second winding and can be considered to be fully coupled with the first winding W1. The main magnetic flux passes through the first strand W2a of the second winding.

[0090] The coupling coefficient between the first winding W1 and the first strand W2a of the second winding is calculated according to formula D10. The formula D10 is: k122 = Rm4 / (Rm2+Rm3*2+Rm4), where k122 is the coupling coefficient between the first winding W1 and the first strand W2a of the second winding, Rm4 is the magnetic resistance of the bypass core 3, Rm2 is the magnetic resistance of the second C-shaped magnetic core 2, and Rm3 is the magnetic resistance of the flux regulation block 6.

[0091] The inductance of the first winding W1 is calculated according to formula D11. The formula D11 is: AL1 = Lm1 / (N1²), where AL1 is the inductance of the first winding W1, Lm1 is the excitation inductance of the first winding W1, and N1 is the number of turns of the first winding W1.

[0092] The leakage inductance of the first winding W1 to the second winding W2 is calculated according to formula D12. The formula D12 is: Lk12 =AL1*(N1*(1–N21 / N2))²*(1-K122²), where Lk12 is the leakage inductance of the first winding W1 to the second winding W2, AL1 is the inductance of the first winding W1, N1 is the number of turns of the first winding W1, N21 is the number of turns of the second strand W2b of the second winding, N2 is the number of turns of the second winding W2, and k122 is the coupling coefficient between the first winding W1 and the first strand W2a of the second winding.

[0093] The present invention also provides a three-port on-board charger suitable for use in on-board chargers. The three-port on-board charger includes an integrated transformer using the aforementioned leakage inductance. Referring to the principle block diagram of the on-board charger of the present invention shown in FIG8 , the three-port on-board charger also includes: a bidirectional high-voltage DC conversion circuit connected to the first winding W1, a primary-side conversion circuit connected to the second winding W2, and a low-voltage DC conversion circuit connected to the third winding W3. The magnetic path length, cross-sectional area, and magnetic permeability of the first C-shaped magnetic core 1, the second C-shaped magnetic core 2, the bypass magnetic core 3, and the flux regulating block 6 are used to change the magnetic resistance of the flux regulating block 6 and the bypass magnetic core 3, thereby controlling the excitation inductance of the first winding W1 and the leakage inductance of the first winding W1 to the second winding W2. FIG9 shows a circuit diagram of the on-board charger of the present invention.

[0094] The following application example is combined with an on-board charger.

[0095] 1. Referring to Figure 9, the circuit has three ports: Port A (connected to the PFC circuit), Port B (connected to the high-voltage battery pack), and Port C (connected to the low-voltage load within the vehicle). Ports A and B are both connected to the full-bridge circuit, while Port C is connected to the Buck converter for magnetic circuit decoupling. Therefore, when Port A transfers energy to Ports B and C simultaneously, the power received by Port B is determined by both Ports A and B. This allows for a variety of control methods, such as phase-shift control, variable frequency control, variable duty cycle control, or any combination of these. The Port C circuit chops the voltage on the low-voltage load side within the vehicle, meaning the received power is regulated by the duty cycle of S11.

[0096] 2. Take the A and B port circuits operating in DAB phase-shift control mode as an example. As shown in Figure 9, S1 and S4 have the same drive signal, and S2 and S3 have the same drive signal. Both drive signals have a duty cycle of 0.5, meaning they are on for 180 degrees and are complementary. It is important to note that a dead time should be added between these two drive signals to prevent the upper and lower transistors of the same bridge arm from conducting and causing a short circuit in the input source.

[0097] 3. As shown in Figure 9, S5, S8, and S10 have the same drive signal, and S6, S7, and S9 have the same drive signal. The duty cycle of these two drive signals is 0.5 and they are complementary. It is worth noting that a dead time should be added between these two drive signals to prevent the upper and lower transistors of the same bridge arm from being turned on, which would cause a short circuit on the output side and a short circuit on the low-voltage load side winding in the vehicle.

[0098] 4. Arms 1, 2, 3, and 4 operate in DAB mode. As shown in Figure 9, when port A transmits energy to port B, the drive signal of S5 must lag behind S1 by a certain phase Dps. Adjusting Dps adjusts the energy transfer from port A to port B. As shown in Figure 10, when port B transmits energy to port A, the drive signal of S1 must lag behind S5 by a certain phase Dps. Adjusting Dps adjusts the energy transfer from port B to port A.

[0099] 5. As shown in Figures 10 and 11, the drive signals for S9 and S10 both have a duty cycle of 0.5 and are phase-matched with the drive signals for S5 and S6, respectively. S11 switches at twice the frequency of the switching device at port A. Adjusting S11's duty cycle, Dbuck, regulates the power transfer to port C. S12 provides a freewheeling path for the inductor current, and its drive signal is complementary to that of S11. It is important to note that a dead time should be added between these two drive signals to prevent short-circuiting of the low-voltage load-side winding within the vehicle.

[0100] The above embodiments are for illustration only and are not intended to be limiting. Any equivalent modifications or variations made thereto without departing from the spirit and scope of this application should be included in the scope of the claims of this application.

Claims

1. An integrated transformer utilizing leakage inductance, comprising a first C-shaped magnetic core (1) and a second C-shaped magnetic core (2). Two ends of the first C-shaped magnetic core (1) and two ends of the second C-shaped magnetic core (2) are butt-jointed in pairs to enclose a closed magnetic ring, and it is characterized in that: The first C-shaped core (1) is wound with a first winding (W1) and a third winding (W3), and the second C-shaped core (2) is wound with a second winding (W2); the integrated transformer further includes a bypass core (3). The bypass core (3) is arranged in the middle of the magnetic ring, dividing the inner part of the magnetic ring into a first chamber (4) and a second chamber (5).

2. The integrated transformer capable of utilizing leakage inductance according to claim 1, characterized in that: At least one of the two docking joints of the first C-shaped core (1) and the second C-shaped core (2) is provided with a magnetic flux adjustment block (6), and the magnetic flux adjustment block (6) is clamped between the ends of the first C-shaped core (1) and the second C-shaped core (2).

3. The integrated transformer capable of utilizing leakage inductance according to claim 2, characterized in that: Both of the two docking joints of the first C-shaped core (1) and the second C-shaped core (2) are provided with magnetic flux adjustment blocks (6), and the magnetic flux adjustment blocks (6) are clamped between the ends of the first C-shaped core (1) and the second C-shaped core (2); one end of the bypass core (3) is connected to one end of the first C-shaped core (1), and the other end of the bypass core (3) is connected to the other end of the first C-shaped core (1).

4. The integrated transformer capable of utilizing leakage inductance according to claim 2, characterized in that: Both of the two docking joints of the first C-shaped core (1) and the second C-shaped core (2) are provided with magnetic flux adjustment blocks (6), and the magnetic flux adjustment blocks (6) are clamped between the ends of the first C-shaped core (1) and the second C-shaped core (2); the bypass core (3) is connected between the two magnetic flux adjustment blocks (6).

5. The integrated transformer capable of utilizing leakage inductance according to claim 2, characterized in that: Both of the two docking joints of the first C-shaped core (1) and the second C-shaped core (2) are provided with magnetic flux adjustment blocks (6), and the magnetic flux adjustment blocks (6) are clamped between the ends of the first C-shaped core (1) and the second C-shaped core (2); wherein, One end of the bypass core (3) is connected to one end of the second C-shaped core (2), and the other end of the bypass core (3) is connected to the other end of the second C-shaped core (2); or, One end of the bypass core (3) is connected to one end of the first C-shaped core (1), and the other end of the bypass core (3) is connected to the magnetic flux adjustment block (6).

6. The integrated transformer capable of utilizing leakage inductance according to claim 1, characterized in that: The third winding (W3) includes a first strand (W3a) and a second strand (W3b) of the third winding. The first winding (W1) is wound closely around the first C-shaped core (1). The first strand (W3a) of the third winding is wound on the outside of the first winding (W1), and the second strand (W3b) of the third winding is wound on the outside of the first strand (W3a) of the third winding.

7. The integrated transformer utilizing leakage inductance according to claim 1, characterized in that: The third winding (W3) includes a first strand (W3a) and a second strand (W3b) of the third winding. The first winding (W1), the first strand (W3a) of the third winding, and the second strand (W3b) of the third winding are all wound closely around the first C-shaped core (1).

8. The integrated transformer capable of utilizing leakage inductance according to claim 1, characterized in that: The bypass core (3) includes a plurality of magnetic sheets (3a), and there are air gaps (3b) between the adjacent magnetic sheets (3a).

9. The integrated transformer utilizing leakage inductance according to claim 3, characterized in that: The second winding (W2) is entirely wound on the second C-shaped core (2).

10. The integrated transformer capable of utilizing leakage inductance according to claim 9, characterized in that: By adjusting the magnetic path lengths, cross-sectional areas, and magnetic permeabilities of the first C-shaped core (1), second C-shaped core (2), bypass core (3), and magnetic flux adjustment block (6), the magnetic resistance of the magnetic flux adjustment block (6) and the magnetic resistance of the bypass core (3) are changed to control the exciting inductance of the first winding (W1) and the leakage inductance of the first winding (W1) with respect to the second winding (W2); Calculate the cross-sectional area of the first C-shaped core (1) according to formula A1, where formula A1 is: Ae1 = V1*T1 / (2 * N1 * B1), where Ae1 is the cross-sectional area of the first C-shaped core (1), V1 is the voltage applied to the connection port of the first winding (W1), T1 is the switching period of the power switch in the DC conversion circuit connected to the first winding (W1), N1 is the number of turns of the first winding (W1), and B1 is the magnetic flux density of the first C-shaped core (1); Calculate the cross-sectional area of the second C-shaped core (2) according to formula A2, where formula A2 is: Ae2 = V2*T2 / (2 * N2 * B2), where Ae2 is the cross-sectional area of the second C-shaped core (2), V2 is the voltage applied to the connection port of the second winding (W2), T2 is the switching period of the power switch in the DC conversion circuit connected to the second winding (W2), N2 is the number of turns of the first winding (W2), and B2 is the magnetic flux density of the second C-shaped core (2); Calculate the maximum number of turns of the third winding W3 according to formula A3, where formula A3 is: N3=V3*T3*D / (Ae1*B1), where N3 is the maximum number of turns of the third winding W3, V3 is the voltage applied to the connection port of the third winding W3, T3 is the switching period of the power switch in the DC conversion circuit connected to the third winding W3, D is the duty cycle of the voltage waveform at the connection port of the third winding W3, Ae1 is the cross-sectional area of the first C-shaped core 1, and B1 is the magnetic flux density of the first C-shaped core 1; Calculate the magnetic resistance of the first C-shaped core (1) according to formula A4, where formula A4 is: Rm1 = Le1 / (u0*ur1*Ae1), where Rm1 is the magnetic resistance of the first C-shaped core (1), Le1 is the magnetic path length of the first C-shaped core (1), u0 is the permeability of free space, ur1 is the relative magnetic permeability of the magnetic material of the first C-shaped core (1), and Ae1 is the cross-sectional area of the first C-shaped core (1); Calculate the magnetic resistance of the second C-shaped magnetic core (2) according to Formula A5, where Formula A5 is: Rm2 = Le2 / (u0*ur2*Ae2), where Rm2 is the magnetic resistance of the second C-shaped magnetic core (2), Le2 is the magnetic path length of the second C-shaped magnetic core (2), u0 is the permeability of free space, ur2 is the relative permeability of the magnetic material of the second C-shaped magnetic core (2), and Ae2 is the cross-sectional area of the second C-shaped magnetic core (2); Calculate the magnetic resistance of the two flux adjustment blocks (6) according to Formula A6, where Formula A6 is: Rm3 = Le3 / (u0*ur3*Ae3), where Rm3 is the magnetic resistance of the two flux adjustment blocks (6), Le3 is the magnetic path length of the two flux adjustment blocks (6), u0 is the permeability of free space, ur3 is the relative permeability of the magnetic material of the flux adjustment block (6), and Ae3 is the cross-sectional area of the flux adjustment block (6); Calculate the magnetic resistance of the bypass magnetic core (3) according to Formula A7, where Formula A7 is: Rm4 = Le4 / (u0*ur4*Ae4), where Rm4 is the magnetic resistance of the bypass magnetic core (3), Le4 is the magnetic path length of the bypass magnetic core (3), u0 is the permeability of free space, ur4 is the relative permeability of the magnetic material of the bypass magnetic core (3), and Ae4 is the cross-sectional area of the bypass magnetic core (3); Calculate the total magnetic resistance of the first winding (W1) according to Formula A8, where Formula A8 is: Rm = Rm1+ (Rm2+Rm3*2)*Rm4 / (Rm2+Rm3*2+Rm4), where Rm is the total magnetic resistance of the first winding (W1), Rm1 is the magnetic resistance of the first C-shaped magnetic core (1), Rm2 is the magnetic resistance of the second C-shaped magnetic core (2), Rm3 is the magnetic resistance of the flux adjustment block (6), and Rm4 is the magnetic resistance of the bypass magnetic core (3); Calculate the exciting inductance of the first winding (W1) according to Formula A9, where Formula A9 is: Lm1 = N1² / Rm, where Lm1 is the exciting inductance of the first winding (W1), N1 is the number of turns of the first winding (W1), and Rm is the total magnetic resistance of the first winding (W1); Calculate the coupling coefficient between the first winding (W1) and the second winding W2 according to Formula A10, where Formula A10 is: k12 = Rm4 / (Rm2+Rm3*2+Rm4), where k12 is the coupling coefficient between the first winding (W1) and the second winding (W2), Rm4 is the magnetic resistance of the bypass magnetic core (3), Rm2 is the magnetic resistance of the second C-shaped magnetic core (2), and Rm3 is the magnetic resistance of the flux adjustment block (6); Calculate the leakage inductance of the first winding (W1) with respect to the second winding (W2) according to formula A11, where formula A11 is: L12 = Lm1 * (1–k12²), L12 is the leakage inductance of the first winding (W1) with respect to the second winding (W2), Lm1 is the magnetizing inductance of the first winding (W1), and k12 is the coupling coefficient between the first winding (W1) and the second winding (W2).

11. The integrated transformer capable of utilizing leakage inductance according to claim 3, characterized in that: The second winding (W2) includes a first strand of the second winding (W2a) and a second strand of the second winding (W2b). The first strand of the second winding (W2a) is wound around the second C-shaped core (2), and the second strand of the second winding (W2b) is wound around the second C-shaped core (2) and the bypass core (3).

12. The integrated transformer capable of utilizing leakage inductance according to claim 11, wherein: By adjusting the magnetic path lengths, cross-sectional areas, and magnetic permeabilities of the first C-shaped core (1), the second C-shaped core (2), the bypass core (3), and the magnetic flux regulating block (6), change the magnetic resistance of the magnetic flux regulating block (6) and the magnetic resistance of the bypass core (3) to control the magnetizing inductance of the first winding (W1) and the leakage inductance of the first winding (W1) with respect to the second winding (W2); Calculate the cross-sectional area of the first C-shaped core (1) according to formula D1, where formula D1 is: Ae1 = V1*T1 / (2 * N1 * B1), where Ae1 is the cross-sectional area of the first C-shaped core (1), V1 is the voltage applied to the connection port of the first winding (W1), T1 is the switching period of the power switch in the DC conversion circuit connected to the first winding (W1), N1 is the number of turns of the first winding (W1), and B1 is the magnetic flux density of the first C-shaped core (1); Calculate the cross-sectional area of the second C-shaped core (2) according to formula D2, where formula D2 is: Ae2 = K12 * Ae1, where Ae2 is the cross-sectional area of the second C-shaped core (2), k12 is the coupling coefficient between the first winding (W1) and the second winding (W2), and Ae1 is the cross-sectional area of the first C-shaped core (1); Calculate the maximum number of turns of the third winding (W3) according to formula D3, where formula D3 is: N3 = V3*T3 / (2 * Ae1 * B1), where N3 is the maximum number of turns of the third winding (W3), V3 is the voltage applied to the connection port of the third winding (W3), T3 is the switching period of the power switch in the DC conversion circuit connected to the third winding (W3), Ae1 is the cross-sectional area of the first C-shaped core (1), and B1 is the magnetic flux density of the first C-shaped core (1); Calculate the magnetic resistance of the first C-shaped magnetic core (1) according to formula D4. The formula D4 is: Rm1 = Le1 / (u0*ur1*Ae1), where Rm1 is the magnetic resistance of the first C-shaped magnetic core (1), Le1 is the magnetic path length of the first C-shaped magnetic core (1), u0 is the permeability of free space, ur1 is the relative permeability of the magnetic material of the first C-shaped magnetic core (1), and Ae1 is the cross-sectional area of the first C-shaped magnetic core (1); Calculate the magnetic resistance of the second C-shaped magnetic core (2) according to formula D5. The formula D5 is: Rm2 = Le2 / (u0*ur2*Ae2), where Rm2 is the magnetic resistance of the second C-shaped magnetic core (2), Le2 is the magnetic path length of the second C-shaped magnetic core (2), u0 is the permeability of free space, ur2 is the relative permeability of the magnetic material of the second C-shaped magnetic core (2), and Ae2 is the cross-sectional area of the second C-shaped magnetic core (2); Calculate the magnetic resistance of the two magnetic flux adjustment blocks (6) according to formula D6. The formula D6 is: Rm3 = Le3 / (u0*ur3*Ae3), where Rm3 is the magnetic resistance of the two magnetic flux adjustment blocks (6), Le3 is the magnetic path length of the two magnetic flux adjustment blocks (6), u0 is the permeability of free space, ur3 is the relative permeability of the magnetic material of the magnetic flux adjustment block (6), and Ae3 is the cross-sectional area of the magnetic flux adjustment block (6); Calculate the magnetic resistance of the bypass magnetic core (3) according to formula D7. The formula D7 is: Rm4 = Le4 / (u0*ur4*Ae4), where Rm4 is the magnetic resistance of the bypass magnetic core (3), Le4 is the magnetic path length of the bypass magnetic core (3), u0 is the permeability of free space, ur4 is the relative permeability of the magnetic material of the bypass magnetic core (3), and Ae4 is the cross-sectional area of the bypass magnetic core (3); Calculate the total magnetic resistance of the first winding (W1) according to formula D8. The formula D8 is: Rm = Rm1+ (Rm2+Rm3*2)*Rm4 / (Rm2+Rm3*2+Rm4), where Rm is the total magnetic resistance of the first winding (W1), Rm1 is the magnetic resistance of the first C-shaped magnetic core (1), Rm2 is the magnetic resistance of the second C-shaped magnetic core (2), Rm3 is the magnetic resistance of the magnetic flux adjustment block (6), and Rm4 is the magnetic resistance of the bypass magnetic core (3); Calculate the exciting inductance of the first winding (W1) according to formula D9. The formula D9 is: Lm1 = N1² / Rm, where Lm1 is the exciting inductance of the first winding (W1), N1 is the number of turns of the first winding (W1), and Rm is the total magnetic resistance of the first winding (W1); Calculate the coupling coefficient between the first winding (W1) and the first strand of the second winding (W2a) according to formula D10, where formula D10 is: k122 = Rm4 / (Rm2 + Rm3*2 + Rm4), where k122 is the coupling coefficient between the first winding (W1) and the first strand of the second winding (W2a), Rm4 is the magnetic resistance of the bypass magnetic core (3), Rm2 is the magnetic resistance of the second C-shaped magnetic core (2), and Rm3 is the magnetic resistance of the magnetic flux adjustment block (6); Calculate the inductance coefficient of the first winding (W1) according to formula D11, where formula D11 is: AL1 = Lm1 / (N1²), AL1 is the inductance coefficient of the first winding (W1), Lm1 is the exciting inductance of the first winding (W1), and N1 is the number of turns of the first winding (W1); Calculate the leakage inductance of the first winding (W1) with respect to the second winding (W2) according to formula D12, where formula D12 is: Lk12 = AL1*(N1*(1–N21 / N2))²*(1 - K122²), Lk12 is the leakage inductance of the first winding (W1) with respect to the second winding (W2), AL1 is the inductance coefficient of the first winding (W1), N1 is the number of turns of the first winding (W1), N21 is the number of turns of the second strand of the second winding (W2b), N2 is the number of turns of the second winding (W2), and k122 is the coupling coefficient between the first winding (W1) and the first strand of the second winding (W2a).

13. A three-port on-vehicle charger applicable to an on-vehicle charger, characterized in that: In addition to the integrated transformer that can utilize leakage inductance according to any one of claims 1 to 12, the three-port on-vehicle charger further includes: a bidirectional high-voltage DC conversion circuit connected to the first winding (W1), a primary conversion circuit connected to the second winding (W2), and a low-voltage DC conversion circuit connected to the third winding (W3); wherein, The magnetic path lengths, cross-sectional areas, and magnetic permeabilities of the first C-shaped magnetic core (1), the second C-shaped magnetic core (2), the bypass magnetic core (3), and the magnetic flux adjustment block (6) are used to change the magnetic resistance of the magnetic flux adjustment block (6) and the magnetic resistance of the bypass magnetic core (3) to control the exciting inductance of the first winding (W1) and the leakage inductance of the first winding (W1) with respect to the second winding (W2).

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