Integrated transformer with adjustable leakage inductance and on-board charger

Abstract

Disclosed in the present invention are an integrated transformer with adjustable leakage inductance and an on-board charger. The integrated transformer comprises a first C-shaped magnetic core and a second C-shaped magnetic core; the first C-shaped magnetic core and the second C-shaped magnetic core are combined to form a closed magnetic ring, and a bypass magnetic core is provided in the middle of the magnetic ring to divide the interior of the magnetic ring into a first cavity and a second cavity; a first winding, a third winding and a first strand of a second winding are wound on the first C-shaped magnetic core and pass through the first cavity; and a second strand of the second winding is wound on the second C-shaped magnetic core and passes through the second cavity. According to the present invention, a magnetic core is reused, the coupling between transformer windings is controlled to obtain leakage inductance as resonant inductance, four power magnetic elements in the prior art are integrated into a transformer for use in an on-board charger, so that the number of magnetic devices of a converter is reduced, thereby reducing costs; the power density of the converter is greatly increased; and the overall loss of the converter is reduced, the efficiency is improved, and the costs of the magnetic devices of the converter are reduced.

Description

Integrated transformer with adjustable leakage inductance and on-board charger Technical Field

[0001] The present invention relates to a transformer, and in particular to an integrated transformer with adjustable leakage inductance and integrating multiple coils, as well as an on-vehicle charger. Background Art

[0002] In existing on-board charger technology, the DC conversion circuit is generally divided into two stages. The front-stage circuit is a bidirectional high-voltage DC conversion circuit, whose output is connected to the on-board power battery and provides a high-voltage DC charging current. This circuit topology is generally an LLC resonant circuit, in which the resonant inductor and the main transformer are independent of each other and are used in the circuit as two discrete components. The rear-stage circuit is a low-voltage DC conversion circuit, whose input is connected in parallel with the on-board power battery to the output of the front-stage bidirectional high-voltage DC conversion circuit, and the output is connected to the low-voltage battery and other low-voltage loads. Its circuit topology is also often an LLC resonant circuit, in which the transformer and resonant inductor are independent of each other and are used as two discrete components. Referring to the circuit diagram of the existing on-board charger transformer shown in Figure 1, the two-stage DC conversion circuit has a total of two resonant inductors, two transformers, and a total of four power magnetic devices. For on-board integrated charging products, there are the following problems: 1. The total loss of multiple power magnetic devices is large, which is not conducive to improving product efficiency; 2. The multiple power magnetic devices generate severe heat overall, and the heat dissipation cost is high; 3. The total volume of multiple magnetic components is large, which is not conducive to the high power density and miniaturization development of the product; 4. The total cost of multiple magnetic components is high.

[0003] Although some transformers in existing technical solutions mention the use of leakage inductance as a resonant inductor, the following problems still exist: 1. The design and control of leakage inductance are relatively difficult; 2. In multi-level output applications such as in-vehicle charging products, the front- and rear-stage DC conversion circuits still require the use of more than two power magnetic devices, which results in relatively high power consumption and cost. Summary of the Invention

[0004] In order to solve the above-mentioned defects in the prior art, the present invention proposes an integrated transformer with adjustable leakage inductance and an on-board charger.

[0005] The technical solution adopted by the present invention is: an integrated transformer with adjustable leakage inductance, comprising a first C-shaped magnetic core and a second C-shaped magnetic core, wherein the two ends of the first C-shaped magnetic core and the two ends of the second C-shaped magnetic core are connected in pairs to form a closed magnetic ring, which also includes a first winding, a second winding, a third winding, and a bypass magnetic core; the bypass magnetic core is arranged in the middle of the magnetic ring, dividing the interior of the magnetic ring into a first cavity and a second cavity; the second winding comprises a first strand of the second winding and a second strand of the second winding connected in series with each other; the first winding, the third winding and the first strand of the second winding are wound on the first C-shaped magnetic core and pass through the first cavity; the second strand of the second winding is wound on the second C-shaped magnetic core and passes through the second cavity.

[0006] Optionally, a magnetic flux regulating block is provided at least at one of the two joints between the first C-shaped magnetic core and the second C-shaped magnetic core, and the magnetic flux regulating block is clamped between the ends of the first C-shaped magnetic core and the second C-shaped magnetic core.

[0007] In an optional solution, a flux regulating block is provided at both joints of the first C-shaped magnetic core and the second C-shaped magnetic core, and the flux regulating block is clamped between the ends of the first C-shaped magnetic core and the second C-shaped magnetic core; one end of the bypass magnetic core is connected to one end of the first C-shaped magnetic core, and the other end of the bypass magnetic core is connected to the other end of the first C-shaped magnetic core.

[0008] In another optional solution, a flux regulating block is provided at both joints of the first C-shaped magnetic core and the second C-shaped magnetic core, and the flux regulating block is clamped between the ends of the first C-shaped magnetic core and the second C-shaped magnetic core; the bypass magnetic core is connected between the two flux regulating blocks.

[0009] In another optional scheme, a flux adjustment block is provided at both joints of the first C-shaped magnetic core and the second C-shaped magnetic core, and the flux adjustment block is clamped between the ends of the first C-shaped magnetic core and the second C-shaped magnetic core; wherein, one end of the bypass magnetic core is connected to one end of the second C-shaped magnetic core, and the other end of the bypass magnetic core is connected to the other end of the second C-shaped magnetic core; or, one end of the bypass magnetic core is connected to one end of the first C-shaped magnetic core, and the other end of the bypass magnetic core is connected to the flux adjustment block.

[0010] In an optional solution, the third winding includes a first strand of the third winding and a second strand of the third winding. The first winding is wound tightly against the first C-shaped magnetic core, the first strand of the third winding is wound on the outside of the first winding, and the second strand of the third winding is wound on the outside of the first strand of the third winding.

[0011] In another optional solution, the third winding includes a third winding first strand and a third winding second strand, and the first winding, the third winding first strand, the third winding second strand and the second winding first strand are all wound closely around the first C-shaped magnetic core.

[0012] Optionally, the bypass magnetic core includes multiple magnetic sheets, and air gaps are provided between connected magnetic sheets.

[0013] In an optional solution, the magnetic resistance of the flux regulating block and the bypass core is changed by adjusting the magnetic path length, cross-sectional area, and magnetic permeability of the first C-shaped magnetic core, the second C-shaped magnetic core, the bypass core, and the flux regulating block to control the excitation inductance of the first winding and the leakage inductance of the first winding to the second winding;

[0014] The cross-sectional area of ​​the first C-shaped magnetic core is calculated according to formula A1: Ae1=V1*T1 / (2*N1*B1), where Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core 1, V1 is the voltage applied to the connection port of the first winding, T1 is the switch circumference of the power switch in the DC conversion circuit connected to the first winding, N1 is the number of turns of the first winding, and B1 is the magnetic flux density of the first C-shaped magnetic core;

[0015] Calculate the cross-sectional area of ​​the second C-shaped magnetic core according to formula A2: Ae2=K122*Ae1, where Ae2 is the cross-sectional area of ​​the second C-shaped magnetic core, and K122 is the coupling coefficient between the first winding and the second winding;

[0016] The maximum number of turns of the third winding is calculated according to formula A3, which is: N3=V3*T3*D / (Ae1*B1), where N3 is the maximum number of turns of the third winding, V3 is the voltage applied to the connection port of the third winding, T3 is the switch circumference of the power switch in the DC conversion circuit connected to the third winding, D is the duty cycle of the voltage waveform at the connection port of the third winding, Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core, and B1 is the magnetic flux density of the first C-shaped magnetic core;

[0017] The magnetic resistance of the first C-shaped magnetic core is calculated according to Formula A4, which is: Rm1= Le1 / (u0*ur1*Ae1), where Rm1 is the magnetic resistance of the first C-shaped magnetic core, Le1 is the magnetic path length of the first C-shaped magnetic core, u0 is the vacuum permeability, ur1 is the relative permeability of the magnetic material of the first C-shaped magnetic core, and Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core;

[0018] The magnetic resistance of the second C-shaped magnetic core is calculated according to Formula A5, which is: Rm2=Le2 / (u0*ur2*Ae2), where Rm2 is the magnetic resistance of the second C-shaped magnetic core, Le2 is the magnetic path length of the second C-shaped magnetic core, u0 is the vacuum permeability, ur2 is the relative permeability of the magnetic material of the second C-shaped magnetic core, and Ae2 is the cross-sectional area of ​​the second C-shaped magnetic core;

[0019] The magnetic resistance of the two flux regulating blocks is calculated according to formula A6, which is: Rm3 = Le3 / (u0*ur3*Ae3), where Rm3 is the magnetic resistance of the two flux regulating blocks, Le3 is the magnetic path length of the two flux regulating blocks, u0 is the vacuum permeability, ur3 is the relative permeability of the magnetic material of the flux regulating block 6, and Ae3 is the cross-sectional area of ​​the flux regulating block;

[0020] The magnetic resistance of the bypass core 3 is calculated according to formula A7, which is: Rm4=Le4 / (u0*ur4*Ae4), where Rm4 is the magnetic resistance of the bypass core, Le4 is the magnetic path length of the bypass core, u0 is the vacuum permeability, ur4 is the relative permeability of the magnetic material of the bypass core, and Ae4 is the cross-sectional area of ​​the bypass core;

[0021] The total magnetic resistance of the first winding W1 is calculated according to Formula A8, which is: Rm=Rm1+(Rm2+Rm3*2)*Rm4 / (Rm2+Rm3*2+Rm4), where Rm is the total magnetic resistance of the first winding, Rm1 is the magnetic resistance of the first C-shaped magnetic core, Rm2 is the magnetic resistance of the second C-shaped magnetic core, Rm3 is the magnetic resistance of the flux regulating block, and Rm4 is the magnetic resistance of the bypass magnetic core.

[0022] Calculate the excitation inductance of the first winding according to formula A9, which is: Lm1=N1² / Rm, where Lm1 is the excitation inductance of the first winding, N1 is the number of turns of the first winding, and Rm is the total magnetic resistance of the first winding;

[0023] Calculate the coupling coefficient between the first winding and the second strand of the second winding according to Formula A10: k122=Rm4 / (Rm2+Rm3*2+Rm4), where k122 is the coupling coefficient between the first winding and the second strand of the second winding, Rm4 is the magnetic resistance of the bypass magnetic core, Rm2 is the magnetic resistance of the second C-shaped magnetic core, and Rm3 is the magnetic resistance of the flux regulation block;

[0024] The inductance of the first winding is calculated according to formula A11, where AL1=Lm1 / (N1²), wherein AL1 is the inductance of the first winding, Lm1 is the magnetizing inductance of the first winding, and N1 is the number of turns of the first winding;

[0025] The leakage inductance of the first winding W1 to the first winding is calculated according to formula A12. The formula A12 is: Lk12=AL1*(N1–N21)²*(1-K122²), wherein Lk12 is the leakage inductance of the first winding to the first winding, AL1 is the inductance coefficient of the first winding, N1 is the number of turns of the first winding, N21 is the number of turns of the first strand of the second winding, and k122 is the coupling coefficient between the first winding and the second strand of the second winding.

[0026] Optionally, the turns ratio of the first strand of the second winding and the second strand of the second winding is adjusted to control the magnetizing inductance of the first winding and the leakage inductance of the first winding to the second winding.

[0027] The present invention also provides an on-board charger, which includes, in addition to the integrated transformer with adjustable leakage inductance described above, a bidirectional high-voltage DC conversion circuit connected to the first winding, a primary-side conversion circuit connected to the second winding, and a low-voltage DC conversion circuit connected to the third winding; wherein, the magnetic path length, cross-sectional area, and magnetic permeability of the first C-shaped magnetic core, the second C-shaped magnetic core, the bypass magnetic core, and the magnetic flux adjustment block are used to change the magnetic resistance of the magnetic flux adjustment block and the magnetic resistance of the bypass magnetic core to control the excitation inductance of the first winding and the leakage inductance of the first winding to the second winding.

[0028] The beneficial effects of the technical solution provided by the present invention are:

[0029] The present invention reuses a magnetic core and controls the coupling between the transformer windings to obtain leakage inductance as a resonant inductance, integrating four power magnetic components in the prior art into an integrated transformer for application in a three-port power supply circuit. The invention has the following advantages: 1. By reusing a magnetic core, the number of magnetic components of the converter is reduced, thereby lowering the cost; 2. The leakage inductance can be adjusted according to usage requirements; 3. The power density of the converter is greatly improved; 4. The overall loss of the converter is reduced, thereby improving the efficiency. BRIEF DESCRIPTION OF THE DRAWINGS

[0030] The present invention is described in detail below with reference to the embodiments and accompanying drawings, in which:

[0031] Figure 1 is a circuit diagram of a conventional vehicle-mounted charger transformer;

[0032] FIG2 is a schematic diagram of the structure of an integrated transformer according to the present invention;

[0033] FIG3 is a schematic diagram of the integrated transformer winding and magnetic circuit of the present invention;

[0034] FIG4 is a schematic diagram of an integrated transformer having only one magnetic sheet according to the present invention;

[0035] FIG5 is a schematic diagram of an integrated transformer in which a bypass core is connected between two magnetic sheets according to the present invention;

[0036] FIG6 is a schematic diagram of an integrated transformer in which a magnetic sheet of the present invention has multiple air gaps;

[0037] FIG7 is a functional block diagram of a vehicle-mounted charger according to the present invention;

[0038] FIG8 is a circuit diagram of a vehicle-mounted charger according to the present invention;

[0039] 9 is a timing diagram of the drive signal when port A of the on-board charger of the present invention transmits energy to ports B and C;

[0040] 10 is a timing diagram of the drive signal when port B of the on-board charger of the present invention transmits energy to ports A and C;

[0041] Explanation of reference numerals: first C-shaped magnetic core (1), second C-shaped magnetic core (2), bypass magnetic core (3), magnetic sheet (3a), air gap (3b), first chamber (4), second chamber (5), magnetic flux regulating block (6), first winding (W1), second winding (W2), first strand of second winding (W2a), second strand of second winding (W2b), third winding (W3), first strand of third winding (W3a), second strand of third winding (W3b). DETAILED DESCRIPTION

[0042] In order to make the purpose, technical solutions and advantages of the present invention more clearly understood, the present invention will be further described in detail below in conjunction with the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are only used to explain the present invention and are not intended to limit the present invention.

[0043] The present invention discloses an integrated transformer with adjustable leakage inductance. The present invention reuses a magnetic core and controls the coupling between transformer windings to obtain leakage inductance as resonant inductance, integrating four power magnetic components into an on-board charger. Referring to an embodiment shown in Figure 3, the integrated transformer includes a first C-shaped magnetic core 1 and a second C-shaped magnetic core 2, where the two ends of the first C-shaped magnetic core 1 and the two ends of the second C-shaped magnetic core 2 are connected in pairs to form a closed magnetic ring. The integrated transformer also includes a first winding W1, a second winding W2, a third winding W3, and a bypass magnetic core 3; the bypass magnetic core 3 is arranged in the middle of the magnetic ring, dividing the interior of the magnetic ring into a first cavity 4 and a second cavity 5; the second winding W2 includes a first strand W2a of the second winding and a second strand W2b of the second winding connected in series with each other; the first winding W1, the third winding W3 and the first strand W2a of the second winding are wound on the first C-shaped magnetic core 1 and pass through the first cavity 4; the second strand W2b of the second winding is wound on the second C-shaped magnetic core 2 and passes through the second cavity 5.

[0044] The first winding W1, the third winding W3, and the first strand W2a of the second winding pass through the first cavity 4 on the left, and the second strand W2b of the second winding passes through the second cavity 5 on the right. The first C-shaped magnetic core 1 and the second C-shaped magnetic core 2 form a main magnetic circuit, and the bypass magnetic core 3 forms a bypass magnetic circuit. By adjusting the magnetic circuit length, cross-sectional area, and magnetic permeability of the first C-shaped magnetic core 1, the second C-shaped magnetic core 2, the bypass magnetic core 3, and the magnetic flux adjustment block 6, the magnetic resistance of the magnetic flux adjustment block 6 and the magnetic resistance of the bypass magnetic core 3 are changed to control the excitation inductance of the first winding W1 and the leakage inductance of the first winding W1 to the second winding W2. These serve as the resonant inductance of the resonant power supply circuit, thereby eliminating the need for discrete inductance, as shown in the resonant inductance Lr in FIG8 .

[0045] To achieve the purpose of regulating magnetic flux, a magnetic flux regulating block 6 is provided at least at one of the two joints between the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2. The magnetic flux regulating block 6 is sandwiched between the ends of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2. Figure 4 shows a schematic diagram of an integrated transformer with only one magnetic flux regulating block 6.

[0046] Referring to another optional embodiment shown in Figure 2, a flux adjustment block 6 is provided at both joints of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2, and the flux adjustment block 6 is clamped between the ends of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2; one end of the bypass magnetic core 3 is connected to one end of the first C-shaped magnetic core 1, and the other end of the bypass magnetic core 3 is connected to the other end of the first C-shaped magnetic core 1.

[0047] In an embodiment shown in Figure 5, a flux regulating block 6 is provided at both joints of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2, and the flux regulating block 6 is clamped between the ends of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2; the bypass magnetic core 3 is connected between the two flux regulating blocks 6.

[0048] In other embodiments, a flux regulating block 6 is provided at both joints of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2, and the flux regulating block 6 is sandwiched between the ends of the first C-shaped magnetic core 1 and the second C-shaped magnetic core 2. Alternatively, one end of the bypass magnetic core 3 is connected to one end of the second C-shaped magnetic core 2, and the other end of the bypass magnetic core 3 is connected to the other end of the second C-shaped magnetic core 2. Alternatively, one end of the bypass magnetic core 3 is connected to one end of the first C-shaped magnetic core 1, and the other end of the bypass magnetic core 3 is connected to the flux regulating block 6. It should be noted that the above embodiments do not include drawings.

[0049] A complete tertiary winding W3 can be adapted for a full-bridge rectifier circuit. To connect to a full-wave rectifier circuit, the tertiary winding W3 needs to be divided into two winding groups with a tap in the middle, resulting in two different winding methods. Referring to FIG2 , one winding method is shown in which the tertiary winding W3 includes a first strand W3a and a second strand W3b. The first winding W1 is wound closely against the first C-shaped magnetic core 1, with the first strand W3a wound outside the first winding W1 and the second strand W3b wound outside the first strand W3a.

[0050] In another winding method, the third winding W3 includes a third winding first strand W3a and a third winding second strand W3b, and the first winding W1, the third winding first strand W3a, the third winding second strand W3b and the second winding first strand W2a are all wound closely around the first C-shaped magnetic core 1.

[0051] 6 , the bypass core 3 comprises a plurality of magnetic sheets 3 a with air gaps 3 b disposed between the connected magnetic sheets 3 a. The magnetic resistance of the bypass core 3 can be adjusted by adjusting the combination of the magnetic sheets and the air gaps.

[0052] The present invention adjusts the magnetic path length, cross-sectional area, and magnetic permeability of the first C-shaped magnetic core 1, the second C-shaped magnetic core 2, the bypass magnetic core 3, and the magnetic flux adjustment block 6 to change the magnetic resistance of the magnetic flux adjustment block 6 and the magnetic resistance of the bypass magnetic core 3, thereby controlling the excitation inductance of the first winding W1 and the leakage inductance of the first winding W1 to the second winding W2. The cross-sectional area of ​​the first C-shaped magnetic core 1 is calculated based on its magnetic density, which is determined by the material properties and heat dissipation parameters of the first C-shaped magnetic core 1. The cross-sectional area of ​​the second C-shaped magnetic core 2 is calculated based on its magnetic density, which is calculated based on the magnetic density of the second C-shaped magnetic core 2 and the coupling coefficient between the first winding W1 wound on the first C-shaped magnetic core 1 and the second winding W2 wound on the second magnetic core.

[0053] The cross-sectional area of ​​the first C-shaped magnetic core 1 is calculated according to formula A1, and the formula A1 is: Ae1=V1*T1 / (2*N1 *B1), where Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core 1, V1 is the voltage applied to the connection port of the first winding W1, T1 is the switch circumference of the power switch in the DC conversion circuit connected to the first winding W1 (i.e., the switch circumference of the power switch in the bidirectional high-voltage DC conversion circuit in Figure 7), N1 is the number of turns of the first winding W1, and B1 is the magnetic flux density of the first C-shaped magnetic core 1.

[0054] The cross-sectional area of ​​the second C-shaped magnetic core 2 is calculated according to formula A2, which is: Ae2=K122*Ae1, where Ae2 is the cross-sectional area of ​​the second C-shaped magnetic core 2, and K122 is the coupling coefficient between the first winding W1 and the second winding W2.

[0055] The maximum number of turns of the third winding W3 is calculated according to formula A3, which is: N3=V3*T3*D / (Ae1*B1), where N3 is the maximum number of turns of the third winding W3, V3 is the voltage applied to the connection port of the third winding W3, T3 is the switch circumference of the power switch in the DC conversion circuit connected to the third winding W3 (i.e., the switch circumference of the power switch in the low-voltage DC conversion circuit in Figure 7), D is the duty cycle of the voltage waveform at the connection port of the third winding W3, Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core 1, and B1 is the magnetic flux density of the first C-shaped magnetic core 1.

[0056] Analyzing the magnetic circuit in conjunction with Figures 8, 7, and 3, when excitation is applied to port B, the total magnetic flux φ generated by the first winding W1 is split at bypass core 3, dividing into main magnetic flux φ1 and bypass magnetic flux φ2. The magnetic flux path of the main magnetic flux φ1 is: first C-shaped magnetic core 1 - flux adjustment block 6 - second C-shaped magnetic core 2 - flux adjustment block 6 - first C-shaped magnetic core 1; the magnetic flux path of the bypass magnetic flux φ2 is: first C-shaped magnetic core 1 - bypass core 3 - first C-shaped magnetic core 1.

[0057] The magnetic resistance of the first C-shaped magnetic core 1 is calculated according to formula A4, and the formula A4 is: Rm1= Le1 / (u0*ur1*Ae1), wherein Rm1 is the magnetic resistance of the first C-shaped magnetic core 1, Le1 is the magnetic path length of the first C-shaped magnetic core 1, u0 is the vacuum magnetic permeability, ur1 is the relative magnetic permeability of the magnetic material of the first C-shaped magnetic core 1, and Ae1 is the cross-sectional area of ​​the first C-shaped magnetic core 1.

[0058] The magnetic resistance of the second C-shaped magnetic core 2 is calculated according to formula A5, and the formula A5 is: Rm2=Le2 / (u0*ur2*Ae2), wherein Rm2 is the magnetic resistance of the second C-shaped magnetic core 2, Le2 is the magnetic path length of the second C-shaped magnetic core 2, u0 is the vacuum magnetic permeability, ur2 is the relative magnetic permeability of the magnetic material of the second C-shaped magnetic core 2, and Ae2 is the cross-sectional area of ​​the second C-shaped magnetic core 2.

[0059] The magnetic resistance of the two flux regulating blocks 6 is calculated according to formula A6: Rm3 = Le3 / (u0*ur3*Ae3), where Rm3 is the magnetic resistance of the two flux regulating blocks 6, Le3 is the magnetic path length of the two flux regulating blocks 6, u0 is the vacuum magnetic permeability, ur3 is the relative magnetic permeability of the magnetic material of the flux regulating blocks 6, and Ae3 is the cross-sectional area of ​​the flux regulating blocks 6. It should be noted that to reduce processing difficulty and computational complexity, the cross-sectional area of ​​the flux regulating blocks 6 can be the same as the cross-sectional area of ​​the first C-shaped magnetic core 1 (or the second C-shaped magnetic core 2).

[0060] The magnetic resistance of the bypass core 3 is calculated according to formula A7: Rm4=Le4 / (u0*ur4*Ae4), where Rm4 is the magnetic resistance of the bypass core 3, Le4 is the magnetic path length of the bypass core 3, u0 is the vacuum permeability, ur4 is the relative permeability of the magnetic material of the bypass core 3, and Ae4 is the cross-sectional area of ​​the bypass core 3. Ae4 can be preset manually.

[0061] The total magnetic resistance of the first winding W1 is calculated according to formula A8, and the formula A8 is: Rm=Rm1+ (Rm2+Rm3*2)*Rm4 / (Rm2+Rm3*2+Rm4), where Rm is the total magnetic resistance of the first winding W1, Rm1 is the magnetic resistance of the first C-shaped magnetic core 1, Rm2 is the magnetic resistance of the second C-shaped magnetic core 2, Rm3 is the magnetic resistance of the flux adjustment block 6 (it should be noted that when the structure of Figure 4 is used for calculation, because there is only one adjustment block 6, Rm3 does not need to be multiplied by 2), and Rm4 is the magnetic resistance of the bypass magnetic core 3.

[0062] The excitation inductance of the first winding W1 is calculated according to formula A9, and the formula A9 is: Lm1=N1² / Rm, where Lm1 is the excitation inductance of the first winding W1, N1 is the number of turns of the first winding W1, and Rm is the total magnetic resistance of the first winding W1.

[0063] Referring to the transformer structure shown in Figure 3, the second winding W2 has N2 turns and is divided into two parts. The first strand W2a of the second winding is wound around the first C-shaped magnetic core 1 and has N21 turns; the second strand W2b of the second winding is wound around the second C-shaped magnetic core 2 and has N22 turns. Here, N2 = N21 + N22. In the above analysis, the total magnetic flux passes through the first strand W2a of the second winding, which can be considered fully coupled with the first winding W1. The main magnetic flux passes through the second strand W2b of the second winding.

[0064] The coupling coefficient between the first winding W1 and the second strand W2b of the second winding is calculated according to formula A10, and the formula A10 is: k122=Rm4 / (Rm2+Rm3*2+Rm4), wherein k122 is the coupling coefficient between the first winding W1 and the second strand W2b of the second winding, Rm4 is the magnetic resistance of the bypass core 3, Rm2 is the magnetic resistance of the second C-shaped magnetic core 2, and Rm3 is the magnetic resistance of the flux regulation block 6.

[0065] The inductance of the first winding W1 is calculated according to formula A11, which is: AL1=Lm1 / (N1²), where AL1 is the inductance of the first winding W1, Lm1 is the excitation inductance of the first winding W1, and N1 is the number of turns of the first winding W1.

[0066] The leakage inductance of the first winding W1 to the first winding W1 is calculated according to formula A12, and the formula A12 is: Lk12=AL1*(N1–N21)²*(1-K122²), wherein Lk12 is the leakage inductance of the first winding W1 to the first winding W1, AL1 is the inductance coefficient of the first winding W1, N1 is the number of turns of the first winding W1, N21 is the number of turns of the first strand W2a of the second winding, and k122 is the coupling coefficient between the first winding W1 and the second strand W2b of the second winding.

[0067] In other embodiments, adjusting the turns ratio (N21:N22) of the first strand W2a of the second winding and the second strand W2b of the second winding can control the magnetic flux interlinked between the first winding W1 and the second winding W2, thereby controlling the excitation inductance of the first winding W1 and the leakage inductance of the first winding W1 to the second winding W2.

[0068] The present invention also provides an on-board charger, which includes an integrated transformer with the above-mentioned adjustable leakage inductance. Referring to the principle block diagram of the on-board charger of the present invention shown in FIG7 , the on-board charger also includes: a bidirectional high-voltage DC conversion circuit connected to the first winding W1, a primary-side conversion circuit connected to the second winding W2, and a low-voltage DC conversion circuit connected to the third winding W3. The magnetic path length, cross-sectional area, and magnetic permeability of the first C-shaped magnetic core 1, the second C-shaped magnetic core 2, the bypass magnetic core 3, and the flux adjustment block 6 are used to change the magnetic resistance of the flux adjustment block 6 and the magnetic resistance of the bypass magnetic core 3 to control the excitation inductance of the first winding W1 and the leakage inductance of the first winding W1 to the second winding W2. FIG8 shows a circuit diagram of the on-board charger of the present invention.

[0069] The following application example is combined with the on-board charger.

[0070] 1. Referring to Figure 8, the circuit has three ports: Port A (connected to the PFC circuit), Port B (connected to the high-voltage battery pack), and Port C (connected to the low-voltage load within the vehicle). Ports A and B are both connected to the full-bridge circuit, while Port C is connected to the Buck converter for magnetic circuit decoupling. Therefore, when Port A transfers energy to Ports B and C simultaneously, the power received by Port B is determined by both Ports A and B. This allows for a variety of control methods, such as phase-shift control, variable frequency control, variable duty cycle control, or any combination of these. The Port C circuit chops the voltage on the low-voltage load side within the vehicle, meaning the received power is regulated by the duty cycle of S11.

[0071] 2. Take the A and B port circuits operating in DAB phase-shift control mode as an example. As shown in Figure 8, S1 and S4 have the same drive signal, and S2 and S3 have the same drive signal. Both drive signals have a duty cycle of 0.5, meaning they are on for 180 degrees and are complementary. It is important to note that a dead time should be added between these two drive signals to prevent the upper and lower transistors of the same bridge arm from conducting and causing a short circuit in the input source.

[0072] 3. As shown in Figure 8, S5, S8, and S10 have the same drive signal, and S6, S7, and S9 have the same drive signal. The duty cycle of these two drive signals is 0.5 and they are complementary. It is worth noting that a dead time should be added between these two drive signals to prevent the upper and lower transistors of the same bridge arm from being turned on, which would cause a short circuit on the output side and a short circuit on the low-voltage load side winding in the vehicle.

[0073] 4. Arms 1, 2, 3, and 4 operate in DAB mode. As shown in Figure 8, when port A transmits energy to port B, the drive signal of S5 must lag behind S1 by a certain phase Dps. Adjusting Dps adjusts the energy transfer from port A to port B. As shown in Figure 9, when port B transmits energy to port A, the drive signal of S1 must lag behind S5 by a certain phase Dps. Adjusting Dps adjusts the energy transfer from port B to port A.

[0074] 5. As shown in Figures 9 and 10, the drive signals for S9 and S10 both have a duty cycle of 0.5 and are phase-matched with the drive signals for S5 and S6, respectively. S11's switching frequency is twice that of the switching device at port A. Adjusting S11's duty cycle, Dbuck, regulates the power transfer to port C. S12 provides a freewheeling path for the inductor current, and its drive signal is complementary to that of S11. It is important to note that a dead time should be added between these two drive signals to prevent short-circuiting of the low-voltage load-side winding within the vehicle.

[0075] The above embodiments are for illustration only and are not intended to be limiting. Any equivalent modifications or variations made thereto without departing from the spirit and scope of this application should be included in the scope of the claims of this application.

Claims

1. An integrated transformer with adjustable leakage inductance, comprising a first C-shaped magnetic core (1) and a second C-shaped magnetic core (2), wherein two ends of the first C-shaped magnetic core (1) and two ends of the second C-shaped magnetic core (2) are butt-jointed in pairs to enclose a closed magnetic ring, and it is characterized in that: It also includes a first winding (W1), a second winding (W2), a third winding (W3), and a bypass magnetic core (3); The bypass magnetic core (3) is arranged in the middle of the magnetic ring, dividing the inner part of the magnetic ring into a first chamber (4) and a second chamber (5); The second winding (W2) includes a first strand (W2a) and a second strand (W2b) of the second winding connected in series with each other; The first winding (W1), the third winding (W3), and the first strand (W2a) of the second winding are wound around the first C-shaped magnetic core (1) and pass through the first chamber (4); The second strand (W2b) of the second winding is wound around the second C-shaped magnetic core (2) and passes through the second chamber (5).

2. The integrated transformer with adjustable leakage inductance according to claim 1, wherein: At least one of the two docking joints of the first C-shaped magnetic core (1) and the second C-shaped magnetic core (2) is provided with a magnetic flux regulating block (6), and the magnetic flux regulating block (6) is clamped between the ends of the first C-shaped magnetic core (1) and the second C-shaped magnetic core (2).

3. The integrated transformer with adjustable leakage inductance according to claim 2, wherein: Both of the two docking joints of the first C-shaped magnetic core (1) and the second C-shaped magnetic core (2) are provided with magnetic flux regulating blocks (6), and the magnetic flux regulating blocks (6) are clamped between the ends of the first C-shaped magnetic core (1) and the second C-shaped magnetic core (2); one end of the bypass magnetic core (3) is connected to one end of the first C-shaped magnetic core (1), and the other end of the bypass magnetic core (3) is connected to the other end of the first C-shaped magnetic core (1).

4. The integrated transformer with adjustable leakage inductance according to claim 2, characterized in that: Both of the two docking joints of the first C-shaped magnetic core (1) and the second C-shaped magnetic core (2) are provided with magnetic flux regulating blocks (6), and the magnetic flux regulating blocks (6) are clamped between the ends of the first C-shaped magnetic core (1) and the second C-shaped magnetic core (2); the bypass magnetic core (3) is connected between the two magnetic flux regulating blocks (6).

5. The integrated transformer with adjustable leakage inductance according to claim 2, wherein: Both of the two docking joints of the first C-shaped magnetic core (1) and the second C-shaped magnetic core (2) are provided with magnetic flux regulating blocks (6), and the magnetic flux regulating blocks (6) are clamped between the ends of the first C-shaped magnetic core (1) and the second C-shaped magnetic core (2); wherein, One end of the bypass magnetic core (3) is connected to one end of the second C-shaped magnetic core (2), and the other end of the bypass magnetic core (3) is connected to the other end of the second C-shaped magnetic core (2); or, One end of the bypass magnetic core (3) is connected to one end of the first C-shaped magnetic core (1), and the other end of the bypass magnetic core (3) is connected to the magnetic flux regulating block (6).

6. The integrated transformer with adjustable leakage inductance according to claim 1, wherein: The third winding (W3) includes a first strand (W3a) and a second strand (W3b) of the third winding. The first winding (W1) is wound closely around the first C-shaped magnetic core (1). The first strand (W3a) of the third winding is wound on the outside of the first winding (W1), and the second strand (W3b) of the third winding is wound on the outside of the first strand (W3a) of the third winding.

7. The integrated transformer with adjustable leakage inductance according to claim 1, characterized in that: The third winding (W3) includes a first strand (W3a) and a second strand (W3b) of the third winding. The first winding (W1), the first strand (W3a) of the third winding, the second strand (W3b) of the third winding, and the first strand (W2a) of the second winding are all wound closely around the first C-shaped magnetic core (1).

8. The integrated transformer with adjustable leakage inductance according to claim 1, characterized in that: The bypass magnetic core (3) includes a plurality of magnetic sheets (3a), and an air gap (3b) is provided between the adjacent magnetic sheets (3a).

9. The integrated transformer with adjustable leakage inductance according to claim 3, wherein: By adjusting the magnetic path lengths, cross-sectional areas, and magnetic permeabilities of the first C-shaped magnetic core (1), the second C-shaped magnetic core (2), the bypass magnetic core (3), and the magnetic flux regulating block (6), the magnetic resistance of the magnetic flux regulating block (6) and the magnetic resistance of the bypass magnetic core (3) are changed to control the exciting inductance of the first winding (W1) and the leakage inductance of the first winding (W1) with respect to the second winding (W2); Calculate the cross-sectional area of the first C-shaped magnetic core (1) according to formula A1. The formula A1 is: Ae1 = V1 * T1 / (2 * N1 * B1), where Ae1 is the cross-sectional area of the first C-shaped magnetic core (1), V1 is the voltage applied to the connection port of the first winding (W1), T1 is the switching period of the power switch in the DC conversion circuit connected to the first winding (W1), N1 is the number of turns of the first winding (W1), and B1 is the magnetic flux density of the first C-shaped magnetic core (1); Calculate the cross-sectional area of the second C-shaped magnetic core (2) according to formula A2. The formula A2 is: Ae2 = K122 * Ae1, where Ae2 is the cross-sectional area of the second C-shaped magnetic core (2), and K122 is the coupling coefficient between the first winding (W1) and the second winding (W2); Calculate the maximum number of turns of the third winding W3 according to formula A3. The formula A3 is: N3 = V3 * T3 * D / (Ae1 * B1), where N3 is the maximum number of turns of the third winding W3, V3 is the voltage applied to the connection port of the third winding W3, T3 is the switching period of the power switch in the DC conversion circuit connected to the third winding W3, D is the duty cycle of the voltage waveform at the connection port of the third winding W3, Ae1 is the cross-sectional area of the first C-shaped magnetic core 1, and B1 is the magnetic flux density of the first C-shaped magnetic core 1; Calculate the magnetic resistance of the first C-shaped magnetic core (1) according to formula A4. The formula A4 is: Rm1 = Le1 / (u0 * ur1 * Ae1), where Rm1 is the magnetic resistance of the first C-shaped magnetic core (1), Le1 is the magnetic path length of the first C-shaped magnetic core (1), u0 is the vacuum permeability, ur1 is the relative magnetic permeability of the magnetic material of the first C-shaped magnetic core (1), and Ae1 is the cross-sectional area of the first C-shaped magnetic core (1); Calculate the magnetic resistance of the second C-shaped magnetic core (2) according to formula A5. The formula A5 is: Rm2 = Le2 / (u0 * ur2 * Ae2), where Rm2 is the magnetic resistance of the second C-shaped magnetic core (2), Le2 is the magnetic path length of the second C-shaped magnetic core (2), u0 is the vacuum permeability, ur2 is the relative magnetic permeability of the magnetic material of the second C-shaped magnetic core (2), and Ae2 is the cross-sectional area of the second C-shaped magnetic core (2); Calculate the magnetic resistance of two magnetic flux regulating blocks (6) according to formula A6, where formula A6 is: Rm3 = Le3 / (u0*ur3*Ae3), where Rm3 is the magnetic resistance of two magnetic flux regulating blocks (6), Le3 is the magnetic path length of two magnetic flux regulating blocks (6), u0 is the magnetic permeability of vacuum, ur3 is the relative magnetic permeability of the magnetic material of the magnetic flux regulating block (6), and Ae3 is the cross-sectional area of the magnetic flux regulating block (6); Calculate the magnetic resistance of the bypass magnetic core (3) according to formula A7, where formula A7 is: Rm4=Le4 / (u0*ur4*Ae4), where Rm4 is the magnetic resistance of the bypass magnetic core (3), Le4 is the magnetic path length of the bypass magnetic core (3), u0 is the magnetic permeability of vacuum, ur4 is the relative magnetic permeability of the magnetic material of the bypass magnetic core (3), and Ae4 is the cross-sectional area of the bypass magnetic core (3); Calculate the total magnetic resistance of the first winding (W1) according to formula A8, where formula A8 is: Rm=Rm1+ (Rm2+Rm3*2)*Rm4 / (Rm2+Rm3*2+Rm4), where Rm is the total magnetic resistance of the first winding (W1), Rm1 is the magnetic resistance of the first C-shaped magnetic core (1), Rm2 is the magnetic resistance of the second C-shaped magnetic core (2), Rm3 is the magnetic resistance of the magnetic flux regulating block (6), and Rm4 is the magnetic resistance of the bypass magnetic core (3); Calculate the exciting inductance of the first winding (W1) according to formula A9, where formula A9 is: Lm1=N1² / Rm, where Lm1 is the exciting inductance of the first winding (W1), N1 is the number of turns of the first winding (W1), and Rm is the total magnetic resistance of the first winding (W1); Calculate the coupling coefficient between the first winding (W1) and the second strand of the second winding (W2b) according to formula A10, where formula A10 is: k122=Rm4 / (Rm2+Rm3*2+Rm4), where k122 is the coupling coefficient between the first winding (W1) and the second strand of the second winding (W2b), Rm4 is the magnetic resistance of the bypass magnetic core (3), Rm2 is the magnetic resistance of the second C-shaped magnetic core (2), and Rm3 is the magnetic resistance of the magnetic flux regulating block (6); Calculate the inductance coefficient of the first winding (W1) according to formula A11, where formula A11 is: AL1=Lm1 / (N1²), where AL1 is the inductance coefficient of the first winding (W1), Lm1 is the exciting inductance of the first winding (W1), and N1 is the number of turns of the first winding (W1); Calculate the leakage inductance of the first winding (W1) with respect to the first winding (W1) according to formula A12, where formula A12 is: Lk12 = AL1*(N1–N21)²*(1-K122²), where Lk12 is the leakage inductance of the first winding (W1) with respect to the first winding (W1), AL1 is the inductance coefficient of the first winding (W1), N1 is the number of turns of the first winding (W1), N21 is the number of turns of the first strand of the second winding (W2a), and k122 is the coupling coefficient between the first winding (W1) and the second strand of the second winding (W2b).

10. The integrated transformer with adjustable leakage inductance according to any one of claims 2, 3, and 4, characterized in that: Adjust the turn ratio of the first strand (W2a) and the second strand (W2b) of the second winding to control the magnetizing inductance of the first winding (W1) and the leakage inductance of the first winding (W1) with respect to the second winding (W2).

11. A vehicle-mounted charger, characterized in that: In addition to the adjustable leakage inductance integrated transformer described in any one of claims 1 to 10, the on-vehicle charger further includes: a bidirectional high-voltage DC conversion circuit connected to the first winding (W1), a primary conversion circuit connected to the second winding (W2), and a low-voltage DC conversion circuit connected to the third winding (W3); wherein, The magnetic path lengths, cross-sectional areas, and magnetic permeabilities of the first C-shaped core (1), the second C-shaped core (2), the bypass core (3), and the magnetic flux adjustment block (6) are used to change the magnetic resistance of the magnetic flux adjustment block (6) and the magnetic resistance of the bypass core (3) to control the magnetizing inductance of the first winding (W1) and the leakage inductance of the first winding (W1) with respect to the second winding (W2).

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