Non-inductive active-disturbance-rejection control method for permanent magnet synchronous motor

Abstract

The invention discloses a permanent magnet synchronous motor non-inductive active-disturbance-rejection control method, which comprises the following steps: establishing a pulsating high-frequency voltage injection model of a permanent magnet synchronous motor, and injecting a high-frequency pulsating voltage signal into a direct axis of an estimated dq axis of the permanent magnet synchronous motor; a high-frequency response current equation under the dq shaft system is obtained and estimated; obtaining a simplified high-frequency response current equation according to the high-frequency response current equation and the high-frequency voltage signal, and obtaining an expression of a q-axis high-frequency response signal # imgabs0 #; based on an expression of a q-axis high-frequency response signal # imgabs1 #, a q-axis high-frequency current response estimation value containing a position estimation error is detected, useful information f (delta theta) is obtained after preset signal processing, and the useful information f (delta theta) serves as input of a PI controller so as to obtain an estimation value of the rotating speed. Compared with the traditional method, the control strategy and the signal processing method adopted by the invention are simpler and more efficient, the control performance and the stability of the system are improved, the complexity of system implementation is reduced, and engineering application and actual deployment are facilitated.

Description

Technical Field

[0001] The present invention relates to the technical field of permanent magnet synchronous motors, and in particular to a non-inductive self-disturbance rejection control method for permanent magnet synchronous motors. Background Art

[0002] Permanent magnet synchronous motors are increasingly used in aerospace, new energy and other fields due to their high power density, high efficiency and flexible control. It is crucial to obtain accurate rotor position information for controlling permanent magnet synchronous motors. The commonly used mechanical position sensors increase the volume, weight and cost of the system, and limit their application in high temperature, strong corrosion and other occasions. Position sensorless control technology is an important way to solve this problem. At present, the rotor position detection technology of permanent magnet synchronous motors in the medium and high speed stages is relatively mature, but the rotor position detection technology in the zero and low speed starting stage is still a key and difficult point in research, which has long restricted the development and application of permanent magnet synchronous motor control technology.

[0003] In the prior art, a high-frequency injection method is usually used to control the zero-low-speed starting phase of a permanent magnet synchronous motor. In the control system of a permanent magnet synchronous motor, a PI controller is usually used for the current regulator and speed regulator in the current loop and speed loop. However, although the PI controller has a simple structure and relatively simple parameter setting, it is easy to cause a large overshoot and a large torque pulsation in the system when facing a sudden change in speed or load, and the stabilization time is long and the control effect is poor; for the rotor position observation link, the bandpass filter and low-pass filter usually used will cause phase lag, which will seriously affect the observation accuracy of the rotor position, thereby causing the system to be unable to accurately observe the rotor position, thereby causing the system control performance to decline, and even causing the motor to fail to start in severe cases.

[0004] Based on this, a new technical solution is needed. Summary of the invention

[0005] In view of this, an embodiment of the present invention provides a permanent magnet synchronous motor non-inductive active disturbance rejection control method to at least solve the deficiencies of the prior art.

[0006] The embodiment of the present invention provides the following technical solutions:

[0007] The embodiment of the present invention provides a permanent magnet synchronous motor non-inductive self-disturbance rejection control method, comprising:

[0008] A pulse high-frequency voltage injection model of the permanent magnet synchronous motor is established, and a high-frequency pulse voltage signal is injected into the direct axis of the estimated dq axis of the permanent magnet synchronous motor. The high-frequency voltage signal is:

[0009]

[0010] The high-frequency voltage signal is converted to the real dq axis system through Park transformation, and the high-frequency response current signal generated in the real dq axis system is converted to the estimated axis system through inverse Park transformation, and the high-frequency response current equation in the estimated dq axis system is obtained, and the high-frequency response current equation is:

[0011]

[0012] Among them, the average inductance L=(L d +L q ) / 2, semi-differential inductance ΔL=(L q -L d ) / 2, and They represent the estimated high frequency voltages of the direct axis and quadrature axis in the dq axis system respectively;

[0013] A simplified high-frequency response current equation is obtained according to the high-frequency response current equation and the high-frequency voltage signal, and a q-axis high-frequency response signal is obtained. An expression, wherein, in the simplified high-frequency response current equation, when the q-axis high-frequency response signal approaches zero, the rotor angle error of the permanent magnet synchronous motor also approaches zero;

[0014] The simplified high frequency response current equation is:

[0015]

[0016] The q-axis high frequency response signal The expression of the expression is as follows:

[0017]

[0018] Based on the q-axis high frequency response signal By detecting the estimated value of the q-axis high-frequency current response containing the position estimation error, and obtaining useful information f(Δθ) after a preset signal processing, and using the useful information f(Δθ) as the input of the PI controller to obtain the estimated value of the rotor position and speed, the expression of the useful information f(Δθ) is as follows:

[0019]

[0020] The estimated value of the rotational speed is subtracted from a given reference value and then sent to a speed loop self-disturbance rejection rotational speed regulator for closed-loop regulation.

[0021] Furthermore, the pulsating high-frequency voltage injection model includes a flux equation, a voltage equation, a motion equation, and a state equation;

[0022] Wherein, the magnetic flux equation is:

[0023]

[0024] Among them, ψ d and ψ q are the magnetic flux of d and q axes respectively; L d and L q are the inductances of the d and q axes, i d and i q are the currents of the d and q axes, ψ f is the permanent magnet flux;

[0025] The voltage equation is:

[0026]

[0027] Among them, u d and u q are the voltages of the d and q axes respectively; R s is the stator resistance, p is the differential operator, ω e is the rotor electrical angular velocity;

[0028] The torque equation is:

[0029]

[0030] Among them, T e is the electromagnetic torque, p n is the pole pair number;

[0031] The equation of motion is:

[0032]

[0033] In the formula, ω r is the rotor mechanical angular velocity, J is the moment of inertia, T L is the load torque, and B is the damping coefficient. e Substituting in, we get:

[0034]

[0035] The state equation is:

[0036]

[0037] Furthermore, the frequency of the high-frequency pulse voltage signal is 0.5-2 kHz.

[0038] Furthermore, when the permanent magnet synchronous motor is in the zero-speed starting stage, the rotation angular frequency is small, and the model of the permanent magnet synchronous motor under high-frequency signal excitation is equivalent to a pure inductance model. At this time, the voltage equation can be simplified as follows:

[0039]

[0040] Among them, u dh 、u qh 、i dh 、i qh , are the high-frequency voltage and current of the d and q axes respectively.

[0041] Furthermore, obtaining useful information after the preset signal processing includes:

[0042] Extracting the q-axis high-frequency response signal by a second-order generalized integrator, and multiplying the q-axis high-frequency response signal by a modulation signal;

[0043] The input required by the PI controller is obtained by low-pass filtering. The formula of the input is as follows:

[0044]

[0045] Furthermore, the method for obtaining the current loop self-interference rejection current regulator of the q-axis is as follows:

[0046] The q-axis current state equation is obtained, and the q-axis current state equation is as follows:

[0047]

[0048] Among them, b q =1 / L q , q-axis current loop system disturbance f1 = -R s i q / L q -ω e L d i d / L q -ω e ψ f / L q , f q is the total disturbance and serves as the new expanded state variable;

[0049] Set state variables x3 and x4, x3 = i q , x4=f q , the expanded state equation is as follows:

[0050]

[0051] The corresponding extended state observer is designed as follows:

[0052]

[0053] The design control rate is in Designq is the proportional coefficient of the control rate, ref q is the reference current given by the q axis.

[0054] Furthermore, the method for obtaining the current loop self-interference rejection current regulator of the d-axis is as follows:

[0055] The d-axis current state equation is obtained, and the d-axis current state equation is as follows:

[0056]

[0057] Among them, b d =1 / L d ; System disturbance f2 = -R s i d / L d +ω e L q i q / L d , f d is the total disturbance and serves as the new expanded state variable;

[0058] Set state variables x5 and x6, x5 = i d , x6=f d ,thereby:

[0059]

[0060] The corresponding extended state observer is designed as follows:

[0061]

[0062] The design control rate is in Design d is the proportional coefficient of the control rate, ref d is the reference current given for the d-axis.

[0063] Furthermore, the method for obtaining the speed loop anti-disturbance speed regulator is as follows:

[0064] Expand the disturbance in the speed loop into a new state variable f ω , the corresponding mechanical motion equation is:

[0065]

[0066] Among them, b ω =3p n i q ψ f / 2J, speed loop system disturbance f3=-T L / J-Bω r / J,fω is the total disturbance and serves as the new expanded state variable;

[0067] Set state variables x7 and x8, x7 = i q , x8=f ω ,thereby:

[0068]

[0069] The corresponding extended state observer is designed as follows:

[0070]

[0071] The design control rate is in Design ω is the proportional coefficient of the control rate, ref ω is the given reference rotor mechanical angular velocity.

[0072] Compared with the prior art, the at least one technical solution adopted in the embodiment of the present invention can achieve the following beneficial effects:

[0073] The present invention provides a method for controlling a permanent magnet synchronous motor without inductive self-disturbance rejection, which replaces the traditional PI controller by introducing a current regulator and a speed regulator based on self-disturbance rejection control, thereby significantly enhancing the robustness and dynamic response performance of the system, reducing the overshoot phenomenon of the system, and improving the stability and control accuracy. The application of self-disturbance rejection control effectively compensates for the external disturbance and internal uncertainty of the system, and enhances the anti-disturbance capability of the system under complex working conditions. By optimizing the current loop regulator, the torque pulsation is reduced, the smoothness of the motor operation is improved, and the energy efficiency of the system is improved;

[0074] Furthermore, a second-order generalized integrator is used to replace the traditional bandpass filter and low-pass filter. The second-order generalized integrator can be used to accurately extract or filter out a specific frequency component signal, greatly improving the extraction accuracy of the high-frequency response signal. At the same time, it improves the filtering effect of the high-frequency current harmonic signal in the current loop feedback loop, enhances the accuracy of the rotor position observation, and thus improves the control accuracy and stability of the motor. BRIEF DESCRIPTION OF THE DRAWINGS

[0075] In order to more clearly illustrate the technical solutions of the embodiments of the present application, the drawings required for use in the embodiments will be briefly introduced below. Obviously, the drawings described below are only some embodiments of the present application. For ordinary technicians in this field, other drawings can be obtained based on these drawings without paying creative work.

[0076] Figure 1 It is the relationship diagram of each coordinate system;

[0077] Figure 2 It is the structure diagram of the rotor position observer;

[0078] Figure 3 It is the structure diagram of the second-order generalized integrator;

[0079] Figure 4 is the amplitude-phase characteristic curve of SOGI;

[0080] Figure 5 This is the basic principle block diagram of the first-order linear auto-disturbance rejection;

[0081] Figure 6 A system block diagram of a permanent magnet synchronous motor non-inductive self-disturbance rejection control method of the present invention;

[0082] Figure 7 It is the speed tracking waveform of the existing permanent magnet synchronous motor;

[0083] Figure 8 The speed tracking waveform of the permanent magnet synchronous motor non-inductive self-disturbance rejection control method of the present invention is

[0084] Fig. 9 An angle tracking waveform for an existing permanent magnet synchronous motor;

[0085] Fig.10 It is an angle tracking waveform of a permanent magnet synchronous motor non-inductive self-disturbance rejection control method of the present invention. DETAILED DESCRIPTION

[0086] The embodiments of the present application are described in detail below with reference to the accompanying drawings.

[0087] The following describes the implementation methods of the present application through specific examples, and those skilled in the art can easily understand other advantages and effects of the present application from the contents disclosed in this specification. Obviously, the described embodiments are only a part of the embodiments of the present application, rather than all the embodiments. The present application can also be implemented or applied through other different specific implementation methods, and the details in this specification can also be modified or changed in various ways based on different viewpoints and applications without departing from the spirit of the present application. It should be noted that, in the absence of conflict, the following embodiments and the features in the embodiments can be combined with each other. Based on the embodiments in the present application, all other embodiments obtained by ordinary technicians in the field without making creative work belong to the scope of protection of the present application.

[0088] It should be noted that various aspects of the embodiments within the scope of the appended claims are described below. It should be apparent that the aspects described herein can be embodied in a wide variety of forms, and any specific structure and / or function described herein is merely illustrative. Based on the present application, it should be understood by those skilled in the art that an aspect described herein can be implemented independently of any other aspect, and two or more of these aspects can be combined in various ways. For example, any number and aspect described herein can be used to implement the device and / or practice the method. In addition, other structures and / or functionalities other than one or more of the aspects described herein can be used to implement this device and / or practice this method.

[0089] It should also be noted that the illustrations provided in the following embodiments are only schematic illustrations of the basic concept of the present application. The drawings only show components related to the present application rather than being drawn according to the number, shape and size of components in actual implementation. In actual implementation, the type, quantity and proportion of each component may be changed arbitrarily, and the component layout may also be more complicated.

[0090] Additionally, in the following description, specific details are provided to facilitate a thorough understanding of the examples. However, it will be understood by those skilled in the art that the examples can be practiced without these specific details.

[0091] In view of the weak back electromotive force signal of the permanent magnet synchronous motor at zero and low speed, the low signal-to-noise ratio of the extracted signal, which cannot meet the control requirements, the present invention adopts the method of pulse high-frequency voltage injection, by injecting a high-frequency sinusoidal voltage signal of a specific frequency into the direct axis under the dq axis system, and superimposing the high-frequency signal. The two-phase voltage under the dq axis system is transformed by inverse PARK to obtain a two-phase static coordinate system, and then the three-phase duty cycle signal is calculated by the SVPWM module and sent to the inverter bridge to control the permanent magnet synchronous motor. At the same time, the three-phase current of the motor when working is detected by the current sensor, and transformed into the two-phase current under the dq axis system through CLARK and PARK. Finally, it is fed back to the current regulator to complete the closed-loop control.

[0092] In order to improve the estimation accuracy of the rotor position, a second-order generalized integrator is used to replace the traditional bandpass filter and low-pass filter, thereby reducing the phase lag, improving the extraction accuracy of the high-frequency response signal, and enhancing the accuracy of the rotor position observation.

[0093] At the same time, a current regulator and speed regulator based on active anti-disturbance control are designed to replace the PI controller in the traditional current loop and speed loop, which significantly enhances the robustness and dynamic response performance of the system, reduces the overshoot and torque pulsation of the system, and thus improves the smoothness and energy efficiency of the motor operation.

[0094] The technical solutions provided by various embodiments of the present application are described below in conjunction with the accompanying drawings.

[0095] like Figures 1 to 6 As shown, a permanent magnet synchronous motor inductive self-anti-disturbance control method of the present invention mainly includes three steps: first, constructing a system model of a permanent magnet synchronous motor (PMSM) pulsating high-frequency voltage injection method; second, designing a high-frequency response signal extraction link based on a second-order generalized integrator (SOGI) to replace the traditional band-pass filter (BPF) extraction method, and at the same time using SOGI to replace the low-pass filter (LPF) in the synchronous rotating dq axis current feedback loop; third, designing a current regulator and a speed regulator based on active anti-disturbance control (ADRC) to replace the PI controllers in the traditional current loop and speed loop respectively.

[0096] Specifically, the permanent magnet synchronous motor non-inductive self-disturbance rejection control method includes:

[0097] Step S102: establish a pulse high-frequency voltage injection model of the permanent magnet synchronous motor, and inject a high-frequency pulse voltage signal into the direct axis of the estimated dq axis of the permanent magnet synchronous motor, wherein the high-frequency voltage signal is:

[0098]

[0099] Specifically, it is assumed that the three-phase PMSM is an ideal motor and satisfies the following conditions: (1) the saturation effect of the core is not considered; (2) eddy current and hysteresis losses are not considered; and (3) the current in the motor is a symmetrical three-phase current.

[0100] The mathematical model of the permanent magnet synchronous motor in the two-phase rotating dq axis system includes the flux equation, voltage equation, motion equation, and state equation:

[0101] Wherein, the magnetic flux equation is:

[0102]

[0103] Among them, ψ d and ψ q are the magnetic flux of d and q axes respectively; L d and L q are the inductances of the d and q axes, i d and i q are the currents of the d and q axes, ψ f is the permanent magnet flux;

[0104] The voltage equation is:

[0105]

[0106] Among them, u d and u q are the voltages of the d and q axes respectively; R sis the stator resistance, p is the differential operator, ω e is the rotor electrical angular velocity;

[0107] The torque equation is:

[0108]

[0109] Among them, T e is the electromagnetic torque, p n is the pole pair number;

[0110] The equation of motion is:

[0111]

[0112] In the formula, ω r is the rotor mechanical angular velocity, J is the moment of inertia, T L is the load torque, and B is the damping coefficient. e Substituting in, we get:

[0113]

[0114] The state equation is:

[0115]

[0116] Step S104: convert the high-frequency voltage signal to the real dq axis system through Park transformation, and convert the high-frequency response current signal generated in the real dq axis system to the estimated axis system through inverse Park transformation, and obtain the high-frequency response current equation in the estimated dq axis system, wherein the high-frequency response current equation is:

[0117]

[0118] Among them, the average inductance L=(L d +L q ) / 2, semi-differential inductance ΔL=(L q -L d ) / 2, and They represent the estimated high-frequency voltages of the direct axis and quadrature axis in the dq axis system respectively.

[0119] Specifically, for the convenience of analysis, two simplifications are usually required: First, in order to facilitate the separation of the injected high-frequency signal and the base frequency signal of the motor rotation, the injected signal frequency ω h Generally, 0.5 to 2kHz is selected, which is much higher than the rotation angular frequency ω of the motor. e At this time, the impedance of the motor winding is mainly determined by the inductive reactance of the self-inductance, and the winding resistance R can be ignored. sSecond, since the high-frequency injection method is mainly used for the zero-speed starting stage of PMSM, the rotation angular frequency of the motor is very small, and the back electromotive force term ω in the voltage equation e ψ f and the cross-coupling term ω e L d and ω e L q Can be ignored. At this time, the PMSM model under high-frequency signal excitation can be equivalent to a pure inductance model, and the voltage equation can be simplified to:

[0120]

[0121] Among them, u dh 、u qh 、i dh 、i qh , are the high-frequency voltage and current of the d and q axes, respectively, which can be further rewritten as follows:

[0122]

[0123] Step S106: Obtain a simplified high-frequency response current equation according to the high-frequency response current equation and the high-frequency voltage signal, and obtain a q-axis high-frequency response signal An expression, wherein, in the simplified high-frequency response current equation, when the q-axis high-frequency response signal approaches zero, the rotor angle error of the permanent magnet synchronous motor also approaches zero;

[0124] The simplified high frequency response current equation is:

[0125]

[0126] The q-axis high frequency response signal The expression of the expression is as follows:

[0127]

[0128] Specifically, the design process involves many coordinate transformations, and the relationship diagrams of various coordinate systems are as follows: Figure 1 As shown, θ is the actual rotor angle. To estimate the rotor angle, the angle error U mh cos(ω h t) is the high-frequency signal injected into the estimated direct axis, where U mh is the high frequency signal amplitude, ω his the high-frequency signal frequency. Since the real dq axis system is difficult to determine, it is necessary to first inject a high-frequency voltage signal into the estimated dq axis system, and then transform it to the real dq axis system through Park transformation. The high-frequency response current signal generated in the real dq axis system is then transformed to the estimated dq axis system through inverse Park transformation. The high-frequency response current equation in the estimated dq axis system is as follows:

[0129]

[0130] After simplification, we can get:

[0131]

[0132] Since the high-frequency sinusoidal voltage signal is injected into the direct axis of the estimated dq axis system as follows:

[0133]

[0134] in, and They represent the estimated high-frequency voltages of the direct axis and quadrature axis in the dq axis system. The estimated high-frequency current response in the dq axis system is as follows:

[0135]

[0136] After further simplification, the simplified high-frequency response current equation can be obtained:

[0137]

[0138] According to the high-frequency current response under the dq axis system, the high-frequency response currents of the direct axis and quadrature axis of the dq axis system contain rotor position error information. When the rotor angle error Δθ is 0, it also approaches 0. When it is 0, it cannot guarantee that the error Δθ approaches 0. So choose Extract the target signal and cross-axis high frequency response signal The expression is as follows

[0139]

[0140] Step S108: Based on the q-axis high frequency response signal By detecting the estimated value of the q-axis high-frequency current response containing the position estimation error, and obtaining useful information f(Δθ) after a preset signal processing, and using the useful information f(Δθ) as the input of the PI controller to obtain the estimated value of the rotor position and speed, the expression of the useful information f(Δθ) is as follows:

[0141]

[0142] Specifically, useful information obtained after preset signal processing includes:

[0143] Extracting the q-axis high-frequency response signal by a second-order generalized integrator, and multiplying the q-axis high-frequency response signal by a modulation signal;

[0144] The input required by the PI controller is obtained by low-pass filtering. The formula of the input is as follows:

[0145]

[0146] More specifically, when the position estimate gradually approaches the actual value, that is, when Δθ approaches 0, It also gradually approaches 0. Therefore, by detecting the estimated value of the quadrature axis high-frequency current response containing the position estimation error, after appropriate signal processing, the useful information f(Δθ) is extracted as the input of the regulator, and then through the PI controller, the estimated value of the rotor position and speed can be obtained. The specific implementation block diagram is as follows Figure 2 First, the high-frequency response signal of a specific injection frequency is extracted through SOGI. Then With the modulation signal sin(ω h t) and then low-pass filtered to obtain the input f(Δθ) required by the regulator as follows:

[0147]

[0148] The structure of the second-order generalized integrator is as follows Figure 3 , including a proportional and two integral links, X is the input signal, ω h is the frequency of the signal to be processed, K is the proportional coefficient. The output frequency of Y is ω h signal, while the Y' output does not contain ω h Therefore, the second-order generalized integrator can accurately extract or filter out a signal with a specific frequency component. Selecting Y as the output in the high-frequency response signal extraction link can improve the estimation accuracy of the rotor position, and selecting Y' as the output in the dq axis current loop feedback loop can accurately filter out the injected frequency signal component in the baseband feedback control signal. The transfer function of the output Y is:

[0149]

[0150] Figure 4The amplitude-phase characteristic curve of SOGI shows that at the characteristic frequency point, different values ​​of the proportional coefficient K have no effect on the system amplitude and no phase shift. Therefore, SOGI can improve the phase delay problem caused by BPF and LPF in the system, thereby improving the estimation accuracy of the rotor position. The value of K is critical. As the value of K increases, the bandwidth increases, and the amplitude remains large in the frequency region near the characteristic frequency, which is not conducive to the extraction of specific frequency signals. When the value of K is small, the system bandwidth is small, and the frequency domain amplitude attenuation outside the characteristic frequency is large, which is conducive to the extraction of characteristic signals. Since the value of K directly affects the performance of the system, the value of K should not be too large or too small. It is necessary to select a suitable proportional coefficient based on the performance requirements and stability requirements of the system to achieve a balance between response speed and stability.

[0151] Step S110: Subtract the estimated value of the rotor speed from the given reference value and send it to the speed loop self-disturbance rejection speed regulator for closed-loop regulation.

[0152] Furthermore, the current baseband signal obtained after the high-frequency component is filtered out by SOGI is subtracted from a given reference value corresponding to the current baseband signal and then sent to the current loop self-interference rejection current regulator for closed-loop regulation.

[0153] Specifically, the basic idea of ​​ADRC is to regard the unknown disturbances and modeling errors in the system as a total disturbance, estimate the total disturbance through the observer, and compensate it to the control input, so as to achieve precise control of the system. The first-order linear ADRC controller mainly includes a tracking differentiator, a linear expanded state observer, and a linear error feedback rate. Among them, the tracking differentiator mainly softens the input signal. The linear expanded state observer is the core link of ADRC, which is used to estimate the state and total disturbance of the system in real time. The linear error feedback rate estimates the state and disturbance of the system based on the linear expanded state observer, calculates the control input, and makes the system output track the desired trajectory. The basic principle block diagram is as follows: Figure 5 shown.

[0154] For a first-order system, we can write In the form of, b is the control gain, which is an inherent parameter of the system. Generally, the true value of b cannot be determined, so it is assumed that the estimated value of b is b0, and f is the total disturbance. If the inaccurate part of the control gain is included in the total disturbance, the system equation can be further written as follows:

[0155]

[0156] Select state variables

[0157]

[0158] The matrix C=[1 0],D=[0],

[0159] Design the linear observer as follows:

[0160]

[0161] L=[β1β2] T are two parameters of the state observer, which can adjust the dynamic performance of the state observer and thus determine the convergence speed. Substituting matrices A, B, C, and L into the extended state observer is as follows:

[0162]

[0163] Where z represents z1 and z2, which are the two outputs of the linear extended state observer. z1 will track the output x1, and z2 will track the total disturbance x2. The design control rate is u = (u0-z2) / b0, where u0 = ω c (ref-z1), design ω c is the proportional coefficient of the control rate, and ref is the given reference signal.

[0164] Furthermore, the method for obtaining the current loop self-interference rejection current regulator of the q-axis is as follows:

[0165] The q-axis current state equation is obtained, and the q-axis current state equation is as follows:

[0166]

[0167] Among them, b q =1 / L q , q-axis current loop system disturbance f1 = -R s i q / L q -ω e L d i d / L q -ω e ψ f / L q , f q is the total disturbance and serves as the new expanded state variable;

[0168] Set state variables x3 and x4, x3 = i q , x4=f q , the expanded state equation is as follows:

[0169]

[0170] The corresponding extended state observer is designed as follows:

[0171]

[0172] The design control rate is in Design q is the proportional coefficient of the control rate, ref q is the reference current given by the q axis.

[0173] Furthermore, the method for obtaining the current loop self-interference rejection current regulator of the d-axis is as follows:

[0174] The d-axis current state equation is obtained, and the d-axis current state equation is as follows:

[0175]

[0176] Among them, b d =1 / L d ; System disturbance f2 = -R s i d / L d +ω e L q i q / L d , f d is the total disturbance and serves as the new expanded state variable;

[0177] Set state variables x5 and x6, x5 = i d , x6=f d ,thereby:

[0178]

[0179] The corresponding extended state observer is designed as follows:

[0180]

[0181] The design control rate is in Design d is the proportional coefficient of the control rate, ref d is the reference current given for the d-axis.

[0182] Furthermore, the method for obtaining the speed loop anti-disturbance speed regulator is as follows:

[0183] Expand the disturbance in the speed loop into a new state variable f ω , the corresponding mechanical motion equation is:

[0184]

[0185] Among them, b ω =3p n iq ψ f / 2J, speed loop system disturbance f3=-T L / J-Bω r / J,f ω is the total disturbance and serves as the new expanded state variable;

[0186] Set state variables x7 and x8, x7 = i q , x8=f ω ,thereby:

[0187]

[0188] The corresponding extended state observer is designed as follows:

[0189]

[0190] The design control rate is in Design ω is the proportional coefficient of the control rate, ref ω is the given reference rotor mechanical angular velocity.

[0191] The simulation verification analysis of the present invention is as follows:

[0192] In order to verify the effectiveness and feasibility of the present invention, a complete PMSM position sensorless control system simulation model of the pulse high-frequency voltage injection method was built in the MATLAB / simulink environment for verification. Among them, the simulation motor parameters are: stator inductance L d =5.2mH,L q =17.4mH; pole pair number p n =2; stator resistance R s =0.33Ω; flux linkage ψ f =0.646Wb; moment of inertia J = 0.008kg·m 2 ; Damping coefficient B = 0.008N·m·s.

[0193] The simulation conditions are set as: DC bus voltage U dc =311V; PWM switching frequency f pwm =5kHz, variable step size ode45 algorithm is used, simulation time is 1.5s. The amplitude of high frequency injected voltage signal U mh =30V, frequency ω h =1kHz.

[0194] The test conditions are as follows: the initial reference speed is 300 rpm, the machine is started without load, and a load of 2 N·m is applied at 0.5 s. Figures 7 to 9 This is the waveform recorded for the simulation experiment.

[0195] Through observation and analysis Figure 7 and Figure 8 It can be seen that during the motor startup phase, the control systems before and after the improvement take basically the same time to stabilize the motor to the target speed, but the PI controller used before the improvement has a large overshoot, while the overshoot after the improvement is almost zero; after the sudden load at 0.5s, the speed drops before and after the improvement, and then rises back to the set speed, but in the process of speed recovery, it can be found that the overshoot after the improvement is significantly smaller than before the improvement. Through observation and analysis Fig. 9 and Fig.10 It can be seen that during the motor startup phase, the overlap of the angle tracking waveform after improvement is higher than that before improvement; when the motor is running stably, the overlap of the angle tracking waveform before and after improvement is basically the same.

[0196] The simulation results show that the position error signal extraction method based on SOGI design in the proposed method significantly improves the extraction accuracy of the rotor position error signal, thereby enhancing the accuracy of the rotor position estimation. The current regulator and speed regulator based on ADRC design can also effectively compensate for the disturbance of the system and improve the anti-interference ability of the system under complex working conditions. Whether facing the initial stage of motor startup or sudden load conditions, the proposed method can ensure a high system response speed while controlling a small overshoot, thereby improving the control accuracy and stability of the motor, and the control effect is significantly better than the traditional control method. The simulation experimental results are consistent with the theoretical analysis of the proposed method, which verifies the effectiveness and feasibility of the proposed method.

[0197] In this specification, the same or similar parts between the various embodiments can be referred to each other, and each embodiment focuses on the differences from other embodiments. In particular, for the product embodiment described later, since it corresponds to the method, the description is relatively simple, and the relevant parts can be referred to the partial description of the system embodiment.

[0198] The above is only a specific implementation of the present application, but the protection scope of the present application is not limited thereto. Any changes or substitutions that can be easily thought of by a person skilled in the art within the technical scope disclosed in the present application should be included in the protection scope of the present application. Therefore, the protection scope of the present application shall be based on the protection scope of the claims.

Claims

1. A method for controlling a permanent magnet synchronous motor without inductive interference, characterized in that: include: A pulse high-frequency voltage injection model of the permanent magnet synchronous motor is established, and a high-frequency pulse voltage signal is injected into the direct axis of the estimated dq axis of the permanent magnet synchronous motor. The high-frequency voltage signal is: The high-frequency voltage signal is converted to the real dq axis system through Park transformation, and the high-frequency response current signal generated in the real dq axis system is converted to the estimated axis system through inverse Park transformation, and the high-frequency response current equation in the estimated dq axis system is obtained, and the high-frequency response current equation is: Among them, the average inductance L=(L d +L q ) / 2, semi-differential inductance ΔL=(L q -L d ) / 2, and They represent the estimated high frequency voltages of the direct axis and quadrature axis in the dq axis system respectively; A simplified high-frequency response current equation is obtained according to the high-frequency response current equation and the high-frequency voltage signal, and a q-axis high-frequency response signal is obtained. An expression, wherein, in the simplified high-frequency response current equation, when the q-axis high-frequency response signal approaches zero, the rotor angle error of the permanent magnet synchronous motor also approaches zero; The simplified high frequency response current equation is: The q-axis high frequency response signal The expression of the expression is as follows: Based on the q-axis high frequency response signal By detecting the estimated value of the q-axis high-frequency current response containing the position estimation error and obtaining useful information f(Δθ) after a preset signal processing, the useful information f(Δθ) is used as the input of the PI controller to obtain the estimated value of the rotor speed. The expression of the useful information f(Δθ) is as follows: The estimated value of the rotational speed is subtracted from a given reference value and then sent to a speed loop self-disturbance rejection rotational speed regulator for closed-loop regulation.

2. The method for controlling a permanent magnet synchronous motor without inductive active disturbance rejection according to claim 1, characterized in that: The pulse high-frequency voltage injection model includes a flux equation, a voltage equation, a motion equation, and a state equation; Wherein, the magnetic flux equation is: Among them, ψ d and ψ q are the magnetic flux of d and q axes respectively; L d and L q are the inductances of the d and q axes, i d and i q are the currents of the d and q axes, ψ f is the permanent magnet flux; The voltage equation is: Among them, u d and u q are the voltages of the d and q axes respectively; R s is the stator resistance, p is the differential operator, ω e is the rotor electrical angular velocity; The torque equation is: Among them, T e is the electromagnetic torque, p n is the pole pair number; The equation of motion is: In the formula, ω r is the rotor mechanical angular velocity, J is the moment of inertia, T L is the load torque, and B is the damping coefficient. e Substituting in, we get: The state equation is:

3. The permanent magnet synchronous motor non-inductive auto-disturbance rejection control method according to claim 2 is characterized in that: The frequency of the high-frequency pulse voltage signal is 0.5-2 kHz.

4. The method for controlling a permanent magnet synchronous motor without inductive active disturbance rejection according to claim 3 is characterized in that: When the permanent magnet synchronous motor is in the zero-speed starting stage, the rotation angular frequency is small, and the model of the permanent magnet synchronous motor under high-frequency signal excitation is equivalent to a pure inductance model. At this time, the voltage equation can be simplified as follows: Among them, u dh 、u qh 、i dh 、i qh , are the high-frequency voltage and current of the d and q axes respectively.

5. The method for controlling a permanent magnet synchronous motor without inductive active disturbance rejection according to claim 4 is characterized in that: The obtaining of useful information after the preset signal processing includes: Extracting the q-axis high-frequency response signal by a second-order generalized integrator, and multiplying the q-axis high-frequency response signal by a modulation signal; The input required by the PI controller is obtained by low-pass filtering. The formula of the input is as follows:

6. The method for controlling a permanent magnet synchronous motor without inductive active disturbance rejection according to claim 5, characterized in that: The method of obtaining the current loop self-disturbance rejection current regulator of the q axis is as follows: The q-axis current state equation is obtained, and the q-axis current state equation is as follows: Among them, b q =1 / L q , q-axis current loop system disturbance f1 = -R s i q / L q -ω e L d i d / L q -ω e ψ f / L q , f q is the total disturbance and serves as the new expanded state variable; Set state variables x3 and x4, x3 = i q , x4=f q , the expanded state equation is as follows: The corresponding extended state observer is designed as follows: The design control rate is in Design q is the proportional coefficient of the control rate, ref q is the reference current given by the q axis.

7. The method for controlling a permanent magnet synchronous motor without inductive active disturbance rejection according to claim 6, characterized in that: The method for obtaining the current loop self-disturbance rejection current regulator of the d-axis is as follows: The d-axis current state equation is obtained, and the d-axis current state equation is as follows: Among them, b d =1 / L d ; System disturbance f2 = -R s i d / L d +ω e L q i q / L d , f d is the total disturbance and serves as the new expanded state variable; Set state variables x5 and x6, x5 = i d , x6=f d ,thereby: The corresponding extended state observer is designed as follows: The design control rate is in Design d is the proportional coefficient of the control rate, ref d is the reference current given for the d-axis.

8. The method for controlling a permanent magnet synchronous motor without inductive active disturbance rejection according to claim 7, characterized in that: The method for obtaining the speed loop anti-disturbance speed regulator is as follows: Expand the disturbance in the speed loop into a new state variable f ω , the corresponding mechanical motion equation is: Among them, b ω =3p n i q ψ f / 2J, speed loop system disturbance f3=-T L / J-Bω r / J,f ω is the total disturbance and serves as the new expanded state variable; Set state variables x7 and x8, x7 = i q , x8=f ω ,thereby: The corresponding extended state observer is designed as follows: The design control rate is in Design ω is the proportional coefficient of the control rate, ref ω is the given reference rotor mechanical angular velocity.

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