Signal transmitting link and method, transceiving link, integrated circuit and electromagnetic wave device

Abstract

A signal transmitting link and method, a transmit-receive link, an integrated circuit and an electromagnetic wave device, the signal transmitting link is applied to a radar system, the signal transmitting link comprises a transmitting end baseband digital module, a digital-to-analog conversion module, a transmitting end local oscillator and a transmitting end quadrature modulator, the digital-to-analog conversion module comprises two identical digital-to-analog converters, and the transmitting end quadrature modulator comprises two identical digital-to-analog converters. Wherein the transmitting-end baseband digital module is configured to generate two paths of orthogonal transmitting-end digital baseband signals and send the two paths of orthogonal transmitting-end digital baseband signals to the digital-to-analog converters respectively; the digital-to-analog conversion module is configured to convert the two paths of orthogonal transmitting end digital baseband signals into two paths of transmitting end analog baseband signals; the transmitting end local oscillator is configured to provide a transmitting end local oscillator signal; and the transmitting quadrature modulator is configured to perform phase shifting operation while performing frequency shifting on the transmitting end local oscillation signal based on the two paths of transmitting end analog baseband signals so as to form an FMCW radio frequency transmitting signal after predetermined phase shifting.

Description

[0001] This application claims priority to the Chinese patent application filed with the China Patent Office on June 14, 2023, with application number 202310702586.5 and invention name “Signal transmission, calibration, compensation and transceiver link, IQ mixer, integrated circuit, sensor and equipment”, the content of which should be understood as incorporated into this application by reference. Technical Field

[0002] The embodiments of the present disclosure relate to, but are not limited to, the technical field of electromagnetic wave devices, and in particular to a signal transmission link and method, a transceiver link, an integrated circuit, and an electromagnetic wave device. Background Art

[0003] At present, the transmitting (TX) end of the frequency modulated continuous wave (FMCW) radar system uses a phase-locked loop (PLL) circuit to perform frequency sweep and then performs phase modulation through an analog phase shifter, such as a vector synthesis phase shifter and a delay line phase shifter.

[0004] However, the phase modulation precision (Resolution) and phase modulation accuracy (Accuracy) of analog phase shifters are both low, which restricts the performance of the system. For example, in automotive radar, in certain specific wave transmission systems, the low accuracy of analog phase shifters will affect the system's detection of targets. In addition, analog phase shifters also have serious problems such as large area, large loss, poor stability and channel coupling. When multiple antenna phase shifters work together, the performance of the phase shifters will affect each other, further deteriorating the performance of the system.

[0005] The phase adjustment precision and accuracy of the analog phase shifter can be improved by offline calibration of the analog phase shifter. However, when the chip works in different states (temperatures), the result of offline calibration may not be accurate. In addition, for analog phase shifters, if higher precision phase shifting is to be achieved, a greater hardware cost is required. Summary of the invention

[0006] The following is a summary of the subject matter described in detail herein. This summary is not intended to limit the scope of the claims.

[0007] An embodiment of the present disclosure provides a signal transmission link, which is applied to a radar system. The signal transmission link includes: a transmitting baseband digital module, a digital-to-analog conversion module, a transmitting local oscillator and a transmitting orthogonal modulator. The digital-to-analog conversion module includes two identical digital-to-analog converters, wherein: the transmitting baseband digital module is configured to generate two orthogonal transmitting digital baseband signals, and send the two orthogonal transmitting digital baseband signals to one of the digital-to-analog converters respectively; the digital-to-analog converter is configured to convert the two orthogonal transmitting digital baseband signals into two transmitting analog baseband signals; the transmitting local oscillator is configured to provide a transmitting local oscillator signal; the transmitting orthogonal modulator is configured to perform frequency shifting and phase shifting on the transmitting local oscillator signal based on the two transmitting analog baseband signals to form a FMCW radio frequency transmission signal after a predetermined phase shift.

[0008] An embodiment of the present disclosure also provides a signal transmission link, which is applied to an electromagnetic wave transmitting device, and the signal transmission link includes a first signal source and a digital phase shift module; wherein the first signal source is configured to generate a first analog signal; and the digital phase shift module is configured to use a digital orthogonal modulation method to perform frequency shifting and / or phase shifting on the first analog signal to form an FMCW RF transmission signal.

[0009] The embodiment of the present disclosure further provides a signal transceiver link, comprising the signal transmission link as described in any embodiment of the present disclosure, and a signal reception link;

[0010] The signal receiving link includes a receiving end local oscillator, a receiving end mixer, an analog-to-digital converter, and a receiving end baseband digital module; wherein the receiving end local oscillator is configured to provide a receiving end local oscillator signal; the receiving end mixer is configured to perform a mixing operation on the received echo signal based on the receiving end local oscillator signal to obtain a receiving end analog baseband signal, wherein the echo signal is a signal formed when the signal transmitted by the signal transmitting link is reflected and / or scattered by a target object; the analog-to-digital converter is configured to convert the receiving end analog baseband signal into a receiving end digital baseband signal; the receiving end baseband digital module is configured to process the receiving end digital baseband signal to achieve target detection and / or wireless communication.

[0011] An embodiment of the present disclosure further provides an integrated circuit, comprising a signal transmission link as described in any embodiment of the present disclosure, or a signal transceiver link as described in any embodiment of the present disclosure.

[0012] An embodiment of the present disclosure also provides an electromagnetic wave device, comprising: a carrier; an integrated circuit as described in any one of the present disclosure, arranged on the carrier; an antenna, arranged on the carrier, or the antenna and the integrated circuit are integrated into a single device and arranged on the carrier; the antenna comprises a transmitting antenna and a receiving antenna; wherein the integrated circuit is connected to the antenna for transmitting electromagnetic wave signals and / or receiving electromagnetic wave signals.

[0013] An embodiment of the present disclosure also provides a device, comprising: a device body; and an electromagnetic wave device as described in any embodiment of the present disclosure, which is arranged on the device body; wherein the electromagnetic wave device is used for target detection and / or wireless communication to provide reference information for the operation of the device body.

[0014] An embodiment of the present disclosure also provides a signal transmission method, which is applied to an electromagnetic wave device having at least one signal transmission link, and the signal transmission method includes: determining the phase of the radio frequency transmission signal of each of the signal transmission links; determining the initial phase of the transmitting end digital baseband signal in each of the signal transmission links according to the phase of the radio frequency transmission signal; generating the transmitting end digital baseband signal according to the determined initial phase; converting the transmitting end digital baseband signal into a transmitting end analog baseband signal, and performing a phase shift operation on the transmitting end local oscillator signal based on the transmitting end analog baseband signal.

[0015] The embodiments of the present disclosure also provide a non-transitory computer-readable storage medium having computer-readable instructions stored thereon. When the instructions are executed by a processor, the processor executes the signal transmission method as described in any embodiment of the present disclosure.

[0016] The signal transmission link and method, transceiver link, integrated circuit and electromagnetic wave device of the embodiments of the present disclosure generate two orthogonal transmitting digital baseband signals through a transmitting baseband digital module; the digital-to-analog conversion module converts the two orthogonal transmitting digital baseband signals into two transmitting analog baseband signals; the transmitting orthogonal modulator performs frequency shifting and phase shifting operations on the transmitting local oscillator signal based on the two transmitting analog baseband signals to form a FMCW radio frequency transmission signal after a predetermined phase shift, thereby realizing a radar system with a high-precision digital phase shifting function, reducing the isolation requirements between antennas, and having the advantages of low link loss, low cost, and no need for offline calibration, and can support more flexible wave transmission schemes, such as high-performance Doppler Division Multiplexing (DDM) and Frequency Division Multiplexing (FDM), etc., and can support frequency response compensation in the digital domain.

[0017] Other aspects will become apparent upon reading and understanding the drawings and detailed description. BRIEF DESCRIPTION OF THE DRAWINGS

[0018] The accompanying drawings are used to provide a further understanding of the technical solution of the present disclosure and constitute a part of the specification. Together with the embodiments of the present disclosure, they are used to explain the technical solution of the present disclosure and do not constitute a limitation on the technical solution of the present disclosure. The shapes and sizes of the components in the accompanying drawings do not reflect the actual proportions and are only intended to illustrate the contents of the present disclosure.

[0019] Figure 1A A simplified schematic diagram of a signal transmission link simulating a phase shifter architecture;

[0020] Figure 1B for Figure 1A A simplified schematic diagram of an analog phase shifter in a signal transmission link is shown;

[0021] Figure 2 A schematic diagram of the structure of a signal transmission link provided for an exemplary embodiment of the present disclosure;

[0022] Figure 3 A waveform diagram of an FMCW transmission signal and an echo signal using sawtooth wave modulation;

[0023] Figure 4 A schematic diagram of the structure of another signal transmission link provided for an exemplary embodiment of the present disclosure;

[0024] Figure 5 A schematic diagram of a digital phase shifter architecture in a signal transmission link according to an embodiment of the present application;

[0025] Figure 6 A schematic diagram of a signal transmission link including a compensation unit in an embodiment of the present application;

[0026] Figure 7 It is a schematic diagram of calibrating and compensating a transmission link using an auxiliary receiving circuit in an embodiment of the present disclosure;

[0027] Figure 8 A schematic diagram of the structure of a signal transceiver link provided for an exemplary embodiment of the present disclosure;

[0028] Fig. 9 A schematic diagram of the structure of another signal transceiver link provided for an exemplary embodiment of the present disclosure;

[0029] Fig.10 A schematic diagram of another transceiver link in an embodiment of the present application;

[0030] Fig.11A schematic diagram of a transceiver link including TX IQ Mod, RX IQ De-Mod and LO Freq Diff in an embodiment of the present application;

[0031] Fig.12 In the embodiment of the present application, a Fig.11 The structure shown is a schematic diagram of the transceiver link combined with BIST;

[0032] Fig.13 It is a schematic diagram of a transceiver link including TX IQ Mod, BIST IQ Mod and RX IQ De-Mod in an embodiment of the present application;

[0033] Fig.14 A schematic diagram of a transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present application;

[0034] Fig.15 It is a schematic diagram of another transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present application;

[0035] Fig.16 This is a schematic diagram of the structure of a mixer in an embodiment of the present application;

[0036] Fig.17 This is a schematic diagram of the structure of a compensation unit in a transmitter in an embodiment of the present application;

[0037] Fig.18 A schematic diagram of a digital pre-compensation HD3 architecture based on a cubic module in an embodiment of the present application;

[0038] Fig.19 A schematic diagram of a digital pre-compensation HD3 architecture based on a frequency doubling waveform generator module in an embodiment of the present application;

[0039] Fig. 20 A schematic diagram of calibration compensation of a transmission link based on a digital phase shifter architecture in an embodiment of the present application;

[0040] Fig.21 A flowchart of a signal transmission method provided by an exemplary embodiment of the present disclosure. DETAILED DESCRIPTION

[0041] In order to make the purpose, technical solution and advantages of the present disclosure more clear, the embodiments of the present disclosure will be described in detail below with reference to the accompanying drawings. It should be noted that the embodiments and features in the embodiments of the present disclosure can be combined with each other arbitrarily without conflict.

[0042] Unless otherwise defined, the technical terms or scientific terms used in the embodiments of the present disclosure should be understood by people with ordinary skills in the field to which the present disclosure belongs. The "first", "second" and similar words used in the embodiments of the present disclosure do not indicate any order, quantity or importance, but are only used to distinguish different components. "Include" or "comprising" and similar words mean that the elements or objects before the word cover the elements or objects listed after the word and their equivalents, without excluding other elements or objects.

[0043] Radar is an electronic device that uses electromagnetic waves to detect targets. The radar chip transmits a beam through a signal transmission link. When the transmitted beam encounters an obstacle, the echo reflected by the obstacle is received by the receiving antenna and transmitted to the radar chip. The radar chip determines the position, distance, speed and other information of the target relative to the electromagnetic wave emission point. With the development of microelectronics and other technologies, radar has gradually been widely used, especially millimeter wave radar (such as automotive radar). Due to its small antenna size, it has been widely used in autonomous driving, smart home devices and industrial automation devices. At present, the miniaturization and integration of radar has become the current development trend.

[0044] Figure 1A This is a simplified schematic diagram of the signal transmission link that simulates the phase shifter architecture. Figure 1B for Figure 1A The following is a simplified schematic diagram of an analog phase shifter in a signal transmission link. Figure 1A As shown, for a transmission link, when the sensor transmits a signal, a signal generator 11 such as a phase-locked loop (PLL) generates a local oscillator (LO) signal (such as a frequency sweep signal in the 77 GHz band), which may be an FMCW signal; an analog phase shifter (Analog PS) 12 performs a phase shift operation on the received LO signal, and then radiates the received LO signal to a predetermined spatial region through a transmitting antenna 13 to perform operations such as target detection and measurement. Optionally, Figure 1A In the transmission chain structure shown in FIG. 1 , the corresponding analog phase shifter architecture can be shown as Figure 1B As shown, the specific phase shift principle can be shown as follows:

[0045]

[0046] in

[0047] That is, by receiving the LO signal (ie, LO IN ) are phase shifted by 0° and 90° respectively to obtain two signals (i.e., LO I and LO Q), and after being amplified by a power amplifier (PA), it is fitted into an output signal LO OUT , that is, the output signal LO OUT Compared with LO IN The input signal is phase-shifted by θ, where the value of the phase shift θ is determined by the two signals LO I The amplitude of A and LO Q is determined by the amplitude B.

[0048] In addition, the above analog phase shifter architecture can also be implemented by means of a delay line unit, that is, by using the narrow-band assumption of the signal to perform phase shifting by means of time delay. The principle is shown in the following formula:

[0049]

[0050] Where τ is the delay time of the delay line.

[0051] Since the above-mentioned analog phase shifters have low phase modulation precision (Resolution) and phase modulation accuracy (Accuracy), they cannot meet the current sensor requirements. At the same time, although the phase modulation precision and phase modulation accuracy can be improved through calibration, the analog architecture phase shifter needs to be calibrated off-line, which will greatly increase the difficulty and complexity of engineering implementation and product mass production. At the same time, analog phase shifters also have serious problems such as large area, high loss, stability and channel coupling.

[0052] Since the signal transmission link adopts the analog phase shifter architecture, it has problems such as low phase modulation precision and accuracy. Therefore, it cannot meet the high performance requirements of the vehicle radar system.

[0053] like Figure 2As shown, the embodiment of the present disclosure provides a signal transmission link, which is applied to a radar system. The signal transmission link includes: a transmitting end baseband (Baseband) digital module 201, a digital-to-analog converter (DAC) module 202, a transmitting end local oscillator 203 and a transmitting end quadrature modulator 204, wherein the DAC module 202 includes two identical digital-to-analog converters, wherein: the transmitting end baseband digital module 201 is configured to generate two orthogonal transmitting end digital The baseband signal is transmitted, and the two orthogonal transmitting digital baseband signals are respectively sent to a digital-to-analog conversion module 202; the digital-to-analog conversion module 202 is configured to convert the two orthogonal transmitting digital baseband signals into two transmitting analog baseband signals; the transmitting local oscillator 203 is configured to provide a transmitting local oscillator signal TX_LO; the transmitting orthogonal modulator 204 is configured to perform frequency shifting and phase shifting operation on the transmitting local oscillator signal TX_LO based on the two transmitting analog baseband signals, so as to form a FMCW RF transmitting signal after a predetermined phase shift.

[0054] In the disclosed embodiment, the transmitting digital baseband signal provided by the transmitting baseband digital module 201 includes preset phase information; the digital-to-analog conversion module 202 performs digital-to-analog conversion on the received transmitting digital baseband signal to convert the transmitting digital baseband signal into a transmitting analog baseband signal (without changing the phase information); the transmitting orthogonal modulator 204 mixes the received transmitting analog baseband signal with the transmitting local oscillator signal TX_LO generated by the transmitting local oscillator 203 to achieve frequency shifting of the transmitting local oscillator signal based on the transmitting analog baseband signal while performing a preset phase shift operation to form a FMCW RF transmission signal after a predetermined phase shift.

[0055] The signal transmission link of the embodiment of the present disclosure forms a digital phase shifter architecture by using a transmitting baseband digital module 201, a digital-to-analog conversion module 202 and a transmitting orthogonal modulator 204. Since the baseband signal of the architecture is generated in the digital domain, it has better orthogonality and lower side lobes, so its phase shift phase can be generated very accurately, making the phase modulation accuracy higher, thereby realizing a vehicle-mounted radar system with a high-precision digital phase shifting function, reducing the isolation requirements between antennas, and having the advantages of small link loss, low cost, and no need for offline calibration, and can support more flexible wave transmission schemes, such as high-performance Doppler division multiplexing and frequency division multiplexing, and can support frequency response compensation in the digital domain.

[0056] In the disclosed embodiment, since the transmitting baseband digital module 201 provides a digital signal, in order to further adapt the signal characteristics, the transmitting modulator is set as an orthogonal modulator (IQ Modulator), and the digital-to-analog conversion module 202 is set as an orthogonal digital-to-analog converter (IQ DAC).

[0057] In the disclosed embodiment, the transmitting local oscillator 203 may include a phase-locked loop (PLL) structure, and may provide an electromagnetic wave (eg, laser, microwave, etc.) signal.

[0058] In some exemplary embodiments, the signal transmission link further includes a power amplifier (PA) 205, wherein the power amplifier 205 is configured to amplify the power of the phase-shifted RF signal and output the amplified signal to the transmitting antenna.

[0059] In some exemplary embodiments, the signal transmission link further includes: a transmitting antenna 206, wherein: the transmitting antenna 206 is configured to radiate the amplified signal to a preset spatial area.

[0060] In the disclosed embodiment, the signal amplified by the power amplifier 205 can be radiated to a preset spatial region through the packaged integrated or external transmitting antenna 206. That is, the transmitting local oscillator 203, the digital phase shifter and the transmitting antenna 206 can be integrated into a device or can be separate components; for example, the transmitting local oscillator 203 and the digital phase shifter can be integrated into a package to form a SoC chip, while the transmitting antenna 206 can be connected through the peripheral port of the chip and formed on a carrier such as a PCB board. At the same time, in some optional embodiments, the transmitting antenna 206 can also be integrated into the chip package to form AiP or AoP, with a chip structure with a packaged antenna.

[0061] In some exemplary embodiments, the frequency bandwidth of the swept frequency signal is above 2 GHz.

[0062] The electromagnetic wave of the transmission signal emitted by the transmitting antenna of the frequency modulated continuous wave radar system is a high-frequency frequency modulated continuous wave, and the echo signal received by the receiving antenna of the frequency modulated continuous wave radar system is the electromagnetic wave reflected / scattered back by the target object. Figure 3 FIG. 1 shows a waveform diagram of an exemplary FMCW transmission signal and an echo signal. Figure 3 As shown, the frequencies of the transmitted signal and the echo signal change regularly with time. The frequency modulated continuous wave is generally sawtooth, triangle, etc. The present disclosure takes the sawtooth shape as an example for explanation. The electromagnetic wave in each frequency modulation period T is called a chirp. The frequency of each chirp signal increases linearly with time. In the embodiment of the present disclosure, the bandwidth range B of a chirp is greater than or equal to 2GHz.

[0063] In some exemplary embodiments, the transmitting end digital baseband signal is a single-tone signal, and the transmitting end local oscillator signal is a swept frequency signal.

[0064] In the disclosed embodiment, the transmitting end local oscillator 203 may be configured to provide an FMCW signal in a centimeter wave band or a millimeter wave band (such as 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.) in microwaves, and the transmitting end baseband digital module 201 may be configured to provide a single-tone transmitting end digital baseband signal at the MHz level (for example, 3 MHz to 5 MHz, such as 3 MHz, 4 MHz, 5 MHz, etc.), that is, the digital-to-analog conversion module 202 performs digital-to-analog conversion on the MHz level single-tone transmitting end digital baseband signal to obtain a single-tone transmitting end analog baseband signal in the corresponding frequency range, and the transmitting end orthogonal modulator 204 may be configured to perform up-mixing or down-mixing operations on the received millimeter wave frequency band FMCW signal based on the received single-tone transmitting end analog baseband signal, so as to realize a preset phase shift operation on the FMCW signal.

[0065] Exemplarily, the FMCW signal in the 3.1 GHz frequency band may include a swept frequency signal between 3.1 GHz and 10.6 GHz, such as 7.163-8.812 GHz; the FMCW signal in the 77 GHz frequency band may include a swept frequency signal between 76 GHz and 81 GHz, or swept frequency signals such as 76 GHz to 77 GHz, 77 GHz to 79 GHz, or 79 GHz to 81 GHz.

[0066] In some other exemplary embodiments, the digital baseband signal at the transmitting end is a swept frequency signal, and the local oscillator signal at the transmitting end is a single tone signal.

[0067] In the disclosed embodiment, the transmitting end local oscillator 203 may be configured to provide a single-tone transmitting end local oscillator signal in a centimeter wave band or a millimeter wave band (such as 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.) in microwaves, and the transmitting end baseband digital module 201 may be configured to provide a MHz (for example, 3 MHz to 5 MHz, such as 3 MHz, 4 MHz, 5 MHz, etc.) level transmitting end digital baseband FMCW signal, that is, the digital-to-analog conversion module 202 performs digital-to-analog conversion on the MHz level transmitting end digital baseband FMCW signal to obtain a transmitting end analog baseband FMCW signal in a corresponding frequency range, and the transmitting end orthogonal modulator 204 may be configured to perform up-mixing or down-mixing operations on the received single-tone transmitting end local oscillator signal in the centimeter wave band or the millimeter wave band based on the received transmitting end analog baseband FMCW signal, so as to realize the preset phase shift and frequency sweep operations on the single-tone transmitting end local oscillator signal.

[0068] Exemplarily, the single-tone transmitting local oscillator signal in the 3.1 GHz frequency band can be a single-tone analog signal in a fixed frequency band such as 3.1 GHz, 5 GHz, 6 GHz, 8 GHz, 10.6 GHz, etc.; the single-tone transmitting local oscillator signal in the 77 GHz frequency band can be a single-tone analog signal in a fixed frequency band such as 76 GHz, 77 GHz, 78 GHz, 79 GHz, 80 GHz, 81 GHz, etc.

[0069] In some exemplary embodiments, the signal transmission link also includes: a low pass filter (LPF) 207, which is arranged between the digital-to-analog conversion module 202 and the transmitting end orthogonal modulator 204, and is configured to low pass filter the transmitting end analog baseband signal output by the digital-to-analog conversion module 202 and output it to the transmitting end orthogonal modulator 204.

[0070] like Figure 2 As shown, the transmitting end baseband digital module 201 generates two orthogonal digital baseband signals, namely, an I-channel digital baseband signal and a Q-channel digital baseband signal, and sends the generated digital baseband signals to the digital-to-analog conversion module 202 (including two identical DACs, namely, IQ DACs) to obtain two analog baseband signals. Then, the two analog baseband signals are input into the low-pass filter 207 to filter out the out-of-band noise signals, and are orthogonally modulated by the transmitting end orthogonal modulator 204 to obtain modulated RF signals, and then the modulated RF signals are radiated out through the power amplifier 205 and the transmitting antenna 206.

[0071] In some exemplary embodiments, the signal transmission link may further include a direct digital frequency synthesizer (DDFS) ( Figure 2 The direct digital frequency synthesizer is not shown in the figure and is arranged between the transmitting baseband digital module 201 and the digital-to-analog conversion module 202. The direct digital frequency synthesizer can be configured to implement at least one of a variety of signal waveforms and transmission modes such as CDM (Code-Division Multiplexing), DDM (Doppler Division Multiplexing), TDM (Time-Division Multiplexing), SDM (Space Division Multiplexing), CSD (Circuit Switch Data), Digital IF (Digital Intermediate Frequency) based on the received source signal to achieve flexible configuration of signal transmission form and transmission waveform.

[0072] like Figure 4 As shown, the embodiment of the present disclosure further provides a signal transmission link, which is applied to an electromagnetic wave transmitting device. The signal transmission link includes a first signal source 41 and a digital phase shift module 42; wherein,

[0073] The first signal source 41 is configured to generate a first analog signal; and

[0074] The digital phase shift module 42 is configured to perform frequency shift and / or phase shift on the first analog signal by using a digital quadrature modulation method to form an FMCW radio frequency transmission signal.

[0075] The signal transmission link provided by the embodiment of the present disclosure includes a first signal source 41 and a digital phase shift module 42. The first signal source 41 can be configured to provide a first analog signal, and the digital phase shift module 42 can be configured to generate a phase shift signal in the digital domain. The digital phase shift module 42 can also phase-shift the first analog signal based on the generated phase shift signal to perform a preset phase shift operation on the first analog signal.

[0076] In some exemplary embodiments, the first signal source 41 may be a transmitting local oscillator, and the first analog signal may be a transmitting LO signal.

[0077] In some exemplary embodiments, the signal transmission chain further includes a power amplifier (not shown in the figure) configured to amplify the FMCW RF transmission signal.

[0078] In some exemplary embodiments, the signal transmission link further includes a transmission antenna 43 configured to radiate the power-amplified FMCW RF transmission signal to a predetermined area.

[0079] like Figure 4 As shown, in some optional embodiments, the signal transmission link may include a first signal source 41, a digital phase shift module (Digital PS) 42 and a transmitting antenna 43, that is, the first signal source 41 may be configured to provide an LO signal, and the digital phase shift module 42 may be configured to perform a preset phase shift operation on the received LO signal, so that the phase-shifted LO signal is radiated to a preset spatial area through the transmitting antenna 43. Among them, the first signal source 41 may also be an architecture including a phase-locked loop PLL, which can provide electromagnetic wave (such as laser, microwave, etc.) signals. Among them, the first signal source 41, the digital phase shift module 42 and the transmitting antenna 43 may be integrated into an integrated device, or may be separate components; for example, the first signal source 41 and the digital phase shift module 42 may be integrated into a package to form a SoC chip, and the transmitting antenna 43 may be connected through the peripheral port of the chip and formed on a carrier such as a PCB board. At the same time, in some optional embodiments, the transmitting antenna 43 may also be integrated on the chip package to form AiP or AoP, with a chip structure of a packaged antenna.

[0080] In some exemplary embodiments, the digital phase shift module 42 includes a second signal source 423, a digital-to-analog conversion module 422, and a mixer 421 connected in sequence; wherein,

[0081] The second signal source 423 is configured to generate a first digital signal;

[0082] The digital-to-analog conversion module 422 is configured to convert the first digital signal into a second analog signal;

[0083] The mixer 421 is configured to perform frequency shift and / or phase shift on the first analog signal based on the second analog signal to form an FMCW RF transmission signal.

[0084] like Figure 4 As shown, the digital phase shift module 42 in the embodiment of the present disclosure may include a mixer (Mixer) 421, a digital-to-analog conversion module (i.e., DAC) 422, and a second signal source (e.g., a digital baseband signal source Baseband) 423, etc., that is, the second signal source 423 may be configured to provide a first digital signal; the digital-to-analog conversion module 422 may be configured to perform digital-to-analog conversion on the received first digital signal to convert the first digital signal into a second analog signal; the mixer 421 may be configured to perform a mixing operation on the received second analog signal with the received first analog signal from the first signal source 41, so as to achieve a phase shift operation of setting the above-mentioned first analog signal using the first digital signal. Optionally, when the above-mentioned signal transmission link provides a swept frequency signal, such as providing an FMCW laser signal or an FMCW microwave signal, a swept frequency transmission signal may be provided based on the first signal source 41, and / or a swept frequency first digital signal may be provided based on the second signal source 423, so that after mixing by the mixer 421, a swept frequency continuous wave signal is output.

[0085] In some optional embodiments, based on Figure 4 In the structure shown, the first signal source 41 can be configured to provide an FMCW signal (i.e., a first analog signal) in a centimeter wave band or a millimeter wave band (e.g., 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.) in microwaves, and the second signal source 423 can be configured to provide a first digital signal at a MHz level (e.g., 3 MHz to 5 MHz, such as 3 MHz, 4 MHz, 5 MHz, etc.), that is, the digital-to-analog conversion module 422 performs digital-to-analog conversion on the first digital signal at the MHz level to obtain a second analog signal in a corresponding frequency range, and the mixer 421 can be configured to perform up-mixing or down-mixing operations on the received millimeter wave band FMCW signal based on the received second analog signal of a fixed frequency band, so as to realize a preset phase shift operation on the FMCW signal.

[0086] In some optional embodiments, the centimeter wave signal in the 3.1 GHz frequency band may include 3.1 GHz to 10.6 GHz, for example, 3.1 GHz, 5 GHz, 5 GHz, 6 GHz, 8 GHz, 10.6 GHz, etc.; the millimeter wave signal in the 77 GHz frequency band may include 76 GHz to 81 GHz signals, for example, swept frequency signals such as 76 GHz to 77 GHz, 77 GHz to 79 GHz, 79 GHz to 81 GHz, or fixed frequency band signals such as 76 GHz, 77 GHz, 78 GHz, 79 GHz, 80 GHz, 81 GHz, etc.

[0087] In some exemplary embodiments, the first digital signal includes two orthogonal transmitting digital baseband signals;

[0088] The second signal source 423 is a baseband digital module at the transmitting end, and the digital-to-analog conversion module 422 includes two identical digital-to-analog converters;

[0089] The transmitting baseband digital module is configured to generate two orthogonal transmitting digital baseband signals and send the two orthogonal transmitting digital baseband signals to a digital-to-analog converter respectively;

[0090] The digital-to-analog conversion module 422 is configured to convert two orthogonal transmitting digital baseband signals into two transmitting analog baseband signals.

[0091] based on Figure 4 In the structure shown, since the second signal source 423 provides the first digital signal, in order to further adapt the signal characteristics, the mixer 421 can be set to an IQ Mixer, and the digital-to-analog conversion module 422 is an IQ DAC. At the same time, the second signal source 423 can be configured to provide a digital baseband signal source (DDFS) for phase shifting and / or provide a corresponding source signal as a waveform controller (Waveform Control). Among them, DDFS is a phase-adjustable digital baseband signal source that generates a digital baseband signal.

[0092] like Figure 5As shown, the signal transmission link (TX digital phase shifter architecture) of the digital phase shifter architecture may include a digital baseband signal source (Baseband), a direct digital frequency synthesizer (Direct Digital Frequency Synthesizer, DDFS for short), an IQ digital to analog converter (Digital to Analog Convertor, DAC for short), a low-pass filter (Low-Pass Filter, LPF for short), an IQ modulator (IQ modulator / IQ Mixer), a power amplifier (Power Amplifier, PA), etc., that is, the baseband signal source is configured to provide a digital phase-shifted source signal (that is, the aforementioned first digital signal), and the direct digital frequency synthesizer may be configured to implement CDM (Code-Division Multiplexing), DDM (Doppler Division Multiplexing), TDM (Time-Division Multiplexing), SDM (Space Division Multiplexing), CSD (Circuit Switch Data), Digital IF (Digital Intermediate Multiplexing) based on the received source signal. Frequency, digital intermediate frequency) and other signal waveforms and wave transmission methods to achieve flexible configuration of signal transmission form and transmission waveform. Among them, the signal amplified by the power amplifier can be radiated to a preset space area through a transmitting antenna packaged as an integral unit or externally.

[0093] With respect to the signal transmission link of the digital phase shifter architecture in the embodiment of the present disclosure, since the digital phase shifter architecture is configured to generate a baseband signal sequence in the digital domain, and can generate an analog baseband signal (i.e., a second analog signal) through a DAC, and then modulate the transmission signal to a high frequency through an orthogonal mixer, that is, because the baseband signal of the architecture is generated in the digital domain, it has better orthogonality and lower side lobes, and therefore its phase shift phase can be generated very accurately, resulting in higher phase modulation accuracy.

[0094] In some optional embodiments, for the signal transmission link of the digital phase shifter architecture, when the RF LO frequency sweep is used to realize the transmitted signal as an FMCW signal, a compensation unit may be added to the signal transmission link to solve the problems of TX IQ imbalance (Imbalance), signal leakage (such as TX LO Leakage) and harmonic distortion (HD) caused by IQ mismatch (Mismatch). Figure 5 As shown, a TX compensation unit can be set between the TX DDFS and the IQ DAC to perform calibration and compensation operations on the signal transmission link of the digital phase shifter architecture, so as to achieve an operation to solve at least one of the above problems. Among them, the HD caused by the third-order nonlinearity of the baseband can be referred to as HD3.

[0095] In some optional embodiments, such as Figure 6 As shown, the TX compensation unit may include at least one of a TX LO leakage compensation unit, a TX IQ imbalance compensation unit, and a TX HD3 compensation unit, wherein the TX LO leakage compensation unit may be configured to compensate for signal leakage, the TX IQ imbalance compensation unit may be configured to compensate for IQ imbalance, and the TX HD3 compensation unit may be configured to compensate for the above HD3. The TX LO leakage compensation unit is configured to compensate for at least one of an IQ modulator imbalance and an IQ channel imbalance. In addition, when the compensation unit includes at least two of the TX LO leakage compensation unit, the TX IQ imbalance compensation unit, and the TX HD3 compensation unit, compensation may be performed synchronously (such as in parallel) or sequentially (such as in series) according to actual needs and signal characteristics. Figure 6 As shown, IQ compensation may be performed first, then LO compensation, and finally HD3 compensation.

[0096] In some optional embodiments, for the signal transmission link of the digital phase shifter architecture, an error correction module (TX DAC Board Error Correction) for DAC and an AWGN (additive white gaussian noise) module for Gaussian white noise may also be included, which are not shown in the figure and may be added or deleted according to actual needs. Among them, I in IQ mentioned in the embodiments of the present disclosure can be expressed as the abbreviation of In-Phase (i.e., in-phase), Q can be expressed as the abbreviation of Quadrature (i.e., orthogonal), and RF can be expressed as the abbreviation of Radio Frequency (i.e., radio frequency).

[0097] In an optional embodiment, the compensation operation for IQ Imbalance can be achieved by compensating the conjugate signal of the BB (baseband) signal to reversely cancel the image component, and this compensation operation method is not affected by the calibration method of IQ Imbalance. The compensation for LO Leakage can be achieved by adjusting the DC component (i.e., DC bias) of the two IQ paths, and the calibration method of LO Leakage has no effect on its compensation scheme. For HD3, since the third-order harmonic distortion of the orthogonal mixer V / IConverter is the main source of HD3, and the harmonic distortion will be affected by the DC bias, when both the LO Leakage and HD3 of the transmission link need to be calibrated, it is necessary to calibrate HD3 after calibrating LO Leakage to ensure the accurate performance of HD3 calibration.

[0098] In addition, the compensation methods of HD3 based on the digital pre-compensation architecture of the digital cubic module and the digital pre-compensation architecture based on the frequency doubling waveform generator module will directly affect the subsequent calibration scheme and subsequent compensation process. Specifically:

[0099] In an optional embodiment, for a digital pre-compensation architecture based on a digital cubic module, LO Leakage may be calibrated and compensated first, and then the root cause of the HD3 problem, i.e., the compensation coefficient of HD3, may be calibrated under a stable DC bias; then, IQ Imbalance may be calibrated, and after IQ Imbalance is further compensated, the third harmonic distortion may be compensated for the two IQ paths based on the results of IQ Imbalance pre-compensation.

[0100] In an optional embodiment, for a digital pre-compensation architecture based on a frequency doubling waveform generator module, after calibrating and compensating for LO Leakage, the compensation coefficient of HD3 can be calibrated, and IQImbalance can be calibrated and IQ Imbalance can be compensated under a stable DC bias; subsequently, the actual waveforms of the IQ and Q signals and the compensation coefficient of HD3 are calculated respectively through the compensation results, and the waveform information of 3x and 5x that need to be pre-compensated is calculated in reverse.

[0101] In another optional embodiment, for the digital pre-compensation architecture based on the frequency-multiplier waveform generator module, LO Leakage can be calibrated and compensated first, and then the pre-compensation coefficients of HD3 and IQ Imbalance can be calibrated simultaneously through multiple (such as three) observations, and then the pre-compensation coefficients at the HD3 mirror position can be calibrated again through observations (such as two times); finally, the 3-fold and 5-fold frequency coefficients that need to be pre-compensated are calculated back through the pre-compensation coefficients at the HD3 and HD3 mirror positions. It should be noted that the observations in the embodiments of the present disclosure are used to represent operations such as testing and comparison and analysis of different test results.

[0102] Figure 7 It is a schematic diagram of using an auxiliary receiving circuit to calibrate and compensate a transmission link in an embodiment of the present disclosure.

[0103] like Figure 7 As shown, the transmitting path (Transmitter) may include a phase shift module PS, a power amplifier PA, a power detector PD, etc. For example, the transmitting path may adopt the signal transmitting link of the digital phase shifter architecture (Digital Phase Shifter) described in any embodiment of the present disclosure. For details, please refer to the relevant figures and text descriptions, which will not be repeated here. Since the transmitting path adopts the digital phase shifter architecture, while achieving more accurate phase shifting operations, the transmitting channel can simultaneously support multiple modes such as DDM and FDM (Frequency Division Multiplexing) of multiple antennas, and can also save the calibration operation of the RF phase shifter (Phase Shifter), reduce the isolation and coupling degree in the phase shifting system, and reduce the link loss and production manufacturing cost. In addition, in response to the possible problems of TX IQ mismatch (Mismatch) and LO leakage (leakage), the transmitting path of the digital phase shifter architecture can also support RF frequency response (Frequency Response) compensation, IQ imbalance and LO leakage calibration operations in the digital domain.

[0104] For problems such as TX IQ mismatch, LO leakage, and frequency response in the transmission path, the auxiliary receiver (ARX) can be set up to perform relevant calibration and compensation operations. Figure 7 As shown, the ARX may include a mixer, TIA, LPF, HPF, IQ ADC, adder and RF calibration module (RF Calib) connected in sequence, that is, by receiving the ARX IQ LO signal at one input end of the mixer, the other input end is connected to the node before the transmission path PD along the signal transmission direction (i.e. the direction of the arrow shown in the figure), or any node after the phase shifter (module). For example, it is connected to the output end of the PA (synchronously realizing the calibration of the PA), the input end of the PA, etc., so as to perform calibration operations on the transmission path through the ARX. There is a set difference frequency between the LO signal frequency in the transmission path and the ARX IQ LO signal frequency, so that there is a frequency error between the two signals to simulate a real transceiver signal loop.

[0105] In an optional embodiment, in order to further improve the calibration accuracy, a corresponding calibration circuit (ie, calibration receiving unit) may be provided for the ARX (ie, auxiliary receiving unit), for example Figure 7 The RF single tone signal generating circuit (RFToneGenerator) shown may include a TX DDFS, an adder, a Real DAC, an LPF, an amplifier, a multiplier, and a band pass filter (BPF) connected in sequence, wherein the adder may be configured to calibrate and compensate for the TX LO leakage (TXLOleakage Waveform), the multiplier may be configured to compensate for the RF Tone Gen LO leakage (leakage), and the BPF may be configured to filter out the DC signal generated by the LO leakage of the RF Tone Generator. That is, the RF Tone Generator may be configured to generate a plurality of stable single tone signals of different frequencies to implement the calibration operation of the ARX.

[0106] In some optional embodiments, such as Figure 7 As shown, the ARX can be calibrated using the RF Tone Generator first, and then the calibrated ARX can be used to calibrate the transmitting channel (Transmitter) including PA, for example, the PD at the PA output, the phase shifter in the transmitting channel, the total gain from DAC to PA output, and the frequency response and other devices and circuits.

[0107] Specifically, Figure 7As shown, the RF Tone Generator can be used to generate multiple stable single-tone signals with different frequencies to assist in calibrating the ARX, and then the calibrated ARX can be used to assist in calibrating the IQ imbalance, local oscillator leakage, frequency response inconsistency and other issues of the transmission path TX.

[0108] The embodiment of the present disclosure also provides a signal transceiver link, including a signal transmission link and a signal reception link, such as Figure 8 or Fig. 9 As shown, the signal transmission link may include: a transmitting baseband digital module 201, a digital-to-analog conversion module 202, a transmitting local oscillator 203 and a transmitting orthogonal modulator 204, wherein: the transmitting baseband digital module 201 is configured to generate two orthogonal transmitting digital baseband signals, and send the generated transmitting digital baseband signals to the digital-to-analog conversion module 202; the digital-to-analog conversion module 202 is configured to convert the transmitting digital baseband signal into a transmitting analog baseband signal; the transmitting local oscillator 203 is configured to provide a transmitting local oscillator signal TX_LO; the transmitting orthogonal modulator 204 is configured to perform a phase shift operation on the transmitting local oscillator signal TX_LO based on the transmitting analog baseband signal to obtain a phase-shifted RF signal.

[0109] The signal receiving link may include a receiving end local oscillator 302, a receiving end mixer 303, an analog-to-digital converter (ADC) 304 and a receiving end baseband digital module 305; wherein the receiving end local oscillator 302 is configured to provide a receiving end local oscillator signal; the receiving end mixer 303 is configured to perform a mixing operation on the received echo signal based on the receiving end local oscillator signal to obtain a receiving end analog baseband signal; the analog-to-digital converter 304 is configured to convert the receiving end analog baseband signal into a receiving end digital baseband signal; the receiving end baseband digital module 305 is configured to process the receiving end digital baseband signal to achieve target detection and / or wireless communication, for example, to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics.

[0110] In the embodiment of the present disclosure, the two ideal I-channel digital baseband signals and Q-channel digital baseband signals generated by the transmitting end baseband digital module 201 can be converted into very ideal complex signals after passing through the digital-to-analog conversion module 202, and the phase of the complex signals can be accurately controlled by the transmitting end baseband digital module 201. Figure 8 or Fig. 9 The receiver structure in the signal transceiver link shown can effectively obtain the phase information of the radio frequency signal of the signal transmission link, thereby realizing phase modulation of multiple antennas.

[0111] In some exemplary embodiments, the signal transmission link may further include: a power amplifier 205, wherein the power amplifier 205 is configured to amplify the power of the phase-shifted radio frequency signal and output the amplified signal to the transmission antenna.

[0112] In some exemplary embodiments, the signal transmission link may further include: a transmission antenna 206, wherein the transmission antenna 206 is configured to radiate the amplified signal to a preset spatial area.

[0113] In some exemplary embodiments, the signal receiving link may further include a receiving antenna 301, wherein the receiving antenna 301 is configured to receive an echo signal, where the echo signal is a signal formed when the signal transmitted by the signal transmitting link is reflected and / or scattered by a target object.

[0114] In some exemplary embodiments, the local oscillator signal at the receiving end may be a swept frequency signal, or the local oscillator signal at the receiving end may be a single tone signal.

[0115] In the embodiment of the present disclosure, in the above signal transceiver link, the frequency of the TX-LO signal received by the transmitting end orthogonal modulator 204 in the signal transmission link and the frequency of the RX-LO signal received by the receiving end mixer 303 in the signal receiving link may be the same. For example, assuming that the signal output by the transmitting end baseband digital module 201 is a sine wave of x MHz, the TX-LO signal and the RX-LO signal may both be sine waves of z GHz, wherein x and z are both positive numbers, generally between 0 and 1000.

[0116] Below through Figure 8 The signal transceiver link shown introduces the principle of the present disclosure. As mentioned above, for the FMCW radar system, the signal transmission link can have two wave transmission schemes: 1) the transmitting end local oscillator signal is swept, and the transmitting end digital baseband signal is single-tone; 2) the transmitting end local oscillator signal is single-tone, and the transmitting end digital baseband signal is swept. Assuming that the transmitting end local oscillator signal TX_LO, the transmitting end digital baseband signal, and the modulated transmission signal are represented by TLO(t), BB(t) and TX(t), respectively, and the subscripts I and q are used to represent the I-path signal and the Q-path signal, and the superscript a is used to represent its complex signal form, then under the two wave transmission schemes, the signals at each stage of the signal transmission link can be represented as follows:

[0117] 1) The local oscillator signal at the transmitting end is swept, and the digital baseband signal at the transmitting end is single tone

[0118]

[0119]

[0120]

[0121]

[0122] 2) Single tone of local oscillator signal at the transmitting end, frequency sweep of digital baseband signal at the transmitting end

[0123]

[0124]

[0125]

[0126]

[0127] in, is the sweep slope of the FMCW signal, f bb is the starting frequency of the transmitting digital baseband signal, f tlo is the starting frequency of the transmitting local oscillator signal, is the initial phase of the transmitting digital baseband signal, is the initial phase of the local oscillator signal at the transmitting end.

[0128] Assumptions Figure 8 The receiving end local oscillator signal RX_LO in the signal receiving link shown is represented by RLO(t). In this embodiment, assuming that the receiving end local oscillator signal RX_LO is a swept frequency signal, RLO(t) can be expressed as:

[0129]

[0130] Among them, f rlo is the starting frequency of the local oscillator signal at the receiving end, Φ 0 is the initial phase of the local oscillator signal at the receiving end. Then, the RF signal transmitted by the signal transmission link generates an echo signal after being reflected / scattered by the target. After passing through the receiving antenna 301, the receiving mixer 303 and the receiving low-pass filter 307, it can be obtained:

[0131]

[0132] Wherein, τ represents the time delay of the RF signal transmitted by the signal transmission link and returned to the signal receiving link after being reflected / scattered by the target.

[0133] The above formula shows that both of the above two frequency sweeping methods can be used to control the initial phase of the digital baseband signal at the transmitting end. To perform precise phase modulation, thereby achieving high-precision and high-accuracy phase shifting, avoiding direct phase shifting at high frequencies.

[0134] In some optional embodiments, the receiving antenna 301 can be connected through the peripheral port of the chip and formed on a carrier such as a PCB board. At the same time, in other optional embodiments, the receiving antenna can also be integrated on the chip package to form AiP or AoP, that is, a chip structure with a packaged antenna.

[0135] In some exemplary embodiments, the signal receiving link may further include a low noise amplifier (LNA) 306 disposed between the receiving antenna 301 and the receiving end mixer 303 , which performs low noise amplification on the echo signal received by the receiving antenna 301 and then sends it to the receiving end mixer 303 .

[0136] In some exemplary embodiments, the signal receiving link may further include a low-pass filter (LPF) 307 and a high-pass filter (HPF) 308 connected in series, which are arranged between the receiving mixer 303 and the analog-to-digital converter 304. The low-pass filter 307 and the high-pass filter 308 form a bandpass filter for filtering out-of-band noise.

[0137] In some exemplary embodiments, Figure 8 As shown, in the signal receiving link, the receiving end mixer 303 may be a real mixer, and the analog-to-digital converter 304 may be a real analog-to-digital converter.

[0138] In the disclosed embodiment, although the signal transmission link adopts a digital phase-shifting architecture, the signal receiving link may include a receiver of an orthogonal receiving architecture or a non-orthogonal receiving architecture. Therefore, it can effectively be compatible with sensors of receiving links of various architectures, effectively reducing the development cost of the entire transceiver link system.

[0139] In other exemplary embodiments, Fig. 9 As shown, in the signal receiving link, the receiving end mixer 303 may be an orthogonal mixer, and the analog-to-digital converter 304 may be an orthogonal analog-to-digital converter.

[0140] In order to match the signal transmission link of the digital phase shifter architecture, the embodiment of the present disclosure adjusts the receiving mixer 303 in the signal receiving link to an IQ demodulator (IQ Demodulator), and adjusts the analog-to-digital converter 304 to an IQ ADC. The echo signal received by the receiving antenna is processed by the above-mentioned low-noise amplifier 306, the receiving mixer 303, the low-pass filter 307, the high-pass filter 308 and the analog-to-digital converter 304 in sequence, and then converted into an IQ digital baseband signal. The subsequent receiving baseband digital module 305 processes the IQ digital baseband signal to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics (i.e., micro-Doppler).

[0141] When the receiving end mixer 303 is an orthogonal mixer and the receiving end local oscillator signal is a swept frequency signal, the receiving end local oscillator signal RX_LO can be expressed as:

[0142]

[0143] In other exemplary embodiments, the local oscillator signal at the receiving end of the signal receiving link may be a single tone signal as shown in formula (12) or formula (13) in addition to being a frequency sweep signal as shown in formula (9) or formula (11).

[0144] The receiving mixer 303 is a real mixer: RLO(t)=cos(2πf rlo t+Φ 0 )(12).

[0145] The receiving end mixer 303 is an orthogonal mixer:

[0146] When the local oscillator signal at the receiving end is a single-tone signal, it is only necessary to add a digital domain signal processing flow (including a digital domain mixing operation) in the receiving end baseband digital module 305 to obtain the required digital baseband signal.

[0147] In summary, the embodiments of the present disclosure can expand various system-level technical solutions according to the combination of different transmission schemes and receiving schemes (for example, whether the transmitting end uses a digital baseband signal single tone, a local oscillator signal sweep frequency or a digital baseband signal sweep frequency, a local oscillator signal single tone; whether the receiving end uses a real mixer, a real analog-to-digital converter or an orthogonal mixer, an orthogonal analog-to-digital converter; whether the receiving end uses a single-tone local oscillator signal or a swept frequency local oscillator signal).

[0148] Fig.10 is a schematic diagram of another transceiver link in an embodiment of the present application, Fig.11 Schematic diagram of a transceiver link including TXIQ Mod, RX IQ De-Mod and LO Freq Diff in an embodiment of the present application, Fig.12 In the embodiment of the present application, a Fig.11 The structure shown is combined with the schematic diagram of the BIST transceiver link. Fig.13 It is a schematic diagram of a transceiver link including TXIQMod, BIST IQ Mod and RX IQ De-Mod in an embodiment of the present application.

[0149] The following is a description of the transmit and receive links formed by the transmit link structure described in the embodiments of the present application:

[0150] like Fig.10As shown, a transceiver link may include a transmitting link and a receiving link, etc. The transmitting link (i.e., a transmitter) may include a digital baseband signal source (Baseband), a direct digital frequency synthesizer (TXDDFS), an IQ digital-to-analog converter (IQ DAC), a low-pass filter (LPF), an IQ modulator (IQ Modulator), a power amplifier (PA), etc., which are connected in sequence. At the same time, the signal amplified by the power amplifier is radiated to a preset spatial area through a transmitting antenna. The receiving link may include a low noise amplifier (LNA), a real mixer (Real Mixer), a trans-impedance amplifier (TIA), a low-pass filter (LPF), a high-pass filter (HPF), a real digital-to-analog converter (Real ADC), etc. connected in sequence. That is, the echo signal received by the receiving antenna is processed by the above-mentioned LNA, Real Mixer, TIA, LPF, HPF and Real ADC in sequence and converted into a real digital baseband signal. The subsequent digital signal processing module processes the real digital baseband signal to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics.

[0151] In the above transceiver link, the frequency of the TX-LO signal received by the IQ modulator in the transmitting link and the frequency of the RX-LO signal received by the Real Mixer in the receiving link may be the same. Fig.10 As shown, if the signal output by the Baseband is a sine wave of x MHz, then the TX-LO signal and the RX-LO signal can both be sine waves of z GHz.

[0152] exist Fig.10 In the embodiment shown, the transmitting link adopts a digital phase-shifting architecture, while the receiving link can adopt components of an analog architecture, that is, no IQ components are used, so it can effectively be compatible with sensors of the receiving link of the analog architecture, effectively reducing the development cost of the entire transceiver link system.

[0153] Optionally, in an embodiment of the present application, the receiving link may include a receiving antenna, that is, the receiving antenna may be connected through a peripheral port of the chip and formed on a carrier such as a PCB board. At the same time, in some optional embodiments, the receiving antenna may also be integrated on the chip package to form AiP or AoP, that is, a chip structure with a packaged antenna.

[0154] In some optional embodiments, in order to match the transmission link of the digital phase shifter architecture, relevant adjustments may be made to the receiving link, such as Fig.11 The transceiver link shown may include Fig.10 The similar transmission link architecture and receiving link in the example (in order to avoid redundancy, the same parts will not be elaborated here), Fig.10 The Real Mixer in the receiving link is adjusted to an IQ demodulator (IQ Demodulator), and the Real ADC is adjusted to an IQ ADC. That is, the receiving link may include a low noise amplifier (LNA), an IQ demodulator (IQ Demodulator), a transimpedance amplifier (TIA), a low pass filter (LPF), a high pass filter (HPF), an IQ digital-to-analog converter (IQADC), etc., which are connected in sequence. That is, the echo signal received by the receiving antenna is processed by the above-mentioned LNA, IQ Demodulator, TIA, LPF, HPF and IQ ADC in sequence and converted into an IQ digital baseband signal. The subsequent digital signal processing module processes the IQ digital baseband signal to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics (i.e., micro-Doppler).

[0155] At the same time, based on Fig.11 When the transceiver link shown in the figure is self-calibrated, it is only necessary to directly connect the signal output port of the transmission link with the signal input port of the receiving link through a transmission line, that is, the transmission link directly sends the transmission signal to the receiving link through the transmission line, so as to realize the self-calibration operation of the receiving and / or transmission link without passing through the transmitting antenna and the receiving antenna. At this time, there is a certain frequency deviation between the TX-LO signal received by the IQ modulator in the transmission link and the RX-LO signal received by the IQ Demodulator in the receiving link. For example, Fig.11 As shown, if the signal output by the Baseband is a sine wave of x MHz, the TX-LO signal can be a sine wave of z GHz. At this time, the RX-LO signal is converted into a digital signal after being processed by a down-mixer (i.e., an IQ Demodulator in the receiving link), a low-pass filter, and a high-pass filter, so as to perform TX IQ imbalance calibration.

[0156] In some optional embodiments, such as Fig.11As shown, the transmitting link (the Transmitter, TX shown in the figure) can be calibrated by adding a receiving link (the Receiver, RX shown in the figure), and the TXIQ imbalance compensation unit in the transmitting link performs compensation based on the calibrated data. At the same time, the transmitting link (the Transmitter, TX shown in the figure) can also be calibrated by multiplexing the receiving link (the Receiver, RX shown in the figure) actually used for signal transmission and reception, and the TX IQ imbalance compensation unit in the transmitting link and / or receiving link performs compensation based on the calibrated data. Among them, Fig.11 This is the signal chain for TX IQ imbalance. Since the signal chain for TX harmonic signal problem is similar to the signal chain for TX IQ imbalance, Fig.11 The TX IQ imbalance compensation unit in the embodiment is replaced by a TX HD3 compensation unit to solve the problem of TX HD3. Similar implementations can also be performed in other embodiments, which will not be described in detail in the following for the sake of simplicity.

[0157] In some optional embodiments, based on Fig.11 Based on the structure shown, in order to achieve accurate calibration of the transceiver link, Fig.11 The RX-LO port of the IQ Demodulator of the receiving link shown in the figure is provided with a built-in self-test module (BIST for short), that is, Fig.12 As shown in Fig.11 On the basis of the transceiver link structure shown, an IQ BIST architecture is set at the RX-LO port of the IQ Demodulator of the receiving link to realize inputting an LO signal with a preset frequency deviation at the RX-LO port of the IQ Demodulator of the receiving link. For example, an IQ BIST composed of a phase angle converter and an IQ modulator (IQ Modulator) is used to form a frequency-shifted signal based on the frequency deviation signal of another input signal BIST-LO of the IQ modulator, such as a TX-LO signal, through a phase angle converter through an IQ modulator, and then input to the RX-LO port of the IQ Demodulator. For example, if the TX-LO signal is a zGHz sine wave, and the BIST-LO signal is a y MHz sine wave, the frequency-shifted signal input to the RX-LO port of the IQ Demodulator is (zGHz-yMHz). It should be noted that, between different embodiments, x, y, and z are all schematic values, and the specific values ​​may be the same or different.

[0158] In some optional embodiments, based on Fig.12The IQ BIST architecture of the transceiver link structure shown in the figure can also calibrate the transmit link of the digital phase shifter architecture by multiplexing the receive link in the transceiver link; wherein, in other embodiments, the calibration operation of the transmit link using the receive link, and the calibration operation of the receive link using the transmit link, can be implemented by multiplexing the corresponding receive link or transmit link in the link that actually performs signal transmission and reception, and can also be implemented by adding a corresponding calibration receive link or calibration transmit link to implement the calibration operation of the corresponding transmit link or receive link in the link that actually performs signal transmission and reception.

[0159] Optionally, the IQBIST may include a phase angle converter and an IQ modulator (IQ Modulator), the phase angle converter is used to realize the calibration of the I path and Q path in the transmission link of the digital architecture respectively, and the other input signal BIST-LO of the IQ modulator may be a y MHz sine wave (sine wave), which is used to simulate the characteristics related to the echo signal formed by the reflection of the transmission signal by the target, wherein Fig.12 Where x, y, and z are all positive numbers, and x≠y≠z, and can generally be between 0 and 1000.

[0160] Optional, in Fig.12 In the transceiver link shown, a TX IQ imbalance compensation unit (TX IQ Imbalance Compensation) may also be provided in the transmitting link (e.g., between the TXDDFS and the IQ DAC), and / or a TX IQ imbalance compensation unit (TX IQ Imbalance Compensation) may be provided in the receiving link (e.g., after the Real ADC), so that the transmitted and / or received signals may be supplemented based on the calibration parameters (or coefficients) obtained by the above self-calibration operation to solve problems such as IQ imbalance.

[0161] In some optional embodiments, based on Fig.12 The structure shown, such as Fig.13 As shown, the above-mentioned IQ BIST module can be set between the signal output port of the transmitting link and the signal input port of the receiving link, that is, the transmitting link directly sends the transmitting signal to the receiving link through the IQBIST module, so as to realize the self-calibration operation of the receiving link and / or the transmitting link without passing through the transmitting antenna and the receiving antenna.

[0162] It should be noted that in Figures 11 to 13In the transmission chain structure shown, only an IQ compensation unit (TXIQImbalance compensation) is illustrated. In actual applications, an LO compensation unit (TXLO leakage compensation) and an HD3 compensation unit (TXHD3compensation) may be added to the transmission chain based on actual needs to form a compensation unit (TXcompensation) including an LO compensation unit (TXLO leakage compensation), an IQ compensation unit (TXIQImbalancecompensation) and / or an HD3 compensation unit (TXHD3 compensation).

[0163] Fig.14 Schematic diagram of a transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present application, Fig.15 Schematic diagram of another transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present application.

[0164] like Fig.14 As shown, a transceiver link, combined with Fig. 9 and Fig.13The structure and related description shown in the figure, the transceiver link may include a transmission link, a receiving link and a calibration link. The transmission link may include a TX digital baseband signal source (TX Baseband), a direct digital frequency synthesizer (TX DDFS), a compensation unit (Compensation), an IQ digital-to-analog converter (IQDAC), a low-pass filter (LPF), an IQ modulator (IQ Modulator) and a power amplifier (PA) connected in sequence, and the signal amplified by the power amplifier is radiated to a preset spatial area through a transmitting antenna. The receiving link may include a low noise amplifier (LNA), a real mixer (RealMixer), a trans-impedance amplifier (TIA), a high-pass filter (HPF), a variable gain amplifier (VGA), a real digital-to-analog converter (RealADC) and an RX Baseband for TXRF calibration, etc., which are connected in sequence. That is, the echo signal received by the receiving antenna is processed by the above-mentioned LNA, Real Mixer, TIA, HPF, VGA and Real ADC in sequence and converted into a real digital baseband signal. The subsequent digital signal processing module processes the real digital baseband signal to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics.

[0165] For the transmit link, the compensation unit (TX compensation) disposed between the TX DDFS and the IQ DAC may include an LO compensation unit (TX LO leakage compensation), an IQ compensation unit (TX IQ Imbalance compensation) and an HD3 compensation unit (TX HD3 compensation) to implement compensation operations corresponding to LO leakage, IQ Imbalance and HD3 in the transmit link of the digital phase shifter architecture.

[0166] In some optional embodiments, a calibration module may be provided between the transmitting link and the receiving link, and the calibration compensation unit may be configured to multiplex the receiving link to perform operations such as calibration on the transmitting link of the above-mentioned digital phase shifter architecture. At the same time, the compensation unit may be based on the parameters or coefficients obtained by the calibration operation of the calibration module to implement compensation operations on the transmitting signal at the transmitting link end. In other embodiments, a corresponding receiving compensation unit may be provided in the receiving link simultaneously or separately, that is, at this time, the receiving compensation unit may be based on the parameters or coefficients obtained by the above-mentioned calibration operation to implement compensation for the echo signal at the receiving link end.

[0167] like Fig.14 As shown, the above-mentioned calibration module may include a BIST unit and an auxiliary circuit unit, that is, the output port of the transmitting link is connected to any node between the RealMixer and the RealADC in the receiving link through the BIST unit and the auxiliary circuit unit. For example, the IQ Modulator in the transmitting link generates a radio frequency signal of (z GHz±x MHz) based on the digital phase-shifted baseband signal of x MHz and the LO signal of zGHz, and then outputs it to the BIST unit through the output port. The BIST unit performs a frequency deviation operation of y MHz on the received radio frequency signal to obtain an analog echo signal of (z GHz±x MHz±y MHz), and then uses the IQ De-Modulator in the auxiliary unit to reduce the frequency to obtain a preset intermediate frequency signal (z GHz±x MHz±y MHz-z GHz=±x MHz±y MHz). The intermediate frequency signal is then input to a preset node in the receiving link to implement the calibration operation in the transmitting link.

[0168] Optionally, the auxiliary circuit unit may be an orthogonal demodulator circuit, and the output end of the auxiliary circuit unit may be connected to any node among the node between the TIA and the HPF, the node between the HPF and the VGA, and the node between the VGA and the Real ADC in the receiving link. In addition, in order to maximize the reuse of the receiving link structure, after the output port of a transmitting link passes through the BIST unit and the auxiliary circuit unit, the I and Q branches may be respectively connected to different transmitting links, that is, Fig.14As shown, one transmission link is calibrated by multiplexing two receiving links. After the calibration of the transmission link is completed, the compensation units such as the LO compensation unit (TX LO leakage compensation), the IQ compensation unit (TX IQ Imbalance compensation) and / or the HD3 compensation unit (TX HD3 compensation) in the compensation module (TX compensation) can be used to implement compensation operations corresponding to the LO leakage, IQ Imbalance and HD3 problems in the transmission link of the digital phase shifter architecture based on the parameters obtained by the calibration.

[0169] In some optional embodiments, the BIST unit may include a phase angle converter and an IQ modulator (IQ Modulator) connected in sequence, and the auxiliary circuit unit may include an LNA, an IQ De-Modulator and a TIA connected in sequence, that is, the phase angle converter receives the RF signal output from the transmission link, and one input end of the IQ Modulator is connected to the output end of the phase angle converter, and the other input end receives the BIST-LO signal of y MHz to generate a preset echo signal. The LNA sends the received echo signal to an input end of the IQ De-Modulator after amplification, and the other input end of the IQ De-Modulator is used to receive the RX-LO signal of z GHz. The two output branches (i.e., the I branch and the Q branch) of the IQ De-Modulator are respectively connected to the corresponding nodes in the corresponding receiving links after the TIA, so as to output the generated preset intermediate frequency signal to the two receiving links, thereby realizing a more efficient multiplexing receiving link design while realizing the calibration operation.

[0170] It should be noted that, for the calibration operation in the embodiment of the present application, if the transmitting link transmits a swept frequency signal, in the actual calibration operation, the TX LO signal can be used as a single tone signal for point-by-point calibration; at the same time, the TXLO signal can also be used as a swept frequency signal for large bandwidth calibration operations, and even the swept frequency bandwidth calibration can be used once to implement the calibration operation for the swept frequency signal of the entire frequency band.

[0171] based on Fig.14 The structure shown in the figure can be realized by cascading at least two BIST units in order to further suppress the preset degree of HD3, LOLeakage, IQ Imbalance, etc. in the transmission link of the digital phase shifter architecture; Fig.15As shown, by two BIST units connected in series, the noise caused by the above defects can be suppressed to a level of -50 dB, thereby effectively reducing the difficulty of development and design of related link analog devices.

[0172] In some optional embodiments, based on the transmission link of the digital phase shifter architecture recorded in the embodiments of the present application, when calibrating and compensating for IQ Imbalance, the compensation coefficient of IQ Imbalance can be obtained in the time domain (Time-Domain) based on spectrum analysis, and the compensation coefficient of IQImbalance can be obtained in the frequency domain (Frequency-Domain) based on the spectrum peak ratio.

[0173] In some optional embodiments, in order to further improve the accuracy of the IQ Imbalance compensation coefficient, the ideal compensation coefficient may be approached by iterative calibration and compensation, or the ideal compensation coefficient may be obtained by multi-observation calibration and compensation.

[0174] For example, for the iterative calibration and compensation method, it can be determined whether to stop the iterative operation based on the size relationship between the compensation coefficients of the two calibration compensations, or whether the difference between the compensation coefficients of the two calibration compensations meets the preset iterative conditions, and the compensation coefficient obtained when the iterative operation is stopped is used as the final compensation coefficient in the current scene for subsequent operations. For the multi-observation calibration and compensation method, after multiple (for example, three) calibration and compensation operations, the measurement data obtained from each operation can be subjected to FFT (fast Fourier transform) and the corresponding amplitude and phase information can be obtained. The measurement can be subtracted and normalized to obtain relevant data, and an observation matrix can be constructed; the corresponding compensation coefficient can be reversely solved based on the data obtained by inverting the observation matrix.

[0175] In some optional embodiments, based on the similar idea of ​​obtaining the IQ Imbalance compensation coefficient, the compensation coefficient of LO leakage and / or HD3 may be obtained by adopting a method such as iterative calibration and compensation, or a method of multiple observation calibration and compensation.

[0176] In the above examples of transmitters with compensation units, the problem of solving harmonic distortion in the transmitter is mentioned. It is found through research that some harmonic distortion may be caused by components including nonlinear characteristics such as mixers in the transmitter.

[0177] by Fig.16 For example, Fig.16 Schematic diagram of the structure of the mixer 421 in the embodiment of the present application. Fig.16As shown, the mixer 421 includes a voltage-current converter (V / I Converter), a current switch (Current Switch) and a current-voltage converter (I / V Converter). The voltage-current converter converts the received voltage signal into a current signal; the current switch is connected to the voltage-current converter and the second signal generator, and is used to process the current signal output by the voltage-current converter using the local oscillator signal; the current-voltage converter is connected to the current switch, and is used to convert the current signal output by the current switch into a voltage signal.

[0178] In the above structure, since the voltage-current converter is provided with a transistor amplifier, based on the nonlinear characteristics of the transistor amplifier and the low frequency characteristics of the baseband signal, the current signal output by the voltage-current converter contains a harmonic signal corresponding to the baseband signal. For example, the HD caused by the third-order nonlinearity of the baseband can be referred to as HD3. Similarly, the harmonic caused by the fifth-order nonlinearity is called HD5. When the current switch processes the current signal output by the voltage-current converter, the frequency of the harmonic is converted to the RF band after up-conversion. Since the operation complexity of suppressing the harmonic signal in the RF band is high, the hardware cost is high. If the harmonic signal in the RF band is not removed, it will affect the signal quality of the radar reception and transmission, and thus affect the accuracy of the radar measurement.

[0179] The compensation unit is used to input the generated cancellation signal into the signal transmission link to cancel the harmonic signal in the radio frequency signal. The compensation unit is independent of the first signal generator.

[0180] To this end, the compensation unit may include a cancellation signal generator. The cancellation signal output by the compensation unit can suppress the harmonic signal in the radio frequency signal, reduce the harmonic component in the radio frequency signal, and thus improve the signal quality of the radio frequency signal output by the transmitter.

[0181] In the embodiment of the present application, for the harmonic signal in the signal transmission link, the compensation unit uses feedback or according to the characteristics of the transmission wave to input the generated cancellation signal into the signal transmission link to cancel the harmonic signal in the RF signal output by the signal transmission link. The cancellation signal has the characteristics of opposite phase and similar amplitude to the harmonic signal transmitted in the RF transmission circuit, so as to achieve the purpose of suppressing the harmonic signal.

[0182] In some examples, the compensation unit generates a compensation signal including a cancellation effect according to parameters such as the phase, frequency, or amplitude of the baseband signal generated by the first signal generator, or even the path length combined with the LO signal.

[0183] For example, Figure 5An example of a transmitter in which a compensation unit is connected to a signal transmission link is shown in FIG. Figure 5 In the structure shown, the compensation unit is a TX compensation unit. The TX compensation unit includes a generator (not shown) that can generate a cancellation signal according to the characteristics of the transmission wave. The cancellation signal generator can be, for example, Figure 6 The TX HD3 compensation unit shown in Figure 2.

[0184] Among them, the baseband processor ( Figure 5 The baseband frame (hereinafter referred to as the baseband frame) controls the orthogonal digital baseband signal generated by the TX DDFS. The TX compensation unit generates an orthogonal compensation signal according to the parameters of the orthogonal digital signal, combines the orthogonal compensation signal and the orthogonal digital signal, and sends them to the IQ DAC to be converted into an analog baseband signal. After LPF filtering, it is sent to the mixer (i.e., Figure 5 The IQ modulator in the RF transmitter is mixed to obtain a radio frequency signal based on the mixing of the TX LO signal and the analog baseband signal. The PA amplifies the mixed signal and outputs it through the transmitting antenna. The compensation signal cancels at least part of the harmonic signal in the RF transmitting circuit, such as the HD3 harmonic signal. Therefore, the clutter in the transmitted RF signal will be greatly reduced. The RF signal can be an FMCW signal.

[0185] In some other examples, the compensation unit generates a compensation signal according to harmonic information obtained through feedback from the radio frequency transmission circuit. Fig.17 , Fig.17 for Figure 5 The schematic diagram of the structure of the compensation unit in the transmitter is shown in FIG. Fig.17 As shown, the compensation unit includes an acquisition circuit and a cancellation signal generator.

[0186] The acquisition circuit is coupled to the RF transmission circuit, and is used to acquire the signal in the RF transmission circuit to obtain an acquisition signal. The acquisition signal (or sampling signal) can reflect the waveform information (also called harmonic parameters) in the harmonic signal, such as the phase of the main frequency signal, the phase of the harmonic signal, the frequency of the harmonic signal, the frequency of the main frequency signal, the power of the harmonic signal, the power of the main frequency signal, etc.

[0187] It should be noted that the harmonic parameters reflected by the acquisition signal are related to the information carried by the signal that can be acquired by the acquisition circuit. For example, if the acquisition circuit is a power acquisition circuit, the corresponding acquisition signal includes the power of the main frequency. For another example, if the acquisition circuit uses at least part of the circuit of the receiver, the acquisition signal reflects the phase of the main frequency signal, the phase of the harmonic signal, the frequency of the harmonic signal, the frequency of the main frequency signal, the power of the harmonic signal, the power of the main frequency signal, etc.

[0188] At least one of the above harmonic parameters can be extracted by analog circuits. For example, the power of the main frequency signal is output through a coupler and a power detector. Or the advantage of the digital circuit in the radar chip in frequency domain calculation is used to extract harmonic parameters. For example, by coupling the RF transmitting circuit, a signal identical to the signal transmitted at the coupling point is obtained as a collection signal. The collection signal carries the main frequency signal and the harmonic signal. The collection signal is converted into a digital signal by an ADC and handed over to the digital circuit for calculation in the frequency domain to obtain more harmonic parameters.

[0189] In one implementation, the input end of the acquisition circuit is connected to the output end or the signal detection end of the mixer. This method can detect the harmonic signal generated by the voltage-to-current converter and has a simplified acquisition circuit. Fig.18 , 19 In one connection mode of the structure shown, the input end of the acquisition circuit is connected to the detection end between the voltage-current converter and the current switch, and is coupled to the ADC.

[0190] In another embodiment, the input end of the acquisition circuit is connected to the RF output end or RF detection end of the RF transmission circuit. The RF output end is exemplified as the output end of the RF transmission circuit. The RF detection end is exemplified as the input end or output end of at least one PA in the RF transmission circuit. This method can acquire more accurate harmonic parameters in the RF transmission circuit, but has a more complex circuit structure.

[0191] In some chips including a BIST module, the acquisition circuit can obtain the acquisition signal through part or all of the circuits in the BIST module. Fig.19 As shown, the input end of the acquisition circuit is coupled to the RF output end, which includes a down converter, a filter, etc. in sequence, and is connected to the IQ ADC to output a digital acquisition signal. Among them, the down converter, the filter, etc. can reuse the BIST module or the receiver.

[0192] The collected signal is input to a cancellation signal generator, which is at least one circuit in the compensation unit. The cancellation signal generator is connected to the first signal generator, so that the signal received by the radio frequency transmitting circuit includes both the baseband signal and the cancellation signal.

[0193] For example, the cancellation signal generator includes the cancellation signal generator mentioned above and a digital circuit for extracting harmonic information, wherein the digital circuit for extracting harmonic information can be configured independently, or at least partially shared with the digital circuit in the radar chip.

[0194] The digital circuit for extracting harmonic information uses, for example, a digital circuit for processing difference frequency baseband signals in a radar chip to extract harmonic information such as harmonic frequency, main frequency, and main frequency power, and provides the information to a cancellation signal generator. The cancellation signal generator generates a cancellation signal according to the received parameters.

[0195] For another example, the digital circuit for extracting harmonic information extracts the main frequency amplitude in the collected signal, and calculates the harmonic amplitude based on the difference between the preset main frequency amplitude and the harmonic amplitude. The cancellation signal generator generates a harmonic signal compensation signal based on the calculated harmonic amplitude and other pre-configured harmonic parameters. Among them, the pre-configured harmonic parameters can be calculated based on the sweep frequency range, phase, etc. of the main frequency signal to be transmitted by the radar chip.

[0196] The cancellation signal generator in the cancellation signal generator can be configured independently with the first signal generator, or at least partially shared. For example, the cancellation signal generated by the cancellation signal generator is input into the first signal generator, so that the baseband signal output by the first signal generator includes the cancellation signal. The cancellation signal generator may include a third harmonic generator and a fifth harmonic generator.

[0197] For another example, the compensation unit further includes an adder, coupled to the cancellation signal generator and the first signal generator, to combine the baseband signal generated by the first signal generator and the cancellation signal generated by the cancellation signal generator. In this embodiment, the cancellation signal includes a cancellation signal Signal_HD3 generated by the third harmonic generator to cancel the third harmonic, and a cancellation signal Signal_HD5 generated by the fifth harmonic generator to cancel the fifth harmonic. The cancellation signals Signal_HD3 and Signal_HD5 and the baseband signal generated by the first signal generator are combined by the adder and output to the RF transmission circuit.

[0198] In summary, the transmitter circuit examples provided in the present application that pre-input a cancellation signal into a radio frequency transmitting circuit in a feedback manner can ensure that the radio frequency signal transmitted by the chip contains sufficiently low harmonic signals under different environments.

[0199] In order to effectively suppress harmonic signals according to the actual operating environment of the chip during use, for example, considering the impact of ambient temperature on semiconductor devices, the present application also provides a method for canceling harmonic signals in the above transmitter using a feedback mechanism, including:

[0200] Step 10: performing a collection operation on the signal in the signal transmission link to obtain a collection signal; wherein the signal transmission link is used to generate a radio frequency signal for radar detection, wherein the radio frequency signal includes a harmonic signal.

[0201] Step 20: Detect the collected signal, generate a cancellation signal for canceling the harmonic signal, and output the cancellation signal to the signal transmission link.

[0202] The method provided in the embodiment of the present application performs a collection operation on the signal in the signal transmission link to obtain a collection signal, and uses the collection signal to generate a cancellation signal, and outputs it to the signal transmission link, so as to use the cancellation signal to suppress the harmonic signal in the radio frequency signal and reduce the harmonic component in the radio frequency signal, thereby improving the signal quality of the radio frequency signal output by the transmitter, and further improving the receiving performance of the receiver for the radio frequency signal.

[0203] The following is combined with Figures 11 to 20 , take the transmitter and its working process as an example:

[0204] For example, Fig.11 An example of using a feedback mechanism to extract harmonic information from a transmitter so that a compensation unit generates a corresponding cancellation signal is shown in FIG. Fig.11 In the structure shown, the compensation unit includes a TX HD3 compensation unit as an example. The TXHD3 compensation unit generates a compensation signal according to the waveform characteristics of the received signal. The feedback mechanism can be executed in the calibration mode of the radar chip to prevent the signal transmission power of the radar chip from being weakened during normal detection.

[0205] The baseband processor in the transmitter (such as Fig.11 The baseband frame in the TX DDFS controls the orthogonal digital baseband signal generated by the TX compensation unit, and then sends it to the IQ DAC to be converted into an analog baseband signal. The analog baseband signal is mixed with an analog cancellation signal for canceling the harmonic signal in the transmission link. The analog baseband signal is filtered by the LPF and then enters the first mixer (i.e., Fig.11 The first mixer uses the TX LO to mix the received filtered signal to obtain an RF signal. The RF signal is coupled through the receiver and output to the TX HD3 calibration circuit (such as Fig.11 The TX HD3 calibration box in FIG. 1 ). The TX HD3 calibration circuit can be regarded as a digital circuit for extracting harmonic information.

[0206] In the receiver, the LNA amplifies the signal output by the transmitter and outputs it to the second mixer (i.e., Fig.11The IQ demodulator in the transmitter is used to obtain a demodulated signal, and the demodulated signal is sent to a transimpedance amplifier for amplification. The demodulated signal is then sent to the TX HD3 calibration circuit after passing through the LPF and HPF in sequence, and then through the IQ ADC for analog-to-digital conversion. The TX HD3 calibration circuit extracts the harmonic information in the transmitter from the feedback signal, and converts it into the parameters required to generate the cancellation signal through the upper controller, and provides it to the TX HD3 compensation unit. Among them, the harmonic information obtained by the TX HD3 calibration circuit includes one or more of the following parameters: the initial phase, starting frequency, cutoff frequency, frequency change duration, center frequency, etc. of the harmonic signal (or main frequency signal). The TX HD3 calibration unit or the upper controller determines the parameters used to generate the cancellation signal in the compensation unit according to the harmonic information, such as the initial phase, the frequency of the cancellation signal, the delay, etc.

[0207] It should be noted that the cancellation operations of the third harmonic and / or the fifth harmonic in the above examples may be determined according to the requirements of the transmitter.

[0208] In some optional embodiments, based on the transmitting link of the digital phase shifter architecture described in the embodiment of the present application, when calibrating and compensating the HD3, the main source of the generation of HD3 in the active mixer is the third harmonic in the nonlinearity of V / IConverter, so the HD3 can be calibrated and compensated by the following method: Fig.18 The cubic module-based compensation architecture shown in Fig.19 The compensation structure shown is based on a triple frequency waveform generator.

[0209] Fig. 20 FIG. 1 is a schematic diagram of calibration compensation of a transmission link based on a digital phase shifter architecture in an embodiment of the present application. Fig. 20 As shown, based on the relevant technical contents of the calibration and compensation operation for IQ Imbalance, LO leakage and HD3 in the embodiment of the present application: the compensation operation for IQ Imbalance can be achieved by compensating the conjugate signal of the BB (baseband) signal to reversely cancel the image component, and this compensation operation mode is not affected by the calibration mode of IQ Imbalance. The compensation for LO Leakage can be achieved by adjusting the DC component (i.e., DC bias) of the IQ two-way, and the calibration mode of LO Leakage has no effect on its compensation scheme. For HD3, since the third-order harmonic distortion of the orthogonal mixer V / I Converter is the main source of HD3, and the harmonic distortion will be affected by the DC bias, when both LO Leakage and HD3 of the transmission link need to be calibrated, it is necessary to calibrate HD3 after calibrating LO Leakage to ensure the accurate performance of HD3 calibration.

[0210] In addition, the compensation methods of HD3 based on the digital pre-compensation architecture of the digital cubic module and the digital pre-compensation architecture based on the frequency doubling waveform generator module will directly affect the subsequent calibration scheme and subsequent compensation process. Specifically:

[0211] In an optional embodiment, for a digital pre-compensation architecture based on a digital cubic module, LO Leakage may be calibrated and compensated first, and then the root cause of the HD3 problem, i.e., the compensation coefficient of HD3, may be calibrated under a stable DC bias; then, IQ Imbalance may be calibrated, and after IQ Imbalance is further compensated, the third harmonic distortion may be compensated for the two IQ paths based on the results of IQ Imbalance pre-compensation.

[0212] In an optional embodiment, for a digital pre-compensation architecture based on a frequency doubling waveform generator module, after calibrating and compensating for LO Leakage, the compensation coefficient of HD3 can be calibrated, and IQImbalance can be calibrated and IQ Imbalance can be compensated under a stable DC bias; subsequently, the third harmonic distortion is compensated for the IQ paths based on the results of IQImbalance pre-compensation according to the results after compensation.

[0213] In another optional embodiment, for a digital pre-compensation architecture based on a frequency doubling waveform generator module, after calibrating and compensating for LO Leakage, the compensation coefficient of HD3 can be calibrated, and IQImbalance can be calibrated and compensated for IQ Imbalance under a stable DC bias; subsequently, the actual waveforms of the IQ and Q signals and the compensation coefficient of HD3 are calculated respectively through the compensation results, and the waveform information of 3x and 5x that need to be pre-compensated is calculated in reverse.

[0214] In another optional embodiment, for the digital pre-compensation architecture based on the frequency doubling waveform generator module, LO Leakage can be calibrated and compensated first, and then the pre-compensation coefficients of HD3 and IQ Imbalance can be calibrated simultaneously through multiple (such as three) observations, and then the pre-compensation coefficients at the HD3 mirror position can be calibrated again through observations (such as two times); finally, the 3-fold and 5-fold frequency coefficients that need to be pre-compensated are calculated back through the pre-compensation coefficients at the HD3 and HD3 mirror positions. It should be noted that the observations in the embodiments of the present application are used to represent operations such as testing and comparative analysis of different test results.

[0215] See also Fig. 20It can be seen from the structure shown that the acquisition circuit has two acquisition branches, and the two acquisition branches can be switched dynamically; wherein, the input end of one acquisition branch is connected between the voltage-current converter and the current switch; the input end of the other acquisition branch is connected to the output end of the power amplifier, and the acquisition branch is provided with an IQ demodulator; in addition, Fig. 20 In the structure shown, the acquisition circuit is further provided with a multiplexer, wherein the input end of the multiplexer is respectively connected to the output ends of the two acquisition branches, and the output end is used to output the acquisition signal.

[0216] exist Figures 18 to 20 In the structure shown, the acquisition signal may be an analog signal, that is, the output end of the acquisition circuit is connected to the IQ ADC; or, the acquisition signal may be a digital signal, and the acquisition circuit at least includes the IQ ADC.

[0217] It should be noted that when the RF signal sent by the signal transmission link is not an orthogonal signal, Figures 18 to 20 The acquisition circuit can be completed by using non-orthogonal components. For example, a single-ended down-conversion mixer can be used to replace the IQ demodulator, and a single-ended analog-to-digital converter can be used to replace the IQ ADC.

[0218] like Fig.21 As shown, the embodiment of the present disclosure further provides a signal transmission method, which is applied to an electromagnetic wave device having at least one signal transmission link, and the signal transmission method includes:

[0219] Step 2101: determine the phase of the radio frequency transmission signal of each signal transmission link;

[0220] Step 2102: determining the initial phase of the transmitting end digital baseband signal in each signal transmission link according to the phase of the radio frequency transmission signal;

[0221] Step 2103: Generate a transmitting end digital baseband signal according to the determined initial phase;

[0222] Step 2104: convert the transmitting end digital baseband signal into a transmitting end analog baseband signal, and perform a phase shift operation on the transmitting end local oscillator signal based on the transmitting end analog baseband signal.

[0223] The signal transmission method of the embodiment of the present disclosure generates a digital baseband signal in the digital domain by using a digital phase shifter architecture, which has better orthogonality and lower sidelobes. The phase shift phase can be generated very accurately, so that the phase modulation accuracy is higher, thereby realizing a vehicle-mounted radar system with a high-precision digital phase shifting function, reducing the isolation requirements between antennas, and having the advantages of small link loss, low cost, and no need for offline calibration. It can also support more flexible wave transmission schemes, such as high-performance Doppler division multiplexing and frequency division multiplexing, and can support frequency response compensation in the digital domain.

[0224] In some exemplary embodiments, the digital baseband signal at the transmitting end is a single-tone signal, and the local oscillator signal at the transmitting end is a swept-frequency signal; or, the digital baseband signal at the transmitting end is a swept-frequency signal, and the local oscillator signal at the transmitting end is a single-tone signal.

[0225] In some exemplary embodiments, the frequency bandwidth of the swept frequency signal is above 2 GHz.

[0226] The embodiment of the present disclosure also provides an integrated circuit, which may include any of the above signal transceiver links. Optionally, the integrated circuit may be a millimeter wave radar chip (chip or die).

[0227] In some optional embodiments, the integrated circuit may be an AiP (Antenna-In-Package) chip structure, an AoP (Antenna-On-Package) chip structure, or an AoC (Antenna-On-Chip) chip structure.

[0228] According to some other embodiments of the present disclosure, an electromagnetic wave device is further proposed in the embodiments of the present disclosure. The electromagnetic wave device may include an antenna, and an integrated circuit as described above. The integrated circuit is electrically connected to the antenna for receiving and transmitting electromagnetic wave signals. For example, the electromagnetic wave device may include: a carrier, an integrated circuit and an antenna as described in any of the above embodiments, etc., wherein the integrated circuit may be arranged on the carrier; the antenna may be arranged on the carrier, or the integrated circuit may be integrated with the integrated circuit and arranged on the carrier (that is, the antenna may be an antenna provided in an AiP, AoP or AoC structure at this time); wherein the integrated circuit is connected to the antenna (that is, the sensor chip or the integrated circuit is not integrated with an antenna at this time, such as a conventional SoC, etc.), for receiving and transmitting electromagnetic wave signals. The carrier may be a printed circuit board PCB.

[0229] An embodiment of the present disclosure provides a device, which may include: a device body; and an electromagnetic wave device as described above, which is arranged on the device body; wherein the electromagnetic wave device is used for target detection and / or communication to provide reference information for the operation of the device body.

[0230] The present disclosure also provides an electronic device, which can be expressed in the form of a general-purpose computing device. The components of the electronic device may include, but are not limited to: at least one processing unit, at least one storage unit, a bus connecting different system components (including a storage unit and a processing unit), a display unit, etc. Among them, the storage unit stores a program code, and the program code can be executed by the processing unit so that the processing unit executes the method described in this specification according to various exemplary embodiments of the present disclosure. The storage unit may include a readable medium in the form of a volatile storage unit, such as a random access memory unit (RAM) and / or a cache memory unit, and may further include a read-only memory unit (ROM).

[0231] The storage unit may also include a program / utility having a set (at least one) of program modules, such program modules including but not limited to: an operating system, one or more application programs, other program modules, and program data, each of which or some combination may include an implementation of a network environment.

[0232] The bus may represent one or more of several types of bus structures, including a memory unit bus or memory unit controller, a peripheral bus, an accelerated graphics port, a processing unit, or a local bus using any of a variety of bus architectures.

[0233] The electronic device may also communicate with one or more external devices (e.g., keyboards, pointing devices, Bluetooth devices, etc.), may communicate with one or more devices that enable a user to interact with the electronic device, and / or may communicate with any device that enables the electronic device to communicate with one or more other computing devices (e.g., routers, modems, etc.). This communication may be performed through an input / output (I / O) interface. In addition, the electronic device may also communicate with one or more networks (e.g., local area networks (LANs), wide area networks (WANs), and / or public networks, such as the Internet) through a network adapter. The network adapter may communicate with other modules of the electronic device through a bus. It should be understood that, although not shown in the figure, other hardware and / or software modules may be used in conjunction with the electronic device, including but not limited to: microcode, device drivers, redundant processing units, external disk drive arrays, RAID systems, tape drives, and data backup storage systems, etc.

[0234] For example, the electronic device in the embodiment of the present disclosure may also include: a device body; and an electromagnetic wave device as described in any of the above embodiments arranged on the device body; wherein the electromagnetic wave device can be used to realize functions such as target detection and / or wireless communication.

[0235] Specifically, based on the above embodiments, in an optional embodiment of the present disclosure, the electromagnetic wave device can be arranged outside the device body, or arranged inside the device body, and in other optional embodiments of the present disclosure, the electromagnetic wave device can also be arranged partially inside the device body and partially outside the device body. The embodiments of the present disclosure are not limited to this, and the specific situation may be determined.

[0236] In an optional embodiment, the device body can be a component or product used in fields such as smart cities, smart homes, transportation, smart homes, consumer electronics, security monitoring, industrial automation, in-cabin detection (such as smart cockpits), medical devices, and health care. For example, the device body can be intelligent transportation equipment (such as cars, bicycles, motorcycles, ships, subways, trains, etc.), security equipment (such as cameras), liquid level / flow rate detection equipment, smart wearable devices (such as bracelets, glasses, etc.), smart home devices (such as sweeping robots, door locks, televisions, air conditioners, smart lights, etc.), various communication devices (such as mobile phones, tablet computers, etc.), and gates, smart traffic lights, smart signs, traffic cameras, and various industrialized robotic arms (or robots), etc. It can also be various instruments for detecting life characteristic parameters and various devices equipped with the instruments, such as life characteristic detection in car cabins, indoor personnel monitoring, smart medical equipment, consumer electronic equipment, etc.

[0237] The embodiment of the present disclosure also provides a non-transitory computer-readable storage medium on which computer-readable instructions are stored. When the instructions are executed by a processor, the processor executes the signal transmission method as described above.

[0238] Through the description of the above implementation methods, it is easy for those skilled in the art to understand that the example implementation methods described here can be implemented by software, or by combining software with necessary hardware. The technical solution according to the implementation methods of the present disclosure can be embodied in the form of a software product, which can be stored in a non-volatile storage medium (which can be a CD-ROM, a USB flash drive, a mobile hard disk, etc.) or on a network, including several instructions to enable a computing device (which can be a personal computer, a server, or a network device, etc.) to execute the above method according to the implementation methods of the present disclosure.

[0239] The software product may use any combination of one or more readable media. The readable medium may be a readable signal medium or a readable storage medium. The readable storage medium may be, for example, but not limited to, a system, device or device of electricity, magnetism, light, electromagnetic, infrared, or semiconductor, or any combination thereof. More specific examples (non-exhaustive list) of readable storage media include: an electrical connection with one or more wires, a portable disk, a hard disk, a random access memory (RAM), a read-only memory (ROM), an erasable programmable read-only memory (EPROM or flash memory), an optical fiber, a portable compact disk read-only memory (CD-ROM), an optical storage device, a magnetic storage device, or any suitable combination thereof.

[0240] Computer readable storage media may include data signals propagated in baseband or as part of a carrier wave, wherein readable program codes are carried. Such propagated data signals may take various forms, including but not limited to electromagnetic signals, optical signals, or any suitable combination thereof. The readable storage medium may also be any readable medium other than a readable storage medium, which may send, propagate, or transmit a program for use by or in conjunction with an instruction execution system, device, or device. The program codes contained on the readable storage medium may be transmitted using any suitable medium, including but not limited to wireless, wired, optical cable, RF, etc., or any suitable combination thereof.

[0241] Program code for performing the operations of the present disclosure may be written in any combination of one or more programming languages, including object-oriented programming languages ​​such as Java, C++, etc., and conventional procedural programming languages ​​such as "C" or similar programming languages. The program code may be executed entirely on the user computing device, partially on the user device, as a separate software package, partially on the user computing device and partially on a remote computing device, or entirely on a remote computing device or server. In cases involving a remote computing device, the remote computing device may be connected to the user computing device through any type of network, including a local area network (LAN) or a wide area network (WAN), or may be connected to an external computing device (e.g., through the Internet using an Internet service provider).

[0242] The computer-readable medium carries one or more programs. When the one or more programs are executed by a device, the computer-readable medium implements the aforementioned functions.

[0243] Those skilled in the art will appreciate that the above modules can be distributed in the device according to the description of the embodiment, or can be changed accordingly and only used in one or more devices different from the embodiment. The modules of the above embodiments can be combined into one module, or further divided into multiple sub-modules.

[0244] According to an embodiment of the present disclosure, a computer program is proposed, including a computer program or an instruction, which, when executed by a processor, can execute the method described above. In an optional embodiment, the above-mentioned integrated circuit can be a millimeter wave radar chip. The type of digital functional module in the integrated circuit can be determined according to actual needs. For example, in a millimeter wave radar chip, the receiving end baseband digital module can be used for functions such as distance-dimensional Doppler transform, velocity-dimensional Doppler transform, constant false alarm detection, wave direction detection, point cloud processing, etc., to obtain information such as the distance, horizontal angle, pitch angle, speed, height, micro-Doppler motion characteristics, shape, size, surface roughness and dielectric properties of the target.

[0245] It should be noted that radio devices can achieve functions such as target detection and / or communication by transmitting and receiving radio signals to provide the device body with detection target information and / or communication information, thereby assisting or even controlling the operation of the device body.

[0246] For example, when the above-mentioned device body is applied to an advanced driver assistance system (i.e., ADAS), the radio device (such as millimeter-wave radar) as a vehicle-mounted sensor can assist the ADAS system to realize application scenarios such as adaptive cruise control, automatic braking assistance (i.e., AEB), blind spot detection warning (i.e., BSD), assisted lane change warning (i.e., LCA), reversing assistance warning (i.e., RCTA), parking assistance, rear vehicle warning, collision avoidance, pedestrian detection, etc.

[0247] The technical features of the above-described embodiments may be arbitrarily combined. To make the description concise, not all possible combinations of the technical features in the above-described embodiments are described. However, as long as there is no contradiction in the combination of these technical features, they should be considered to be within the scope of this specification.

[0248] The above-described embodiments only express the preferred embodiments of the present disclosure and the technical principles used. The description is relatively specific and detailed, but it cannot be understood as limiting the scope of the invention patent. It is possible for those skilled in the art to make various obvious changes, readjustments and substitutions without departing from the protection scope of the present disclosure. Therefore, although the present disclosure is described in more detail through the above embodiments, the present disclosure is not limited to the above embodiments, and may also include more other equivalent embodiments without departing from the concept of the present disclosure, and the protection scope of the patent of the present disclosure is determined by the scope of the attached claims.

Claims

1. A signal transmission link, characterized in that: Applied in a radar system, the signal transmission link includes: a transmitting end baseband digital module, a digital-to-analog conversion module, a transmitting end local oscillator and a transmitting end orthogonal modulator, and the digital-to-analog conversion module includes two identical digital-to-analog converters, wherein: The transmitting end baseband digital module is configured to generate two orthogonal transmitting end digital baseband signals, and send the two orthogonal transmitting end digital baseband signals to one of the digital-to-analog converters respectively; The digital-to-analog conversion module is configured to convert the two orthogonal transmitting digital baseband signals into two transmitting analog baseband signals; The transmitting local oscillator is configured to provide a transmitting local oscillator signal; and The transmitting end orthogonal modulator is configured to perform frequency shifting and phase shifting operations on the transmitting end local oscillator signal based on the two transmitting end analog baseband signals to form a FMCW radio frequency transmitting signal after a predetermined phase shift.

2. The signal transmission link according to claim 1, characterized in that: The transmitting end digital baseband signal is a single tone signal, and the transmitting end local oscillator signal is a swept frequency signal; or, The transmitting end digital baseband signal is a frequency sweep signal, and the transmitting end local oscillator signal is a single tone signal.

3. The signal transmission link according to claim 2, characterized in that: The frequency bandwidth of the frequency sweep signal is above 2 GHz; and / or, The signal transmission link also includes a transmitting antenna configured to radiate the predetermined phase-shifted FMCW radio frequency transmission signal toward a predetermined area.

4. A signal transmission link, characterized in that: In the application of electromagnetic wave transmitting device, the signal transmitting link includes a first signal source and a digital phase shift module; wherein, The first signal source is configured to generate a first analog signal; and The digital phase shift module is configured to perform frequency shift and / or phase shift on the first analog signal by using a digital quadrature modulation method to form an FMCW radio frequency transmission signal.

5. The signal transmission link according to claim 4, characterized in that: The digital phase shift module includes a second signal source, a digital-to-analog conversion module and a mixer connected in sequence; wherein, The second signal source is configured to generate a first digital signal; The digital-to-analog conversion module is configured to convert the first digital signal into a second analog signal; and The mixer is configured to perform frequency shift and / or phase shift on the first analog signal based on the second analog signal to form the FMCW radio frequency transmission signal.

6. The signal transmission link according to claim 5, characterized in that: The first digital signal includes two orthogonal transmitting digital baseband signals; The second signal source is a transmitting baseband digital module, and the digital-to-analog conversion module includes two identical digital-to-analog converters; The transmitting end baseband digital module is configured to generate two orthogonal transmitting end digital baseband signals, and send the two orthogonal transmitting end digital baseband signals to one of the digital-to-analog converters respectively; The digital-to-analog conversion module is configured to convert the two orthogonal transmitting digital baseband signals into two transmitting analog baseband signals.

7. The signal transmission link according to any one of claims 4 to 6, characterized in that: The first signal source is a transmitting end local oscillator, and the first analog signal is a transmitting end intrinsic signal.

8. The signal transmission link according to claim 7, characterized in that: The system also includes a power amplifier configured to amplify the FMCW radio frequency transmission signal.

9. The signal transmission link according to claim 8, characterized in that: It also includes a transmitting antenna configured to radiate the power-amplified FMCW radio frequency transmitting signal to a predetermined area.

10. A signal transceiver link, characterized in that: A signal transmission link and a signal receiving link as claimed in any one of claims 1 to 9; The signal receiving link includes a receiving end local oscillator, a receiving end mixer, an analog-to-digital converter, and a receiving end baseband digital module; wherein, The receiving end local oscillator is configured to provide a receiving end local oscillator signal; The receiving end mixer is configured to perform a mixing operation on the received echo signal based on the receiving end local oscillator signal to obtain a receiving end analog baseband signal; the echo signal is a signal formed by the signal transmitted by the signal transmission link being reflected and / or scattered by the target object; The analog-to-digital converter is configured to convert the receiving end analog baseband signal into a receiving end digital baseband signal; The receiving end baseband digital module is configured to process the receiving end digital baseband signal to achieve target detection and / or wireless communication.

11. The signal transceiver link according to claim 10, characterized in that: The receiving mixer is a real mixer, and the analog-to-digital converter is a real analog-to-digital converter; or, The receiving-end mixer is an orthogonal mixer, and the analog-to-digital converter is an orthogonal analog-to-digital converter.

12. The signal transceiver link according to claim 10, characterized in that: The receiving end local oscillator signal is a swept frequency signal; or, the receiving end local oscillator signal is a single tone signal.

13. The signal transceiver link according to claim 10, characterized in that: The device also includes a receiving antenna, which is configured to receive the echo signal.

14. An integrated circuit, characterized in that: It comprises a signal transmission link as described in any one of claims 1 to 9, or a signal transceiver link as described in any one of claims 10 to 13.

15. The integrated circuit according to claim 14, characterized in that The integrated circuit is a millimeter wave chip.

16. An electromagnetic wave device, characterized in that: include: Carrier; An integrated circuit as claimed in any one of claims 14 to 15, arranged on a carrier; An antenna is arranged on the carrier, or the antenna and the integrated circuit are integrated into a single device and arranged on the carrier; the antenna includes a transmitting antenna and a receiving antenna; Wherein, the integrated circuit is connected to the antenna and is used for transmitting electromagnetic wave signals and / or receiving electromagnetic wave signals.

17. A user terminal device, characterized in that: include: Equipment body; as well as The electromagnetic wave device according to claim 16 is disposed on the device body; Wherein, the electromagnetic wave device is used for target detection and / or wireless communication to provide reference information for the operation of the device body.

18. A signal transmission method, characterized in that: Applied to an electromagnetic wave device having at least one signal transmission link, the signal transmission method comprises: Determining the phase of the radio frequency transmission signal of each of the signal transmission links; Determining the initial phase of the transmitting end digital baseband signal in each of the signal transmission links according to the phase of the radio frequency transmission signal; Generating the transmitting end digital baseband signal according to the determined initial phase; The transmitting end digital baseband signal is converted into a transmitting end analog baseband signal, and a phase shift operation is performed on a transmitting end local oscillator signal based on the transmitting end analog baseband signal.

19. A non-transitory computer-readable storage medium having computer-readable instructions stored thereon, which, when executed by a processor, causes the processor to execute the signal transmission method according to claim 18.

Images