Isolated DC boost converter of two-phase parallel boost circuit and control method thereof

Abstract

The present invention belongs to the field of DC converters, and specifically relates to an isolated DC boost converter of a two-phase parallel boost circuit and a control method thereof, aiming to solve the problem that the existing isolated DC converter cannot adapt to a wide voltage range, high gain, and achieve high-efficiency conversion within the entire load range. The present invention includes: two-phase input inductors respectively form a two-phase parallel Boost boost circuit with two bridge arms of a full-bridge unit to boost and stabilize the input DC voltage; a DC bus capacitor performs voltage filtering after boosting and stabilizing; a series resonant circuit resonates and transmits power from the primary side to the secondary side through a high-frequency transformer; an output rectifier and filter circuit performs high-frequency rectification and filtering of the output voltage of the high-frequency transformer to obtain an output DC voltage. The present invention introduces high-frequency series resonance with leakage inductance to eliminate voltage spikes caused by leakage inductance, and realizes soft switching of switch tubes within a wide voltage range by combining high-frequency resonance with PWM modulation, thereby improving gain and efficiency while reducing losses.

Description

Technical Field

[0001] The invention belongs to the field of direct current converters, and in particular relates to an isolated direct current boost converter of a two-phase parallel boost circuit and a control method thereof. Background Art

[0002] DC boost converters have broad application prospects in renewable energy power generation such as photovoltaics, wind power, and electric vehicles. In applications with higher voltage and power, the converter is generally required to have isolation function in consideration of system safety. And because photovoltaics, wind power, fuel cells, etc. have the characteristics of wide range of output voltage fluctuations, DC converters are required to achieve maximum power point tracking (MPPT) and output voltage regulation in a wide input voltage range.

[0003] Currently, the commonly used isolated DC boost converters based on two-phase parallel boost circuits mainly adopt push-pull topology, phase-shifted full-bridge topology, series / parallel resonant topology, full-bridge Boost topology and other structures: (1) The push-pull topology circuit uses two switches with the same parameters, working alternately in the positive and negative half-waves in a push-pull manner. Its structure is simple and the transformer core utilization rate is higher than that of the forward and flyback circuit topologies. However, there are problems such as high switch tube voltage stress (more than twice the input voltage), voltage spikes caused by transformer leakage inductance, and large ripple current. It is not suitable for applications with high voltage and high power. (2) The phase-shifted full-bridge topology adopts the phase-shifted PWM control method, which can reduce the switching loss and switching noise of the circuit through soft switching, reduce electromagnetic interference during the switching process, and at the same time maintain a series of advantages in the conventional full-bridge PWM circuit, such as simple topology structure, simple control method, constant switching frequency, and small component voltage and current stress. However, in applications with a wide voltage range, the phase-shifted full-bridge topology has problems such as limited soft switching range, loss of secondary duty cycle, and large conduction loss. It is not suitable for applications with wide voltage fluctuations such as photovoltaics, wind power, and fuel cells. (3) The series / parallel resonant topology is a basic resonant topology. It uses a half-bridge or full-bridge to generate a voltage square wave, and after passing through an LC series or parallel resonant network, it generates a sinusoidal AC current in the resonant circuit and adjusts the voltage by changing the switching frequency. It has a small switching loss and a wide range of soft switching. However, since the series / parallel resonant topology uses variable frequency control, its frequency conversion range is wide within a wide input voltage range, so it faces great challenges in circuit filtering, device selection, and transformer design. (4) The full-bridge Boost topology can achieve a higher boost ratio by combining the conventional Boost circuit with the conventional full-bridge circuit, and can adapt to a wider voltage input range. However, due to the influence of the transformer leakage inductance, the switch tube generates higher voltage and current spikes at the switching moment, affecting the reliability of the device and reducing efficiency. In addition, this topology has problems such as difficult soft start and large switch tube current peak, and is not suitable for occasions with high requirements for efficiency and power density.

[0004] In general, some of the above-mentioned topologies can achieve optimal efficiency at certain operating points after optimization. However, within a wider operating voltage range, it is difficult to achieve global optimization due to problems such as narrow soft switching range, large circulating current, and high voltage stress. In addition, the above-mentioned traditional various isolated DC boost converters based on two-phase parallel boost circuits generally have problems such as narrow voltage variation range and poor controllability, and cannot adapt to the wide range of random voltage changes of fuel cells, photovoltaic cells, etc. Summary of the invention

[0005] In order to solve the above-mentioned problems in the prior art, that is, the existing isolated DC converter cannot adapt to a wide voltage operating range, has a low switching frequency and high switching loss, resulting in low efficiency and low power density, and thus cannot meet the requirements of electric vehicles, new energy power generation and other systems for a wide voltage range, high power density and high efficiency, the present invention provides an isolated DC boost converter based on a two-phase parallel boost circuit, including an input power supply, a two-phase input inductor, a DC bus capacitor, a full-bridge unit, a series resonant circuit, a high-frequency transformer, an output rectifier bridge and a filter capacitor;

[0006] The positive electrode of the input power supply is connected to the two-phase input inductor, and the negative electrode is connected to the DC bus capacitor;

[0007] The two-phase input inductor and the DC bus capacitor are connected to the full-bridge unit together;

[0008] The full-bridge unit is connected to a series resonant circuit;

[0009] The series resonant circuit is connected in sequence to an output rectifier bridge and a filter capacitor via a high-frequency transformer.

[0010] In some preferred embodiments, the two-phase input inductor includes an A-phase input inductor L1 and a B-phase input inductor L2;

[0011] The full-bridge unit comprises a first bridge arm and a second bridge arm, wherein the first bridge arm comprises a first bridge arm upper tube S1 and a first bridge arm lower tube S2, and the second bridge arm comprises a second bridge arm upper tube S3 and a second bridge arm lower tube S4;

[0012] The series resonant circuit includes a resonant capacitor C r and resonant inductor L r ; The resonant frequency of the series resonant circuit is higher than the switching frequency of the full-bridge unit, and the voltage of the DC boost converter is adjusted by adjusting the duty cycle of the full-bridge unit;

[0013] The high-frequency transformer comprises a primary winding, a secondary winding and an excitation inductor, the primary winding comprises a first end and a second end, and the secondary winding comprises a third end and a fourth end;

[0014] The first bridge arm upper tube S1 and the first bridge arm lower tube S2 are connected to form the midpoint of the first bridge arm, and the second bridge arm upper tube S3 and the second bridge arm lower tube S4 are connected to form the midpoint of the second bridge arm;

[0015] The A-phase input inductor is connected in series between the positive electrode of the input power supply and the midpoint of the first bridge arm of the full-bridge unit; the B-phase input inductor is connected in series between the positive electrode of the input power supply and the midpoint of the second bridge arm of the full-bridge unit; the first bridge arm and the second bridge arm are connected in parallel with the DC bus capacitor;

[0016] A resonant capacitor is connected in series between the midpoint of the first bridge arm and the first end of the primary winding of the high-frequency transformer; a resonant inductor is connected in series between the midpoint of the second bridge arm and the second end of the primary winding of the high-frequency transformer;

[0017] The primary excitation inductance of the high-frequency transformer is connected to both ends of the primary winding;

[0018] An anode of a diode in the third bridge arm of the output rectifier bridge is connected to a cathode of another diode to form the midpoint of the third bridge arm; an anode of a diode in the fourth bridge arm of the output rectifier bridge is connected to a cathode of another diode to form the midpoint of the fourth bridge arm; the third end and the fourth end of the secondary winding of the high-frequency transformer are respectively connected to the midpoints of the third bridge arm and the fourth bridge arm of the output rectifier bridge; the third bridge arm and the fourth bridge arm are connected in parallel with the output filter capacitor.

[0019] In some preferred embodiments, the two-phase input inductor, the first bridge arm and the second bridge arm form a two-phase parallel Boost circuit, the current on the inductor in the Boost circuit changes linearly, and the current of the two-phase input inductor in the DC boost converter is a superposition of the linear current and the resonant current, thereby achieving the same current effective value while reducing its peak value and reducing the current stress of the power device.

[0020] In some preferred embodiments, the full-bridge unit is controlled by a pulse width modulation (PWM) method, and high-frequency resonance is combined with PWM to achieve soft switching of all switch tubes in the full-bridge unit and reduce the current peak value and effective value of all switch tubes; the DC boost converter achieves power output by applying a high-frequency resonant sinusoidal current to the primary side of the high-frequency transformer.

[0021] Another aspect of the present invention provides a control method for an isolated DC boost converter based on a two-phase parallel boost circuit, which is applied to the above-mentioned isolated DC boost converter based on a two-phase parallel boost circuit, and specifically includes:

[0022] Step S100, providing an initial voltage to the entire circuit through an input power supply;

[0023] Step S200, based on the initial voltage, boosting and stabilizing the voltage through two-phase input inductors;

[0024] Step S300, controlling the first bridge arm upper tube S1, the first bridge arm lower tube S2, the second bridge arm upper tube S3, and the second bridge arm lower tube S4 in the full-bridge unit through a set driving signal to adjust the working mode and output a boost voltage to the series resonant circuit;

[0025] At the same time, the boost voltage output by the full-bridge unit is filtered through the DC bus capacitor;

[0026] Step S400, based on the boost voltage, two resonant half-waves are completed in one switching cycle of the converter through a series resonant circuit, the switch tube voltage spike caused by the transformer leakage inductance is eliminated, and the primary voltage is output;

[0027] Step S500, converting the primary voltage into a secondary voltage through a high frequency transformer;

[0028] Step S600: After each resonant half cycle of the primary voltage is completed, the resonant action is blocked by the output rectifier bridge, and the DC supply voltage is output to the load after filtering by the filter capacitor.

[0029] In some preferred embodiments, the working mode includes:

[0030] Mode 1, D=0.5; Mode 2, D>0.5; Mode 3, D<0.5; wherein D represents the duty cycle of the main switch tube, and the main switch tubes are the first bridge arm lower tube S2 and the second bridge arm lower tube S4; in Mode 1, the converter voltage equivalent gain is 2; in Mode 2, the converter voltage equivalent gain is <2; in Mode 3, the converter voltage equivalent gain is >2.

[0031] In some preferred embodiments, the mode 1, the switching cycle includes mode 1, mode 2, mode 3, mode 4, mode 5, mode 6, mode 7 and mode 8;

[0032] Among them, in modes 1, 2 and 3, the upper tube S1 of the first bridge arm and the lower tube S4 of the second bridge arm are opened, and the lower tube S2 of the first bridge arm and the upper tube S3 of the second bridge arm are closed;

[0033] In mode 4, the first bridge arm upper tube S1, the second bridge arm lower tube S4, the first bridge arm lower tube S2 and the second bridge arm upper tube S3 are turned off;

[0034] In modes 5, 6 and 7, the lower tube S2 of the first bridge arm and the upper tube S3 of the second bridge arm are opened, and the upper tube S1 of the first bridge arm and the lower tube S4 of the second bridge arm are closed;

[0035] In mode 8, the first bridge arm upper tube S1, the second bridge arm lower tube S4, the first bridge arm lower tube S2 and the second bridge arm upper tube S3 are turned off. After mode 8 ends, mode 1 is re-entered to start a new switching cycle.

[0036] In some preferred embodiments, in Mode 1, the circuit states of each mode are:

[0037] In mode 1, the parasitic output capacitance of each switch tube resonates with the two-phase input inductance. The upper tube S1 of the first bridge arm provides a freewheeling circuit for the A-phase input inductance L1. The B-phase input inductance L2 is excited by the lower tube S4 of the second bridge arm. The resonant inductance L r and resonant capacitor C rThe first half cycle of series resonance is performed, and power is output to the secondary circuit through the high-frequency transformer. The output rectifier diodes D1 and D4 in the output rectifier bridge are turned on for high-frequency rectification and the supply voltage is output to the load. The rectifier diodes D2 and D3 remain turned off.

[0038] In mode 2, the resonant inductor L r and resonant capacitor C r The series resonance has completed the first half cycle, the resonant current is 0, and the resonant inductance L r The resonant voltage reaches the maximum value; the output rectifier diodes D1 and D4 in the output rectifier bridge are cut off;

[0039] In the mode 3, the freewheeling current of the A-phase input inductor L1 drops to 0, and the current of the B-phase input inductor L2 continues to increase linearly;

[0040] Mode 4 is the driving dead time of the switch tube. After the upper tube S1 of the first bridge arm and the lower tube S4 of the second bridge arm are turned off, due to the presence of parasitic output capacitance in each switch tube, the drain-source voltage of S1 and S4 increases slowly, and the parasitic capacitance of S1 and S4 is charged by the current of the A-phase input inductor L1 and the B-phase input inductor L2 respectively; at the same time, the current of the A-phase input inductor L1 and the B-phase input inductor L2 also discharges the parasitic capacitance of the lower tube S2 of the first bridge arm and the upper tube S3 of the second bridge arm through the effect of resonance, and the drain-source voltage of S2 and S3 decreases slowly, and the rectifier diodes D2 and D3 are turned on for high-frequency rectification; the voltage of S1 and S4 has increased to C L The voltage of S2 and S3 has dropped to 0, and the supply voltage is output to the load;

[0041] The modality 5, C L The voltage is applied to the resonant cavity and the high-frequency transformer excitation inductor L through the first bridge arm lower tube S2 and the second bridge arm upper tube S3. m In the example, the magnetizing inductance L m The current starts to decrease, and the resonant inductor L r and resonant capacitor C r The second half cycle of series resonance is continued, and power is output to the secondary circuit through the high-frequency transformer, and the supply voltage is output to the load;

[0042] The mode 6, the resonant inductance L r and resonant capacitor C r The series resonance has completed the second half cycle, the resonant current drops to 0, the output rectifier diodes D2 and D3 are cut off, the A-phase input inductor L1 is excited through the first bridge arm lower tube S2, the B-phase input inductor L2 is freewheeling through the second bridge arm upper tube S3, and the sixth power supply voltage is output to the secondary circuit through the high-frequency transformer;

[0043] In the mode 7, the freewheeling current of the B-phase input inductor L2 drops to 0, and the current of the A-phase input inductor L1 continues to increase linearly. The primary circuit outputs the seventh power supply voltage to the secondary circuit through the high-frequency transformer, and the secondary circuit outputs the seventh power supply voltage through the output capacitor C H Perform power output;

[0044] The mode 8 is the driving dead time of the switch tube. The currents of the A-phase input inductor L1 and the B-phase input inductor L2 charge the parasitic capacitances of the first bridge arm lower tube S2 and the second bridge arm upper tube S3 respectively. The drain-source voltages of the first bridge arm lower tube S2 and the second bridge arm upper tube S3 increase slowly. The currents of the two-phase input inductors also discharge the parasitic capacitances of S1 and S4 through resonance. The drain-source voltages of S1 and S4 decrease slowly. At the same time, the primary circuit outputs the eighth power supply voltage to the secondary circuit. The rectifier diodes D1 and D4 are turned on for high-frequency rectification. Before the end of mode 8, the drain-source voltages of S2 and S3 increase to C L The voltage of S1 and S4 drops to 0 at the same time. L The voltage clamping effect is achieved, and the series resonant circuit begins the series resonance of the first half cycle of the next round.

[0045] In some preferred embodiments, the series resonance includes a positive half cycle and a negative half cycle:

[0046] The positive half cycle, specifically, starts from mode 8 and lasts until the end of mode 1, and the resonant capacitor C r The current i Cr is a sine curve;

[0047] The negative half cycle, specifically, starts from mode 4 and lasts until the end of mode 5, and the resonant capacitor C r The current i Cr is a sine curve.

[0048] In some preferred embodiments, the series resonance, its resonant current i ac for:

[0049]

[0050] Among them, T r is the resonant period, T s is the switching period of the converter, I O is the DC current output by the secondary circuit, ω r is the resonant angular frequency, t is the operating time, and n represents the high-frequency transformer ratio;

[0051] The component v of the output rectified voltage at the resonant frequency O—F for:

[0052]

[0053] Among them, D is the duty cycle of the main switch tube, V H express;

[0054] Output load converted to the equivalent load R of the primary side eq for:

[0055]

[0056] Among them, R L Represents the load resistance.

[0057] In some preferred implementations, the design constraints of the series resonant circuit are:

[0058]

[0059] Where Q is the quality factor of the series resonant cavity;

[0060] The equivalent load R that the converter can apply eq satisfy:

[0061]

[0062] The switching frequency of the converter is fixed, and it works in a mixed mode of resonance and PWM. The power output and soft switching of all switches are achieved through high-frequency resonance, and the output voltage is adjusted by adjusting the PWM duty cycle. Under the premise of keeping the resonance action unaffected by PWM control, the minimum duty cycle of the switch is D min =T r / T s , the maximum duty cycle is D max =1-T r / T s ;

[0063] The gain G range is:

[0064]

[0065] With the resonance period T r As the value of is reduced, the gain range of the converter becomes wider. However, the peak value of the resonant current will also increase, the time for the transformer to transfer energy will be shortened, and the conduction loss will increase.

[0066] Beneficial effects of the present invention:

[0067] (1) The present invention is an isolated DC boost converter based on a two-phase parallel boost circuit, which adopts a high-frequency resonance and PWM hybrid control mode. The power output and soft switching of all switch tubes are achieved through high-frequency resonance, and the output voltage is controlled by adjusting the PWM duty cycle. Therefore, high switching frequency, high power density and high efficiency can be achieved.

[0068] (2) The present invention is based on an isolated DC boost converter with a two-phase parallel boost circuit. The design frequency of the converter series resonance is higher than the switching frequency (generally designed to be more than 2 times), and through reasonable design of the resonant inductor and resonant capacitor, the resonance process is not affected by the switching process. Therefore, the converter can adopt a simple PWM control method and has a very wide voltage input range.

[0069] (3) The present invention is based on an isolated DC boost converter of a two-phase parallel boost circuit, which introduces a series resonant capacitor and performs high-frequency series resonance with the leakage inductance of a high-frequency transformer or introduces an independent inductor, thereby eliminating the voltage spike problem of the switching tube caused by the leakage inductance of the transformer. The series resonant frequency is higher than the switching frequency, and the resonance process is not affected by the PWM switching action of the switching tube.

[0070] (4) The present invention is based on an isolated DC boost converter of a two-phase parallel boost circuit. By combining high-frequency resonance with PWM, the current peak value and effective value of the switch tube are reduced, and zero-current switching of the output diode is achieved, thereby reducing device losses and improving system efficiency.

[0071] (5) The isolated DC boost converter of the present invention is based on a two-phase parallel boost circuit. By reasonably designing the inductance of the input inductor, soft switching of the switch tube can be achieved within a wide voltage range, thereby reducing the switching loss of the circuit and achieving a higher power density by increasing the switching frequency.

[0072] (6) The present invention is based on an isolated DC boost converter of a two-phase parallel boost circuit, which achieves a higher boost ratio through a primary-side Boost topology and a high-frequency transformer. Therefore, the voltage stress of the primary-side switch tube is relatively low, and a switch device with a lower voltage and a smaller on-resistance can be used to improve efficiency. BRIEF DESCRIPTION OF THE DRAWINGS

[0073] Other features, objects and advantages of the present application will become more apparent by reading the detailed description of non-limiting embodiments made with reference to the following drawings:

[0074] Figure 1 It is a circuit structure schematic diagram of an isolated DC boost converter based on a two-phase parallel boost circuit of the present invention;

[0075] Figure 2 It is a waveform diagram of main working parameters of the isolated DC boost converter based on the two-phase parallel boost circuit of the present invention;

[0076] Figure 3 It is a schematic diagram of an equivalent circuit structure of an isolated DC boost converter based on a two-phase parallel boost circuit according to an embodiment of the present invention, in mode 1;

[0077] Figure 4 It is a schematic diagram of an equivalent circuit structure of an isolated DC boost converter based on a two-phase parallel boost circuit according to an embodiment of the present invention, in mode 2;

[0078] Figure 5 It is a schematic diagram of an equivalent circuit structure of an isolated DC boost converter based on a two-phase parallel boost circuit according to an embodiment of the present invention, in mode 3;

[0079] Figure 6 It is a schematic diagram of an equivalent circuit structure of an isolated DC boost converter based on a two-phase parallel boost circuit according to an embodiment of the present invention, in mode 4;

[0080] Figure 7 It is a schematic diagram of an equivalent circuit structure of an isolated DC boost converter based on a two-phase parallel boost circuit according to an embodiment of the present invention, in mode 5;

[0081] Figure 8 It is a schematic diagram of an equivalent circuit structure of an isolated DC boost converter based on a two-phase parallel boost circuit according to an embodiment of the present invention, in a circuit in mode 6;

[0082] Fig. 9 1 is a schematic diagram of an equivalent circuit structure of an isolated DC boost converter based on a two-phase parallel boost circuit according to an embodiment of the present invention, in mode 7;

[0083] Fig.10 It is a schematic diagram of an equivalent circuit structure of an isolated DC boost converter based on a two-phase parallel boost circuit according to an embodiment of the present invention, in mode 8;

[0084] Fig.11 It is a derivation diagram of the resonant cavity output equivalent impedance of an embodiment of an isolated DC boost converter based on a two-phase parallel boost circuit of the present invention. DETAILED DESCRIPTION

[0085] The present application will be further described in detail below in conjunction with the accompanying drawings and embodiments. It is to be understood that the specific embodiments described herein are only used to explain the relevant invention, rather than to limit the invention. It is also necessary to explain that, for ease of description, only the parts related to the relevant invention are shown in the accompanying drawings.

[0086] It should be noted that, in the absence of conflict, the embodiments and features in the embodiments of the present application can be combined with each other. The present application will be described in detail below with reference to the accompanying drawings and in combination with the embodiments.

[0087] The present invention aims at the shortcomings of the existing isolated DC converter topology, such as narrow voltage control range and low efficiency, and proposes a new type of isolated DC boost converter based on a two-phase parallel boost circuit. The voltage spike problem caused by leakage inductance is eliminated by introducing high-frequency series resonance with leakage inductance. By combining high-frequency resonance with PWM, the current peak value and effective value of the switching device are reduced, and the zero current switch of the output diode is realized to reduce loss and improve efficiency. At the same time, the high-frequency resonance technology is combined to realize the soft switching of the switch tube in a wide voltage range, improve the voltage gain while reducing the circuit switching loss, which is conducive to the increase of the switching frequency and the power density of the converter.

[0088] The present invention discloses an isolated DC boost converter based on a two-phase parallel boost circuit, the converter comprising an input power supply, a two-phase input inductor, a DC bus capacitor, a full-bridge unit, a series resonant circuit, a high-frequency transformer, an output rectifier bridge and a filter capacitor;

[0089] The two-phase input inductor and the first bridge arm and the second bridge arm of the full-bridge unit respectively form a two-phase parallel Boost circuit for boosting and stabilizing the input DC voltage;

[0090] The DC bus capacitor is used for filtering the output voltage of the two-phase parallel Boost circuit;

[0091] The series resonant circuit completes two resonant half-waves within one switching cycle of the converter, and realizes the transmission of power from the primary side to the secondary side through the high-frequency transformer;

[0092] The output rectification and filtering circuit is used for high-frequency rectification and filtering of the AC voltage output by the high-frequency transformer to obtain an output DC voltage.

[0093] In order to more clearly explain the isolated DC boost converter based on the two-phase parallel boost circuit of the present invention, Figure 1 Each module in the embodiment of the present invention is described in detail.

[0094] The isolated DC boost converter based on the two-phase parallel boost circuit of the first embodiment of the present invention includes an input power supply 1, a two-phase input inductor 2, a first bridge arm 3 of a full-bridge unit, a second bridge arm 4 of a full-bridge unit, a DC bus capacitor 5, a series resonant circuit 6, a high-frequency transformer 7, an output rectifier bridge 8, an output filter capacitor 9 and a DC load 10. Each module is described in detail as follows:

[0095] The positive electrode of the input power supply is connected to the two-phase input inductor, and the negative electrode is connected to the DC bus capacitor; the input power supply 1 in this embodiment is a wide-range fluctuating DC power supply output by photovoltaic or fuel cell, or other wide-range fluctuating DC power supplies, which are not limited in the present invention.

[0096] The two-phase input inductor and the DC bus capacitor are connected to the full-bridge unit together;

[0097] The full-bridge unit is connected to a series resonant circuit;

[0098] The series resonant circuit is connected in sequence to an output rectifier bridge and a filter capacitor via a high-frequency transformer.

[0099] In this embodiment, the two-phase input inductor includes an A-phase input inductor L1 and a B-phase input inductor L2;

[0100] The full-bridge unit comprises a first bridge arm and a second bridge arm, wherein the first bridge arm comprises a first bridge arm upper tube S1 and a first bridge arm lower tube S2, and the second bridge arm comprises a second bridge arm upper tube S3 and a second bridge arm lower tube S4;

[0101] The series resonant circuit includes a resonant capacitor C r and resonant inductor L r ; The resonant frequency of the series resonant circuit is higher than the switching frequency of the full-bridge unit, and the voltage of the DC boost converter is adjusted by adjusting the duty cycle of the full-bridge unit; In this embodiment, the full-bridge unit is controlled by a pulse width modulation PWM method, and the soft switching of all the switch tubes in the full-bridge unit is achieved by combining high-frequency resonance with PWM, and the current peak value and effective value of the switching device are reduced. In this functional module, unlike the conventional DC converter that outputs power to the secondary side by applying a square wave current to the primary side of the high-frequency transformer, the DC boost converter outputs power by applying a high-frequency resonant sinusoidal current to the primary side of the high-frequency transformer.

[0102] The high-frequency transformer comprises a primary winding, a secondary winding and an excitation inductor, the primary winding comprises a first end and a second end, and the secondary winding comprises a third end and a fourth end;

[0103] The first bridge arm upper tube S1 and the first bridge arm lower tube S2 are connected to form the midpoint of the first bridge arm, and the second bridge arm upper tube S3 and the second bridge arm lower tube S4 are connected to form the midpoint of the second bridge arm;

[0104] The A-phase input inductor is connected in series between the positive electrode of the input power supply and the midpoint of the first bridge arm of the full-bridge unit; the B-phase input inductor is connected in series between the positive electrode of the input power supply and the midpoint of the second bridge arm of the full-bridge unit; the first bridge arm and the second bridge arm are connected in parallel with the DC bus capacitor;

[0105] A resonant capacitor is connected in series between the midpoint of the first bridge arm and the first end of the primary winding of the high-frequency transformer; a resonant inductor is connected in series between the midpoint of the second bridge arm and the second end of the primary winding of the high-frequency transformer;

[0106] The primary excitation inductance of the high-frequency transformer is connected to both ends of the primary winding; the primary excitation inductance L mIt can also replace the resonant inductor L r As a resonant inductor in a series resonant circuit;

[0107] The anode of a diode in the third bridge arm of the output rectifier bridge is connected to the cathode of another diode to form the midpoint of the third bridge arm, and the anode of a diode in the fourth bridge arm of the output rectifier bridge is connected to the cathode of another diode to form the midpoint of the fourth bridge arm; the third end and the fourth end of the secondary winding of the high-frequency transformer are respectively connected to the midpoints of the third bridge arm and the fourth bridge arm of the output rectifier bridge; the third bridge arm and the fourth bridge arm are connected in parallel with the output filter capacitor. The current of each diode in the third bridge arm and the fourth bridge arm of the output rectifier bridge is a half-sine wave.

[0108] In this embodiment, the two-phase input inductor, the first bridge arm and the second bridge arm form a two-phase parallel Boost boost circuit, and the Boost boost circuit is used for boosting and stabilizing the input DC voltage. The current on the inductor in the Boost boost circuit changes linearly, and the current of the two-phase input inductor in the DC boost converter is the superposition of the linear current and the resonant current, which greatly reduces the current ripple, reduces the volume of the input filter, achieves the same current effective value while reducing its peak value, and reduces the current stress of the power device. In the Boost circuit, the lower tube current of the bridge arm of the full-bridge unit first bridge arm 3 and the second bridge arm 4 of the full-bridge unit is the sum of the input inductor current and the primary current of the high-frequency transformer 7, and the upper tube current of the bridge arm of the full-bridge unit first bridge arm 3 and the second bridge arm 4 of the full-bridge unit is the difference between the input inductor current and the primary current of the high-frequency transformer 7.

[0109] A second embodiment of the present invention discloses a control method for an isolated DC boost converter based on a two-phase parallel boost circuit, which is applied to the above-mentioned isolated DC boost converter based on a two-phase parallel boost circuit, and specifically includes:

[0110] Step S100, providing an initial voltage to the entire circuit through an input power supply;

[0111] Step S200, based on the initial voltage, boosting and stabilizing the voltage through two-phase input inductors;

[0112] Step S300, controlling the first bridge arm upper tube S1, the first bridge arm lower tube S2, the second bridge arm upper tube S3, and the second bridge arm lower tube S4 in the full-bridge unit through a set driving signal to adjust the working mode and output a boost voltage to the series resonant circuit;

[0113] At the same time, the boost voltage output by the full-bridge unit is filtered through the DC bus capacitor;

[0114] Step S400, based on the boost voltage, two resonant half-waves are completed in one switching cycle of the converter through a series resonant circuit, the switch tube voltage spike caused by the transformer leakage inductance is eliminated, and the primary voltage is output;

[0115] Step S500, based on the primary voltage, converting it into a secondary voltage through a high-frequency transformer; in this embodiment, the turns ratio of the primary winding to the secondary winding is set to 1:n;

[0116] Step S600: After each resonant half cycle of the primary voltage is completed, the resonant action is blocked by the output rectifier bridge, and the DC supply voltage is output to the load after filtering by the filter capacitor.

[0117] The output DC voltage is used as a supply voltage for the DC load 10. The output DC voltage can also be connected to a DC grid.

[0118] In this embodiment, the working modes include: mode 1, D=0.5; mode 2, D>0.5; mode 3, D<0.5; wherein D represents the duty cycle of the main switch tube, and the main switch tube is the first bridge arm lower tube S2 and the second bridge arm lower tube S4; in mode 1, the converter voltage equivalent gain is 2; in mode 2, the converter voltage equivalent gain is <2; in mode 3, the converter voltage equivalent gain is >2. The upper tube and the lower tube of the first bridge arm 3 of the full-bridge unit and the second bridge arm 4 of the full-bridge unit are complementary turned on, and the phase difference of the driving signal of the first bridge arm and the second bridge arm is 180°, and the output voltage can be controlled by changing the duty cycle of the upper tube and the lower tube of the bridge arm.

[0119] Taking mode 1 as an example, the switching cycle includes mode 1 (t0-t1), mode 2 (t1-t2), mode 3 (t2-t3), mode 4 (t3-t4), mode 5 (t4-t5), mode 6 (t5-t6), mode 7 (t6-t7) and mode 8 (t7-t8); the main operating parameter waveforms of the isolated DC boost converter of the two-phase parallel boost circuit of each mode are as follows: Figure 2 As shown; V gs1 、V gs2 Represent the driving signals of S1 and S2 respectively, i L1 、i L2 Represents the current of the input inductor of phase A and phase B respectively, i Cr Represents the current of the resonant capacitor Cr, i ds1 、i ds2 Represent the drain-source current of S1 and S2 respectively, v ds1 、v ds2 Represent the drain-source voltage of S1 and S2 respectively, i d5 、i d6 Represent the current flowing through diodes D3 and D4 respectively;

[0120] Among them, in modes 1, 2 and 3, the upper tube S1 of the first bridge arm and the lower tube S4 of the second bridge arm are opened, and the lower tube S2 of the first bridge arm and the upper tube S3 of the second bridge arm are closed;

[0121] In mode 1, the circuit state is as follows Figure 3 As shown, before t0, S1, S2, S3, and S4 are all in the off state, and the parasitic capacitance of each switch tube resonates with the two-phase input inductance to achieve soft switching. At t0, S1 and S4 are turned on, S1 provides a freewheeling circuit for L1, the first bridge arm upper tube S1 provides a freewheeling circuit for the A-phase input inductor L1, and the B-phase input inductor L2 is excited through the second bridge arm lower tube S4. The resonant inductor L r and resonant capacitor C r The first half cycle of series resonance is performed, and power is output to the secondary circuit through the high-frequency transformer. The output rectifier diodes D1 and D4 in the output rectifier bridge are turned on for high-frequency rectification, and the supply voltage is output to the load. The rectifier diodes D2 and D3 remain off. The resonant current at this stage is a half-sine wave, so D1 and D4 are turned on and off with zero current, which can reduce the reverse recovery loss of the diode. During the resonance process, the current of S1 is the difference between the current of L1 and the resonant current and the excitation inductor current, and the current of S4 is the sum of the current of L2 and the resonant current and the excitation inductor current. Therefore, since the resonant frequency is higher than the switching frequency, the resonant current has dropped to 0 before the inductor current reaches its peak value, which can reduce the current peak and effective value of the switch tube and reduce the conduction loss of the switch tube.

[0122] In this embodiment, the series resonance includes a positive half cycle and a negative half cycle:

[0123] The positive half cycle, specifically, starts from mode 8 and lasts until the end of mode 1, and the resonant capacitor C r The current i Cr is a sine curve;

[0124] The negative half cycle, specifically, starts from mode 4 and lasts until the end of mode 5, and the resonant capacitor C r The current i Cr is a sine curve;

[0125] In mode 2, the circuit state is as follows Figure 4 As shown, at time t1, the resonant inductor L r and resonant capacitor C r The series resonance has completed the first half cycle, the resonant current is 0, and the resonant inductance L r The resonant voltage reaches its maximum value; the output rectifier diodes D1, D4, D2 and D3 in the output rectifier bridge are cut off; due to L r and C r The quality factor of the resonant cavity is less than 1, C r The resonant voltage vCr Less than C L voltage, so the series resonance process stops at time t1; during this stage, L1 continues to flow through S1, and L2 continues to be excited through S4.

[0126] The circuit state of mode 3 is as follows: Figure 5 As shown, at time t2, the freewheeling current of the A-phase input inductor L1 drops to 0, and the current of the B-phase input inductor L2 continues to increase linearly; the secondary side still passes through the output capacitor C H When power is output, the reverse current of the A-phase input inductor L1 provides the condition for S1 to achieve soft shutdown;

[0127] In mode 4, the first bridge arm upper tube S1, the second bridge arm lower tube S4, the first bridge arm lower tube S2 and the second bridge arm upper tube S3 are turned off;

[0128] The circuit state of mode 4 is as follows: Figure 6 As shown in FIG. 1 , it is the driving dead time of the switch tube. At time t3, after the upper tube S1 of the first bridge arm and the lower tube S4 of the second bridge arm are turned off, due to the existence of parasitic capacitance in each switch tube, the drain-source voltage of the switch tubes S1 and S4 slowly increases, realizing the zero voltage turn-off of S1 and S4. The parasitic output capacitance of S1 and S4 is charged by the current of the A-phase input inductor L1 and the B-phase input inductor L2 respectively. At the same time, the current of the A-phase input inductor L1 and the B-phase input inductor L2 also discharges the parasitic output capacitance of the lower tube S2 of the first bridge arm and the upper tube S3 of the second bridge arm through the effect of resonance. The drain-source voltage of S2 and S3 slowly decreases, and the rectifier diodes D2 and D3 are turned on for high-frequency rectification. Before the end of mode 4, the voltage of S1 and S4 has increased to C L The voltage of S2 and S3 has dropped to 0 at the same time; the supply voltage is output to the load; and the condition for zero voltage switching is provided. L Due to the voltage clamping effect, the resonant cavity begins to resonate in series and outputs energy to the secondary side through the high-frequency transformer. Since the resonant current is a sine wave, D2 and D3 are turned on with zero current.

[0129] In modes 5, 6 and 7, the lower tube S2 of the first bridge arm and the upper tube S3 of the second bridge arm are opened, and the upper tube S1 of the first bridge arm and the lower tube S4 of the second bridge arm are closed;

[0130] The circuit state of mode 5 is as follows: Figure 7 As shown, at time t4, the lower tube S2 of the first bridge arm and the upper tube S3 of the second bridge arm are opened at zero pressure, C L The voltage is applied to the resonant cavity and the high-frequency transformer excitation inductor L through the first bridge arm lower tube S2 and the second bridge arm upper tube S3. m In the example, the magnetizing inductance L m The current starts to decrease, and the resonant inductor Lr and resonant capacitor C r The second half cycle of series resonance is continued, and power is output to the secondary circuit through the high-frequency transformer, and the supply voltage is output to the load;

[0131] The circuit state of mode 6 is as follows: Figure 8 As shown, at time t5, the resonant inductor L r and resonant capacitor C r The series resonance has completed the second half cycle, the resonant current drops to 0, the output rectifier diodes D2 and D3 are cut off, the A-phase input inductor L1 is excited through the first bridge arm lower tube S2, the B-phase input inductor L2 is freewheeling through the second bridge arm upper tube S3, and the sixth power supply voltage is output to the secondary circuit through the high-frequency transformer; the circuit parameter state at this time is similar to mode 2. During the resonance process, the current of S3 is the difference between the L2 current and the resonant current and the excitation inductor current, and the current of S2 is the sum of the L1 current and the resonant current and the excitation inductor current; due to C r The resonant voltage v Cr Less than C L The voltage of , so the series resonance process stops at t5;

[0132] The circuit state of mode 7 is as follows: Fig. 9 As shown, the circuit parameters are similar to those of Mode 3. At time t6, the freewheeling current of the B-phase input inductor L2 drops to 0, and the current of the A-phase input inductor L1 continues to increase linearly. The primary circuit outputs the seventh power supply voltage to the secondary circuit through the high-frequency transformer, and the secondary circuit outputs the seventh power supply voltage through the output capacitor C H Perform power output;

[0133] In mode 8, the first bridge arm upper tube S1, the second bridge arm lower tube S4, the first bridge arm lower tube S2 and the second bridge arm upper tube S3 are turned off. After mode 8 ends, mode 1 is re-entered to start a new switching cycle.

[0134] The circuit state of mode 8 is as follows: Fig.10 As shown in FIG. 1 , it is the driving dead time of the switch tube. At time t7, the current of the A-phase input inductor L1 and the B-phase input inductor L2 respectively charges the parasitic capacitance of the first bridge arm lower tube S2 and the second bridge arm upper tube S3, and the drain-source voltage of the switch tube increases slowly, realizing the zero voltage shutdown of the first bridge arm lower tube S2 and the second bridge arm upper tube S3. The two-phase input inductor current also discharges the parasitic capacitance of S1 and S4 through the effect of resonance, and the drain-source voltage of the switch tube decreases slowly. At the same time, the primary circuit outputs the eighth power supply voltage to the secondary circuit, and the rectifier diodes D1 and D4 are turned on for high-frequency rectification. Before the end of mode 8, the drain-source voltage of S2 and S3 increases to C L The voltage of S1 and S4 drops to 0 at the same time. LThe voltage clamping effect is achieved, and the series resonant circuit begins the series resonance of the first half cycle of the next round.

[0135] After t1 or t5 (mode 2 and mode 6), due to C L The voltage clamping effect is achieved, the series resonance is cut off, and in the series resonance stage the circuit outputs the supply voltage to the load. That is, v Cr <v CL According to this constraint, the resonant cavity design conditions and the maximum load impedance of the converter proposed in the present invention can be derived.

[0136] The simplified model of the resonance stage is as follows Fig.11 As shown, C r is the resonant capacitor, L r is the resonant inductance, v in is the equivalent input voltage of the resonant cavity, v O is the equivalent voltage of the output voltage after passing through the transformer and converted to the primary side, R eq I is the equivalent load converted from the output load to the primary side. O is the output DC current, i ac is the resonant current, v O—F is the component of the output rectified voltage at the resonant frequency, D is the duty cycle of the switches S2 and S4, T s is the switching period of the converter.

[0137] In this embodiment, the series resonance, its resonant current i ac for:

[0138]

[0139] Among them, T r is the resonant period, T s is the switching period of the converter, I O is the DC current output by the secondary circuit, ω r is the resonant angular frequency, t is the operating time, and n represents the high-frequency transformer ratio;

[0140] The component v of the output rectified voltage at the resonant frequency O—F for:

[0141]

[0142] Among them, D is the duty cycle of the main switch tube, V H express;

[0143] Output load converted to the equivalent load R of the primary side eq for:

[0144]

[0145] Among them, R L Represents the load resistance.

[0146] In this embodiment, the design constraints of the series resonant circuit are:

[0147]

[0148] Where Q is the quality factor of the series resonant cavity;

[0149] The equivalent load R that the converter can apply eq satisfy:

[0150]

[0151] Under the premise of keeping the resonant action unaffected by PWM control, the minimum duty cycle of the switch tube is D min =T r / T s , the maximum duty cycle is D max =1-T r / T s ;

[0152] The gain G range is:

[0153]

[0154] With the resonance period T r The gain range of the converter becomes wider as the voltage decreases. However, the peak value of the resonant current will also increase, the time for the transformer to transfer energy will be shortened, and the conduction loss will increase. Therefore, in practical applications, a larger resonant period should be selected as much as possible according to the voltage fluctuation range to design L. r and C r .

[0155] Since the resonant frequency of the series resonant circuit 6 is higher than the switching frequency of the converter (generally designed to be more than 2 times), within a certain duty cycle range, the series resonance process is not affected by the duty cycle, so that the superposition of resonance and PWM control can be achieved, which greatly reduces the conduction loss of the device and increases the service life of the device.

[0156] The terms "first", "second", etc. are used to distinguish similar objects rather than to describe or indicate a particular order or sequence.

[0157] The term "comprise" or any other similar term is intended to cover non-exclusive inclusion, such that a process, method, article, or apparatus / device that includes a list of elements includes not only those elements but also other elements not expressly listed, or also includes elements inherent to such process, method, article, or apparatus / device.

[0158] So far, the technical solutions of the present invention have been described in conjunction with the preferred embodiments shown in the accompanying drawings. However, it is easy for those skilled in the art to understand that the protection scope of the present invention is obviously not limited to these specific embodiments. Without departing from the principle of the present invention, those skilled in the art can make equivalent changes or substitutions to the relevant technical features, and the technical solutions after these changes or substitutions will fall within the protection scope of the present invention.

Claims

1. An isolated DC boost converter with a two-phase parallel boost circuit, characterized in that: The converter includes an input power supply, a two-phase input inductor, a DC bus capacitor, a full-bridge unit, a series resonant circuit, a high-frequency transformer, an output rectifier bridge and a filter capacitor; The positive electrode of the input power supply is connected to the two-phase input inductor, and the negative electrode is connected to the DC bus capacitor; The two-phase input inductor and the DC bus capacitor are connected to the full-bridge unit together; The full-bridge unit is connected to a series resonant circuit; The series resonant circuit is connected in sequence to an output rectifier bridge and a filter capacitor via a high-frequency transformer; The two-phase input inductor includes an A-phase input inductor L1 and a B-phase input inductor L2; The full-bridge unit comprises a first bridge arm and a second bridge arm, wherein the first bridge arm comprises a first bridge arm upper tube S1 and a first bridge arm lower tube S2, and the second bridge arm comprises a second bridge arm upper tube S3 and a second bridge arm lower tube S4; The series resonant circuit includes a resonant capacitor C r and resonant inductor L r ; The resonant frequency of the series resonant circuit is higher than the switching frequency of the full-bridge unit, and the output voltage of the converter is adjusted by adjusting the duty cycle of the full-bridge unit; The high-frequency transformer comprises a primary winding, a secondary winding and an excitation inductor, the primary winding comprises a first end and a second end, and the secondary winding comprises a third end and a fourth end; The first bridge arm upper tube S1 and the first bridge arm lower tube S2 are connected to form the midpoint of the first bridge arm, and the second bridge arm upper tube S3 and the second bridge arm lower tube S4 are connected to form the midpoint of the second bridge arm; The A-phase input inductor is connected in series between the positive electrode of the input power supply and the midpoint of the first bridge arm of the full-bridge unit; the B-phase input inductor is connected in series between the positive electrode of the input power supply and the midpoint of the second bridge arm of the full-bridge unit; the first bridge arm and the second bridge arm are connected in parallel with the DC bus capacitor; A resonant capacitor is connected in series between the midpoint of the first bridge arm and the first end of the primary winding of the high-frequency transformer; a resonant inductor is connected in series between the midpoint of the second bridge arm and the second end of the primary winding of the high-frequency transformer; The primary excitation inductance of the high-frequency transformer is connected to both ends of the primary winding; An anode of a diode of the third bridge arm of the output rectifier bridge is connected to a cathode of another diode to form a midpoint of the third bridge arm, and an anode of a diode of the fourth bridge arm of the output rectifier bridge is connected to a cathode of another diode to form a midpoint of the fourth bridge arm; the third end and the fourth end of the secondary winding of the high-frequency transformer are respectively connected to the midpoints of the third bridge arm and the fourth bridge arm of the output rectifier bridge; the third bridge arm and the fourth bridge arm are connected in parallel with the filter capacitor; The control method of the isolated DC boost converter applied to the two-phase parallel boost circuit specifically includes: Step S100, providing an initial voltage to the entire circuit through an input power supply; Step S200, based on the initial voltage, boosting and stabilizing the voltage through two-phase input inductors; Step S300, controlling the first bridge arm upper tube S1, the first bridge arm lower tube S2, the second bridge arm upper tube S3, and the second bridge arm lower tube S4 in the full-bridge unit through a set driving signal to adjust the working mode and output a boost voltage to the series resonant circuit; At the same time, the boost voltage output by the full-bridge unit is filtered through the DC bus capacitor; Step S400, based on the boost voltage, two resonant half-waves are completed in one switching cycle of the converter through a series resonant circuit, the switch tube voltage spike caused by the transformer leakage inductance is eliminated, and the primary voltage is output; Step S500, converting the primary voltage into a secondary voltage through a high frequency transformer; Step S600, after each resonant half cycle of the primary voltage is completed, the resonant action is blocked by the output rectifier bridge, and a DC supply voltage is output to the load after filtering by a filter capacitor; The series resonance, its resonant current i ac for: Among them, T r is the resonant period, T s is the switching period of the converter, I O is the DC current output by the secondary circuit, ω r is the resonant angular frequency, t is the operating time, and n represents the high-frequency transformer ratio; The component v of the output rectified voltage at the resonant frequency O—F for: Among them, D is the duty cycle of the main switch tube, V H Indicates the load voltage; Output load converted to the equivalent load R of the primary side eq for: Among them, R L represents the load resistance; The design constraints of the series resonant circuit are: Where Q is the quality factor of the series resonant cavity, v Cr is the resonant capacitor C r The resonant voltage on CL is the DC bus capacitance C L Voltage on L r is the resonant inductor; The equivalent load R that the converter can apply eq satisfy: The switching frequency of the converter is fixed, and it works in a mixed mode of resonance and PWM. The power output and soft switching of all switches are achieved through high-frequency resonance, and the output voltage is adjusted by adjusting the PWM duty cycle. Under the premise of keeping the resonance action unaffected by PWM control, the minimum duty cycle of the switch is D min =T r / T s , the maximum duty cycle is D max =1-T r / T s ; The gain G range is: With the resonance period T r As the value of is reduced, the gain range of the converter becomes wider. However, the peak value of the resonant current will also increase, the time for the transformer to transfer energy will be shortened, and the conduction loss will increase.

2. The isolated DC boost converter of the two-phase parallel boost circuit according to claim 1, characterized in that: The two-phase input inductor, the first bridge arm and the second bridge arm form a two-phase parallel Boost circuit. The current on the inductor in the Boost circuit changes linearly. The current of the two-phase input inductor in the DC boost converter is a superposition of the linear current and the resonant current, so as to achieve the same current effective value while reducing its peak value and reduce the current stress of the power device.

3. The isolated DC boost converter of the two-phase parallel boost circuit according to claim 1, characterized in that: The full-bridge unit is controlled by a pulse width modulation (PWM) method, and at the same time, high-frequency series resonance is combined with PWM modulation to achieve soft switching of all switch tubes in the full-bridge unit and reduce the current peak value and effective value of all switch tubes; the DC boost converter realizes power output by applying a high-frequency resonant sinusoidal current to the primary side of the high-frequency transformer.

4. The isolated DC boost converter of the two-phase parallel boost circuit according to claim 3, characterized in that: The working modes include: Mode 1, D=0.5; Mode 2, D>0.5; Mode 3, D<0.5; wherein D represents the duty cycle of the main switch tube, and the main switch tubes are the first bridge arm lower tube S2 and the second bridge arm lower tube S4; in Mode 1, the converter voltage equivalent gain is 2; in Mode 2, the converter voltage equivalent gain is <2; in Mode 3, the converter voltage equivalent gain is >2.

5. The isolated DC boost converter of the two-phase parallel boost circuit according to claim 4, characterized in that: The mode 1, the switching cycle includes mode 1, mode 2, mode 3, mode 4, mode 5, mode 6, mode 7 and mode 8; Among them, in modes 1, 2 and 3, the upper tube S1 of the first bridge arm and the lower tube S4 of the second bridge arm are opened, and the lower tube S2 of the first bridge arm and the upper tube S3 of the second bridge arm are closed; In mode 4, the first bridge arm upper tube S1, the second bridge arm lower tube S4, the first bridge arm lower tube S2 and the second bridge arm upper tube S3 are turned off; In modes 5, 6 and 7, the lower tube S2 of the first bridge arm and the upper tube S3 of the second bridge arm are opened, and the upper tube S1 of the first bridge arm and the lower tube S4 of the second bridge arm are closed; In mode 8, the first bridge arm upper tube S1, the second bridge arm lower tube S4, the first bridge arm lower tube S2 and the second bridge arm upper tube S3 are turned off. After mode 8 ends, mode 1 is re-entered to start a new switching cycle.

6. The isolated DC boost converter of the two-phase parallel boost circuit according to claim 5, characterized in that: In mode 1, the circuit states of each mode are: In mode 1, the parasitic capacitance of each switch tube resonates with the two-phase input inductance. The upper tube S1 of the first bridge arm provides a freewheeling circuit for the A-phase input inductance L1. The B-phase input inductance L2 is excited by the lower tube S4 of the second bridge arm. r and resonant capacitor C r The first half cycle of series resonance is performed, and power is output to the secondary circuit through the high-frequency transformer. The output rectifier diodes D1 and D4 in the output rectifier bridge are turned on for high-frequency rectification and the supply voltage is output to the load. The rectifier diodes D2 and D3 remain turned off. In mode 2, the resonant inductor L r and resonant capacitor C r The series resonance has completed the first half cycle, the resonant current is 0, and the resonant inductance L r The resonant voltage reaches the maximum value; the output rectifier diodes D1 and D4 in the output rectifier bridge are cut off; In the mode 3, the freewheeling current of the A-phase input inductor L1 drops to 0, and the current of the B-phase input inductor L2 continues to increase linearly; Mode 4 is the driving dead time of the switch tube. After the upper tube S1 of the first bridge arm and the lower tube S4 of the second bridge arm are turned off, due to the presence of parasitic output capacitance in each switch tube, the drain-source voltage of S1 and S4 increases slowly, realizing zero voltage turn-off of S1 and S4, and the parasitic capacitance of S1 and S4 is charged by the current of the A-phase input inductor L1 and the B-phase input inductor L2 respectively; at the same time, the current of the A-phase input inductor L1 and the B-phase input inductor L2 also discharges the parasitic capacitance of the lower tube S2 of the first bridge arm and the upper tube S3 of the second bridge arm through the effect of resonance, and the drain-source voltage of S2 and S3 decreases slowly, and the rectifier diodes D2 and D3 are turned on for high-frequency rectification; the voltage of S1 and S4 has increased to C L The voltage of S2 and S3 has dropped to 0 at the same time; the supply voltage is output to the load; The modality 5, C L The voltage is applied to the resonant cavity and the high-frequency transformer excitation inductor L through the first bridge arm lower tube S2 and the second bridge arm upper tube S3. m In the example, the magnetizing inductance L m The current starts to decrease, and the resonant inductor L r and resonant capacitor C r The second half cycle of series resonance is continued, and power is output to the secondary circuit through the high-frequency transformer, and the supply voltage is output to the load; The mode 6, the resonant inductance L r and resonant capacitor C r The series resonance has completed the second half cycle, the resonant current drops to 0, the output rectifier diodes D2 and D3 are cut off, the A-phase input inductor L1 is excited through the first bridge arm lower tube S2, the B-phase input inductor L2 is freewheeling through the second bridge arm upper tube S3, and the sixth power supply voltage is output to the secondary circuit through the high-frequency transformer; In the mode 7, the freewheeling current of the B-phase input inductor L2 drops to 0, and the current of the A-phase input inductor L1 continues to increase linearly. The primary circuit outputs the seventh power supply voltage to the secondary circuit through the high-frequency transformer, and the secondary circuit outputs the seventh power supply voltage through the output capacitor C H Perform power output; The mode 8 is the driving dead time of the switch tube. The currents of the A-phase input inductor L1 and the B-phase input inductor L2 charge the parasitic capacitances of the first bridge arm lower tube S2 and the second bridge arm upper tube S3 respectively. The drain-source voltages of the first bridge arm lower tube S2 and the second bridge arm upper tube S3 increase slowly. The currents of the two-phase input inductors also discharge the parasitic capacitances of S1 and S4 through resonance. The drain-source voltages of S1 and S4 decrease slowly. At the same time, the primary circuit outputs the eighth power supply voltage to the secondary circuit. The rectifier diodes D1 and D4 are turned on for high-frequency rectification. Before the end of mode 8, the drain-source voltages of S2 and S3 increase to C L The voltage of S1 and S4 drops to 0 at the same time. L The voltage clamping effect is achieved, and the series resonant circuit begins the series resonance of the first half cycle of the next round.

7. The isolated DC boost converter of the two-phase parallel boost circuit according to claim 5, characterized in that: The series resonance includes positive half cycle and negative half cycle: The positive half cycle, specifically, starts from mode 8 and lasts until the end of mode 1, and the resonant capacitor C r The current i Cr is a sine curve; The negative half cycle, specifically, starts from mode 4 and lasts until the end of mode 5, and the resonant capacitor C r The current i Cr is a sine curve.

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