LPST air-gap magnetic field modeling method and inductance parameter acquisition method

Abstract

The invention belongs to the field of transformers, and particularly relates to an LPST (linear phase shift transformer) air-gap magnetic field modeling method and an inductance parameter acquisition method thereof, and the method comprises the steps: analyzing the relation between the primary and secondary sides of the LPST and an air gap, and building an air-gap magnetic field analysis model considering the saturation of a ferromagnetic material; analyzing the relationship between the LPST cogging and the air gap, and building an air gap magnetic field correction model considering the cogging effect; and analyzing the relationship between the LPST side end and the air gap, and building an air gap magnetic field correction model considering the on-off of the iron core. The established LPST air-gap magnetic field model can consider the influence of ferromagnetic material saturation, cogging effect and iron core breaking at the same time, and the calculation precision of each inductance parameter is high, the calculation dimension is small, and the calculation time is short based on the LPST air-gap magnetic field model.

Description

Technical Field

[0001] The present invention belongs to the field of transformers, and in particular relates to an air gap magnetic field modeling method of an LPST (i.e., linear phase-shifting transformer) and a method for obtaining inductance parameters thereof. Background Art

[0002] like Figure 1 As shown in the figure, the linear phase-shifting transformer (LPST) is a new type of phase-shifting transformer designed based on the structure and working principle of the linear motor, and is an important component of the multiple superposition inverter system. The structure of the LPST is basically the same as that of the linear motor, except that the primary and secondary cores of the LPST have the same length and are symmetrical about the air gap. Unlike conventional phase-shifting transformers, the energy conversion of the LPST is mainly achieved through the air gap traveling wave magnetic field. When the primary winding of the LPST is energized, a traveling wave magnetic field is generated inside the core, inducing three-phase alternating current on the secondary side.

[0003] Compared with phase-shifting transformers of other structures, LPST has the advantages of wide phase-shifting angle range, no need for special winding turns ratio, high utilization of core and winding, and ability to achieve electrical isolation. It can be used in rectifier / inverter equipment to effectively eliminate low-order harmonics, improve output waveform quality, and reduce grid harmonic pollution. It can be used in microgrids such as ship integrated power systems and electric vehicles.

[0004] The air gap magnetic field is the medium for LPST energy conversion, and the accuracy of its calculation results directly affects the calculation of electromagnetic characteristics such as transformer impedance, efficiency, loss, and vibration noise. However, the interruption of the transformer core will lead to asymmetric three-phase impedance, and generate end-face flux at the interruption point, causing air gap magnetic field distortion. At the same time, there are many slots on the primary and secondary sides of the transformer, and the slot effect causes the increase of harmonic components in the air gap magnetic field, making the accurate calculation of the air gap magnetic flux density complicated. Due to the special working conditions of ships, the weight and volume of the transformer must be as small as possible, and the air gap magnetic flux density is selected to be relatively high. The influence of the saturation of ferromagnetic materials on the transformer air gap magnetic field cannot be ignored.

[0005] The existing LPST air gap magnetic field model still has the following shortcomings: First, it is unable to simultaneously consider the influence of multiple factors such as ferromagnetic material saturation, cogging effect and core breaking; Second, the model is relatively complex, the calculation process is cumbersome, and the amount of calculation is large. Summary of the invention

[0006] The purpose of the present invention is to solve the technical problems existing in the background technology. To this end, a LPST air gap magnetic field modeling method and an inductance parameter acquisition method thereof are provided.

[0007] In order to achieve the above object, the technical solution adopted by the present invention is as follows:

[0008] A method for modeling a LPST air gap magnetic field comprises the following steps:

[0009] Step S1: Analyze the relationship between the primary and secondary sides of the LPST and the air gap, and build an analytical model of the air gap magnetic field considering the saturation of ferromagnetic materials;

[0010] Step S101: Divide the air gap area, primary side tooth area, primary side yoke area, secondary side tooth area, and secondary side yoke area of ​​the LPST along the length direction into blocks at equal intervals. Within the length range of the LPST, the air gap center line is equally spaced into N blocks, with a total of N+1 nodes;

[0011] Obtain the total magnetic pressure drop of the loop, the normal magnetic pressure drop of the air gap at the mth and m+1th nodes, the longitudinal magnetic pressure drop of the yoke at the mth node of the primary core, the normal magnetic pressure drop of the virtual tooth at the mth and m+1th nodes of the primary core, the longitudinal magnetic pressure drop of the yoke at the mth node of the secondary core, and the normal magnetic pressure drop of the virtual tooth at the mth and m+1th nodes of the secondary core;

[0012] Step S102: fitting the BH curve of the nonlinear ferromagnetic material to correct the actual normal flux density of the tooth when the ferromagnetic material is saturated;

[0013] Step S103: According to the flux continuity theorem, the yoke transverse magnetic flux at the mth node is equal to the total longitudinal magnetic flux of the air gap from the first node to the mth node, and the yoke magnetic flux is obtained;

[0014] Step S104: combining the distribution characteristics of the LPST winding, simplifying the magnetomotive force of each node of the air gap, and obtaining the air gap flux density at the mth node;

[0015] Step S105: Based on the simplified magnetomotive force of each node of the air gap, iterative judgment is performed until the judgment condition is met. If the condition is met, the air gap magnetic flux density is output; if the condition is not met, the air gap magnetic flux density is corrected;

[0016] Step S2: Analyze the relationship between the LPST tooth slot and the air gap, and build an air gap magnetic field correction model considering the tooth slot effect;

[0017] Step S201: determining the relative magnetic permeability of the air gap under the influence of the cogging effect;

[0018] Step S3: Analyze the relationship between the LPST edge and the air gap, and build an air gap magnetic field correction model considering the core breaking;

[0019] Step S301: simplify the edge model and regularize the magnetic field distribution through Xuck transformation;

[0020] Step S302: taking into account the end magnetic flux density of the core being broken;

[0021] Step S303: using the flux continuity theorem, introducing the core breaking flux density change value to represent the lost flux density;

[0022] Step S4: The final air gap magnetic field is the product of the slotless air gap magnetic field under the condition of ferromagnetic material saturation and the air gap relative permeability function under the influence of tooth slots and core breaking factors.

[0023] The following is a technical solution further defined by the modeling method in the present invention: the primary and secondary sides of the LPST are symmetrical about the center line of the air gap.

[0024] The following is a technical solution further defined by the modeling method in the present invention, and the total magnetic pressure drop of the loop is:

[0025] F ∑ (m) = F δ_n (m+1)+F t1_n (m+1)-F δ_n (m)-F j1_l (m)-F t1_n (m)

[0026] In the formula, F δ_n (m), F δ_n (m+1) are the normal magnetic pressure drops of the air gap at the mth and m+1th nodes, respectively, F j1_l (m) is the longitudinal magnetic voltage drop of the yoke at the mth node of the primary core, F t1_n (m), F t1_n (m+1) is the normal magnetic voltage drop of the virtual tooth at the mth and m+1th nodes of the primary core. The positive direction of the coordinate axis is the positive direction of the magnetic voltage drop of each section. According to Ampere's loop theorem:

[0027]

[0028] In the formula, H δ_n is the magnetic field strength along the normal direction of the air gap; B t1_n , H t1_n is the normal magnetic flux density and magnetic flux density of the virtual tooth of the primary core; B j1_l , H j1_l is the longitudinal magnetic flux density and magnetic flux density of the primary core yoke; δ is the air gap length; h t1 is the virtual tooth height of the primary core; μ0 is the air magnetic permeability; μ j1 is the magnetic permeability of each node of the primary core yoke.

[0029] The following is a technical solution further defined by the modeling method in the present invention, and the actual normal magnetic flux density of the tooth is:

[0030] B t_n (m) = B t_n′(m)-μ0H t_n (m)k δ

[0031] In the formula, B′ t_n (m) is the apparent normal flux density of the tooth at the mth node, which means the flux density when all the magnetic flux enters the tooth; B t_n (m) is the actual normal flux density of the tooth at the mth node; H t_n (m) is the actual normal magnetic field intensity of the tooth at the mth node; k δ is the slot coefficient, k δ =(H·b s ) / (K Fe ·L·b t ).

[0032] The following is a technical solution further defined by the modeling method in the present invention, and the yoke magnetic flux density is:

[0033]

[0034] The following is a technical solution further defined by the modeling method in the present invention, and the simplified magnetomotive force of each node of the air gap is:

[0035]

[0036] In the formula,

[0037]

[0038] The air gap flux density at the mth node is:

[0039]

[0040] In the formula, K B It is a saturation coefficient preset according to the saturation degree of the primary side core.

[0041] The following is a technical solution further defined by the modeling method in the present invention, and the judgment conditions are:

[0042]

[0043] If the judgment condition is not met, the air gap magnetic flux density is corrected:

[0044]

[0045] In the formula, k s is the iteration coefficient.

[0046] The following is a technical solution further defined by the modeling method in the present invention, and the relative magnetic permeability of the air gap is:

[0047]

[0048] In the formula, t1 is the tooth pitch, T is the extension period, then:

[0049]

[0050] The nonlinear function β(y) is:

[0051]

[0052] Wherein, the expression of v is obtained by software fitting.

[0053] The following is a technical solution further defined by the modeling method in the present invention, taking into account the end magnetic flux density of the core being broken:

[0054]

[0055] Using the flux continuity theorem, the core breaking flux change value Q is introduced to represent the lost flux density:

[0056]

[0057] The final air gap magnetic field is the product of the slotless air gap magnetic field under the condition of ferromagnetic material saturation and the air gap relative permeability function under the influence of tooth slot and core breaking factors:

[0058] B _x (x) = (1-Q)B δ_sl_n (x)·λ YB_c_n ·λ FB_c_n

[0059] In the formula, B δ0_sl_n (x) is the air gap flux density expression considering the saturation of ferromagnetic materials; λ YB_c_n , FB_c_n They are the relative permeability functions of the air gap when slots are opened on the primary and secondary sides respectively.

[0060] A method for obtaining inductance parameters, obtaining inductance parameters through a model built by the above-mentioned LPST air gap magnetic field modeling method, comprising:

[0061] In the LPST effective region, the energy storage of the air gap magnetic field is:

[0062]

[0063] Solve the self-inductance and mutual inductance parameters of each LPST winding, including:

[0064]

[0065]

[0066] When solving the self-inductance of the i-th winding, a small current perturbation ΔI is made to the i-th winding. i , the other winding currents remain unchanged, and the magnetic field energy W(I i +ΔI i ) and W(I i -ΔI i ), solve for self-inductance When solving the mutual inductance of the i-th winding and the j-th winding, a small current perturbation ΔI is made to the i-th and j-th windings respectively. i , ΔI j , we get the magnetic field energy W(I i +ΔI i ,I j +ΔI j )、W(I i -ΔI i ,I j +ΔI j )、W(I i -ΔI i ,I j +ΔI j ) and W(I i -ΔI i ,I j -ΔI j ), solve for self-inductance

[0067] Compared with the prior art, the present invention has the following technical effects:

[0068] The LPST air gap magnetic field model constructed by the present invention can simultaneously consider the influence of ferromagnetic material saturation, cogging effect, and core breaking. Based on the LPST air gap magnetic field model of the present invention, the calculation accuracy of each inductance parameter is high, the calculation dimension is small, and the calculation time is short.

[0069] The present invention is further described below in conjunction with the accompanying drawings and embodiments. BRIEF DESCRIPTION OF THE DRAWINGS

[0070] In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the drawings required for use in the embodiments or the prior art will be briefly introduced below. Obviously, the drawings described below are only some embodiments of the present invention. For ordinary technicians in this field, other drawings can be obtained based on these drawings without paying creative work.

[0071] Figure 1 It is a schematic diagram of the structure of a linear phase-shifting transformer;

[0072] Figure 2 It is a simplified diagram of the model of the present invention;

[0073] Figure 3 is an iterative flow chart of the present invention;

[0074] Figure 4 It is the end Hook transformation analysis diagram of the present invention;

[0075] Figure 5 is a comparison diagram of the air gap magnetic field model of the present invention and the finite element simulation result when the DC bus voltage of LPST is 100V;

[0076] Figure 6 is a comparison diagram of the calculated values ​​of each winding of the LPST of the present invention and the finite element simulation values ​​when the DC bus voltage of the LPST is 100V;

[0077] Figure 7 It is a schematic diagram of the LPST winding structure. DETAILED DESCRIPTION

[0078] In order to make the above-mentioned objects, features and advantages of the present invention more obvious and easy to understand, the specific embodiments of the present invention are described in detail below in conjunction with the accompanying drawings. In the following description, many specific details are set forth to facilitate a full understanding of the present invention. However, the present invention can be implemented in many other ways different from those described herein, and those skilled in the art can make similar improvements without violating the connotation of the present invention, so the present invention is not limited by the specific embodiments disclosed below.

[0079] like Figure 2-7 As shown, this embodiment provides a LPST air gap magnetic field modeling method and an inductance parameter acquisition method thereof.

[0080] A LPST air gap magnetic field modeling method is used to build an LPST air gap magnetic field model based on electromagnetic field theory. The LPST air gap magnetic field modeling method includes:

[0081] Step 1: Analyze the relationship between the primary and secondary sides of the LPST and the air gap, and build an analytical model of the air gap magnetic field considering the saturation of ferromagnetic materials.

[0082] Step 2: Analyze the relationship between the LPST tooth slot and the air gap, and build an air gap magnetic field correction model considering the tooth slot effect.

[0083] Step 3: Analyze the relationship between the LPST edge and the air gap, and build an air gap magnetic field correction model considering the core breaking.

[0084] To further elaborate on step one, the specific process is as follows:

[0085] (1) The five regions along the length direction of the LPST (air gap (I), primary side teeth (II), primary side yoke (III), secondary side teeth (IV), secondary side yoke (V)) are divided into blocks at equal intervals. Within the length range of the LPST, the air gap centerline is divided into N blocks at equal intervals, with a total of N+1 nodes. Since the primary and secondary sides of the LPST are symmetrical about the air gap centerline, only one side of the core is analyzed, and the model is simplified as follows Figure 1 shown.

[0086] Figure 1 In the figure, 1 indicates an air gap, 2 indicates a primary side tooth portion, and 3 indicates a primary side yoke portion.

[0087] At this time, the total magnetic pressure drop of the circuit is:

[0088] F ∑ (m) = F δ_n (m+1)+F t1_n (m+1)-F δ_n (m)-F j1_l (m)-F t1_n (m) (1)

[0089] In the formula, F δ_n (m), F δ_n (m+1) are the normal magnetic pressure drops of the air gap at the mth and m+1th nodes, respectively, F j1_l (m) is the longitudinal magnetic voltage drop of the yoke at the mth node of the primary core, F t1_n (m), F t1_n (m+1) is the normal magnetic voltage drop of the virtual tooth at the mth and m+1th nodes of the primary core, and the positive direction of the coordinate axis is the positive direction of the magnetic voltage drop of each section.

[0090] According to Ampere's circuit theorem:

[0091]

[0092] In the formula, H δ_n is the magnetic field strength along the normal direction of the air gap; B t1_n , H t1_n is the normal magnetic flux density and magnetic flux density of the virtual tooth of the primary core; B j1_l , H j1_l is the longitudinal magnetic flux density and magnetic flux density of the primary core yoke; δ is the air gap length; h t1 is the virtual tooth height of the primary core; μ0 is the air magnetic permeability; μ j1 is the magnetic permeability of each node of the primary core yoke.

[0093] (2) The BH curve of the nonlinear ferromagnetic material (DW465-50) is fitted to correct the actual magnetic flux expression of the tooth when the ferromagnetic saturation occurs, as shown in formula (3):

[0094] B t_n (m) = B t_n ′(m)-μ0H t_n (m)k δ (3)

[0095] In the formula, B′ t_n (m) is the apparent normal flux density of the tooth at the mth node, which means the flux density when all the magnetic flux enters the tooth; B t_n (m) is the actual normal flux density of the tooth at the mth node; H t_n (m) is the actual normal magnetic field intensity of the tooth at the mth node; k δ is the slot coefficient, k δ =(H·b s ) / (K Fe ·L·b t ).

[0096] (3) According to the flux continuity theorem, the transverse magnetic flux of the yoke at the mth node is equal to the total longitudinal magnetic flux of the air gap from the first node to the mth node. The magnetic flux of the yoke is obtained as:

[0097]

[0098] (4) Combined with the distribution characteristics of the LPST winding, the magnetomotive force of each node in the air gap is simplified to formula (5):

[0099]

[0100] In the formula,

[0101]

[0102] The air gap flux density at the mth node is:

[0103]

[0104] In the formula, K B It is a saturation coefficient preset according to the saturation degree of the primary side core.

[0105] (5) According to the above solution process, the judgment iteration is performed until the judgment condition is met. The specific judgment condition is as shown in formula (7):

[0106]

[0107] If the judgment condition is not met, the air gap magnetic flux density is corrected:

[0108]

[0109] In the formula, k s is the iteration coefficient.

[0110] Perform multiple iterations until the judgment condition is met. At this time, the air gap magnetic flux distribution within the LPST length range can be obtained. The specific iteration flow chart is as follows: Figure 3 The specific iteration process includes:

[0111] Set the DC bus voltage;

[0112] Calculate the effective value of primary current, air gap magnetomotive force, and initial air gap flux density;

[0113] Selection and calculation: Calculate the magnetic flux of the primary yoke and the primary tooth; correct the magnetic flux of the tooth; calculate the total magnetic voltage drop of the circuit where each node is located;

[0114] Determine whether the conditions are met;

[0115] If the conditions are met, the air gap magnetic flux density is output;

[0116] If the conditions are not met, reselect the magnetic flux density of each node in the air gap and then make the selection and calculation again.

[0117] To further elaborate on step 2, the specific process is as follows:

[0118] Determine the relative permeability of the air gap (per unit value) considering the influence of the cogging effect:

[0119]

[0120] In the formula, t1 is the tooth pitch, T is the extension period (extended with the tooth pitch as the period), so:

[0121]

[0122] The nonlinear function β(y) is:

[0123]

[0124] Wherein, the expression of v is obtained by software fitting.

[0125] To further elaborate on step three, the specific process is as follows:

[0126] Simplify the edge model and regularize the magnetic field distribution through Xu Ke transformation, such as Figure 4 shown.

[0127] The end magnetic flux density considering the core breaking is:

[0128]

[0129] Using the flux continuity theorem, the core breaking flux change value Q is introduced to represent the lost flux density:

[0130]

[0131] The final air gap magnetic field is the product of the slotless air gap magnetic field under the condition of ferromagnetic material saturation and the air gap relative permeability function under the influence of tooth slot and core breaking factors:

[0132] B _x (x) = (1-Q)B δ_sl_n (x)·λ YB_c_n ·λ FB_c_n (14)

[0133] In the formula, B δ0_sl_n (x) is the air gap flux density expression considering the saturation of ferromagnetic materials. YB_c_n , FB_c_n They are the relative permeability functions of the air gap when slots are opened on the primary and secondary sides respectively.

[0134] Based on the above model construction method, an LPST air gap magnetic field model based on electromagnetic field theory is constructed to obtain the main inductance parameters of the LPST based on the air gap magnetic field distribution.

[0135] (1) Modeling of the inductance of each LPST winding using the above analysis and based on the energy perturbation method;

[0136] (2) According to the LPST air gap magnetic field distribution, the inductance parameters of each winding are obtained;

[0137] Furthermore, in the LPST effective region, the energy storage of the air gap magnetic field is given by equation (15), where the magnetic flux density B _x (m) From the above analysis, we can know that:

[0138]

[0139] Furthermore, the self-inductance and mutual inductance parameters of each LPST winding are solved, including:

[0140]

[0141] When solving the self-inductance of the i-th winding, a small current perturbation ΔI is made to the i-th winding. i , the other winding currents remain unchanged, and the magnetic field energy W(I i +ΔI i ) and W(I i -ΔI i ), and then solve the self-inductance according to formula (16) When solving the mutual inductance of the i-th winding and the j-th winding, a small current perturbation ΔI is made to the i-th and j-th windings respectively. i , ΔI j , we get the magnetic field energy W(I i +ΔIi ,I j +ΔI j )、W(I i -ΔI i ,I j +ΔI j )、W(I i -ΔI i ,I j +ΔI j ) and W(I i -ΔI i ,I j -ΔI j ), and then solve the self-inductance according to formula (17)

[0142] The LPST air gap magnetic field model calculation program was written based on MATLAB software, and the LPST simulation model was built based on finite element software. The accuracy of the LPST air gap magnetic field model in the present invention can be verified by the operating characteristics of the transformer under different working conditions of the finite element model. Figure 5 , Figure 6 .

[0143] Compared with the finite element model, the LPST air gap magnetic field distribution error calculated by the present invention is less than 2%, the self-inductance parameter error of the 12 phases on the primary side is 0.812%, and the mutual inductance error (taking the a1 phase as an example) is 2.43%; the self-inductance parameter error of the 3 phases on the secondary side is 4.002%, and the mutual inductance error is 0.352%; the mutual inductance error between the primary and secondary sides (taking the a1 phase as an example) is 1.499%.

[0144] Figure 5 The figure shows the comparison between the air gap magnetic field model proposed in the present invention and the finite element simulation results when the DC bus voltage of LPST is 100V.

[0145] Figure 6 Comparison between the calculated values ​​of each LPST winding and the finite element simulation values ​​when the DC bus voltage is 100V.

[0146] In summary, if Figure 7 As shown in the figure, a LPST air gap magnetic field model can be obtained that can simultaneously take into account the saturation of ferromagnetic materials, the slot effect and the influence of core breaking. Under the premise of ensuring the calculation accuracy, the calculation dimension is reduced and the calculation time is shortened. The magnetic field modeling method can meet the preliminary design requirements of LPST and is universal. The inductance parameters obtained through precise magnetic field analysis are more accurate.

[0147] The above is only a preferred embodiment of the present invention, and does not limit the present invention in any form. Any technician familiar with the art can make many possible changes and modifications to the technical solution of the present invention by using the above disclosed methods and technical contents without departing from the scope of the technical solution of the present invention, or modify it into an equivalent embodiment of equivalent changes. Therefore, all equivalent changes made according to the shape, structure and principle of the present invention without departing from the content of the technical solution of the present invention should be included in the protection scope of the present invention.

Claims

1. A LPST air gap magnetic field modeling method, characterized in that: The following steps are involved: Step S1: Analyze the relationship between the primary and secondary sides of the LPST and the air gap, and build an analytical model of the air gap magnetic field considering the saturation of ferromagnetic materials; Step S101: Divide the air gap area, primary side tooth area, primary side yoke area, secondary side tooth area, and secondary side yoke area of ​​the LPST along the length direction into blocks at equal intervals. Within the length range of the LPST, the air gap center line is equally spaced into N blocks, with a total of N+1 nodes; Obtain the total magnetic pressure drop of the loop, the normal magnetic pressure drop of the air gap at the mth and m+1th nodes, the longitudinal magnetic pressure drop of the yoke at the mth node of the primary core, the normal magnetic pressure drop of the virtual tooth at the mth and m+1th nodes of the primary core, the longitudinal magnetic pressure drop of the yoke at the mth node of the secondary core, and the normal magnetic pressure drop of the virtual tooth at the mth and m+1th nodes of the secondary core; Step S102: fitting the BH curve of the nonlinear ferromagnetic material to correct the actual normal flux density of the tooth when the ferromagnetic material is saturated; Step S103: According to the flux continuity theorem, the yoke transverse magnetic flux at the mth node is equal to the total longitudinal magnetic flux of the air gap from the first node to the mth node, and the yoke magnetic flux is obtained; Step S104: combining the distribution characteristics of the LPST winding, simplifying the magnetomotive force of each node of the air gap, and obtaining the air gap flux density at the mth node; Step S105: Based on the simplified magnetomotive force of each node of the air gap, iterative judgment is performed until the judgment condition is met. If the condition is met, the air gap magnetic flux density is output; if the condition is not met, the air gap magnetic flux density is corrected; Step S2: Analyze the relationship between the LPST tooth slot and the air gap, and build an air gap magnetic field correction model considering the tooth slot effect; Step S201: determining the relative magnetic permeability of the air gap under the influence of the cogging effect; Step S3: Analyze the relationship between the LPST edge and the air gap, and build an air gap magnetic field correction model considering the core breaking; Step S301: simplify the edge model and regularize the magnetic field distribution through Xuck transformation; Step S302: taking into account the end magnetic flux density of the core being broken; Step S303: using the flux continuity theorem, introducing the core breaking flux density change value to represent the lost flux density; Step S4: The final air gap magnetic field is the product of the slotless air gap magnetic field under the condition of ferromagnetic material saturation and the air gap relative permeability function under the influence of tooth slots and core breaking factors.

2. A LPST air gap magnetic field modeling method as claimed in claim 1, characterized in that: The primary and secondary sides of the LPST are symmetrical about the center line of the air gap.

3. A LPST air gap magnetic field modeling method as claimed in claim 2, characterized in that: The total magnetic pressure drop of the circuit is: F ∑ (m)=F δ_n (m+1)+F t1_n (m+1)-F δ_n (m)-F j1_l (m)-F t1_n (m) In the formula, F δ_n (m), F δ_n (m+1) are the normal magnetic pressure drops of the air gap at the mth and m+1th nodes, respectively, F j1_l (m) is the longitudinal magnetic voltage drop of the yoke at the mth node of the primary core, F t1_n (m), F t1_n (m+1) is the normal magnetic voltage drop of the virtual tooth at the mth and m+1th nodes of the primary core. The positive direction of the coordinate axis is the positive direction of the magnetic voltage drop of each section. According to Ampere's loop theorem: In the formula, H δ_n is the magnetic field strength along the normal direction of the air gap; B t1_n , H t1_n is the normal magnetic flux density and magnetic flux density of the virtual tooth of the primary core; B j1_l , H j1_l is the longitudinal magnetic flux density and magnetic flux density of the primary core yoke; δ is the air gap length; h t1 is the virtual tooth height of the primary core; μ0 is the air magnetic permeability; μ j1 is the magnetic permeability of each node of the primary core yoke.

4. A LPST air gap magnetic field modeling method as claimed in claim 3, characterized in that: The actual normal magnetic flux density of the tooth is: B t_n (m)=B t_n ′(m)-μ0H t_n (m)k δ In the formula, B′ t_n (m) is the apparent normal flux density of the tooth at the mth node, which means the flux density when all the magnetic flux enters the tooth; B t_n (m) is the actual normal flux density of the tooth at the mth node; H t_n (m) is the actual normal magnetic field intensity of the tooth at the mth node; k δ is the slot coefficient, k δ =(H·b s ) / (K Fe ·L·b t ).

5. A LPST air gap magnetic field modeling method as claimed in claim 4, characterized in that: The magnetic flux density of the yoke is:

6. A LPST air gap magnetic field modeling method as claimed in claim 5, characterized in that: The simplified magnetomotive force of each node in the air gap is: In the formula, The air gap flux density at the mth node is: In the formula, K B It is a saturation coefficient preset according to the saturation degree of the primary side core.

7. A LPST air gap magnetic field modeling method as claimed in claim 6, characterized in that: The judgment conditions are: If the judgment condition is not met, the air gap magnetic flux density is corrected: In the formula, k s is the iteration coefficient.

8. A LPST air gap magnetic field modeling method as claimed in claim 1, characterized in that: The relative magnetic permeability of the air gap is: In the formula, t1 is the tooth pitch, T is the extension period, then: The nonlinear function β(y) is: Wherein, the expression of v is obtained by software fitting.

9. A LPST air gap magnetic field modeling method as claimed in claim 1, characterized in that: The end magnetic flux density considering the core breaking is: Using the flux continuity theorem, the core breaking flux change value Q is introduced to represent the lost flux density: The final air gap magnetic field is the product of the slotless air gap magnetic field under the condition of ferromagnetic material saturation and the air gap relative permeability function under the influence of tooth slot and core breaking factors: B _x (x)=(1-Q)B δ_sl_n (x)·λ YB_c_n ·l FB_c_n In the formula, B δ0_sl_n (x) is the air gap flux density expression considering the saturation of ferromagnetic materials; λ YB_c_n , FB_c_n They are the relative permeability functions of the air gap when slots are opened on the primary and secondary sides respectively.

10. A method for obtaining inductance parameters, wherein the inductance parameters are obtained by using a model constructed by the LPST air gap magnetic field modeling method according to any one of claims 1 to 9, characterized in that: include: In the LPST effective region, the energy storage of the air gap magnetic field is: Solve the self-inductance and mutual inductance parameters of each LPST winding, including: When solving the self-inductance of the i-th winding, a small current perturbation ΔI is made to the i-th winding. i , the other winding currents remain unchanged, and the magnetic field energy W(I i +ΔI i ) and W(I i -ΔI i ), solve for self-inductance When solving the mutual inductance of the i-th winding and the j-th winding, a small current perturbation ΔI is made to the i-th and j-th windings respectively. i , ΔI j , we get the magnetic field energy W(I i +ΔI i ,I j +ΔI j )、W(I i -ΔI i ,I j +ΔI j )、W(I i -ΔI i ,I j +ΔI j ) and W(I i -ΔI i ,I j -ΔI j ), solve for self-inductance

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