TCM control-based inverter output current THD optimization method

Abstract

The invention discloses an inverter output current THD optimization method based on TCM control, and belongs to the technical field of power electronic converter control. The inverter based on TCM control realizes soft switching of a switching device by controlling bidirectional flow of inductive current, and during a dead zone, resonance of parasitic output capacitance and power inductance of the switching device causes large deviation between an actual boundary value of the inductive current and a boundary value set in a digital controller. Considering the influence of the parasitic output capacitance of the switching device and the nonlinearity of the capacitance value of the switching device on the control of the output current of the inverter, the invention provides an inductive current boundary value adaptive correction method on the basis of TCM control, so that the control error caused by the parasitic output capacitance of the switching device is corrected in real time, the control precision of the inductive current is improved, and the control precision of the inverter is improved. And the THD of the output current of the inverter is optimized.

Description

Technical Field

[0001] The present invention relates to the technical field of power electronic converter control, and more specifically, to a method for optimizing THD of inverter output current based on TCM control. Background Art

[0002] As the core device of electric energy conversion, the inverter is widely used in the fields of renewable energy generation, energy storage and motor control. In recent years, the inverter has developed towards high efficiency, high power density and low cost, and soft switching technology is one of the core technologies to achieve the above goals. Traditional inverter control generally adopts a sinusoidal pulse width modulation strategy, and the driving signal of the power switch tube is obtained by comparing the sinusoidal modulation wave with the triangular carrier. The control is simple and easy to implement, but it is difficult to achieve soft switching without auxiliary devices.

[0003] Triangular Current Mode (TCM) is a common control method for power converters to achieve soft switching. It controls the bidirectional flow of inductor current, utilizes the resonance between the power inductor and the parasitic output capacitance of the switching device, and discharges the parasitic output capacitance of the switching device, thereby achieving zero voltage switching. The control scheme of the inverter based on TCM control generally adopts a combination of software and hardware. Taking the positive half cycle of the grid voltage as an example, the upper boundary value of the inductor current is controlled by calculating the conduction time of the corresponding switching device through software, and the lower boundary value of the inductor current is controlled by hardware reset. It has the characteristics of fast, accurate and flexible.

[0004] However, in actual control, after the inductor current drops to the set boundary value, the inductor current further drops due to the resonance between the power inductor and the parasitic output capacitance of the switching device during the dead zone. The actual boundary value of the inductor current has a large deviation from the boundary value set in the digital controller, which affects the overall control of the inductor current, and ultimately causes the output current to be distorted, and the total harmonic distortion (THD) of the current increases. Most existing optimization schemes use the compensation active tube conduction time to offset the negative current generated by resonance, so that the average value of the inductor current is close to the control reference value, thereby optimizing the THD of the output current. This compensation method can improve the THD of the output current, but increases the conduction time of the active tube, resulting in an increase in the ripple of the inductor current, increasing the conduction loss of the power switching device, and affecting the overall efficiency of the inverter. In addition, most compensation methods regard the capacitance of the parasitic output capacitance of the switching device as a constant value, ignoring the influence of its nonlinearity on the control of the inductor current, and have limited improvement on the THD of the inverter output current. Summary of the invention

[0005] The purpose of the present invention is to propose a method for optimizing the THD of inverter output current based on TCM control in view of the defects of the prior art. The present invention takes into account the influence of the parasitic output capacitance of the switching device and the nonlinearity of its capacitance on the inverter output current control, and proposes an adaptive correction method for the boundary value of the inductor current on the basis of TCM control, so as to correct the control error caused by the parasitic output capacitance of the switching device in real time, improve the control accuracy of the inductor current, and optimize the THD of the inverter output current.

[0006] The objective of the present invention is achieved through the following technical solutions:

[0007] A method for optimizing inverter output current THD based on TCM control, comprising:

[0008] Sample the input voltage V on the DC side of the inverter dc , the output voltage v on the AC side o ; and according to the parasitic output capacitance C of the switching device oss With its drain-source voltage v ds The nonlinear relationship curve is used to calculate the charge Q c ;

[0009] Based on the sampling data and the inductor current boundary value correction formula, the lower boundary value I of the inductor current to be set in the digital controller is calculated in real time bs ; Digital controller according to I bs A shutdown signal is sent to the corresponding switch device to correct the inductor current boundary and optimize the inverter output current THD.

[0010] Furthermore, the charge Q c Equal to the parasitic output capacitance C of the switching device oss Charge from 0 volts to v b Accumulated charge Q c , the calculation formula is as follows:

[0011]

[0012] Among them, v b is the boundary voltage, C oss (v ds ) is the parasitic output capacitance C of the switching device oss With its drain-source voltage v ds The non-linear relationship curve.

[0013] Furthermore, the inverter main power circuit topology is a half-bridge inverter, a full-bridge inverter or a T-type three-level inverter.

[0014] Furthermore, the inverter operates in a grid-connected mode or an off-grid mode.

[0015] Furthermore, the inverter modulation mode is unipolar modulation, bipolar modulation or unipolar and bipolar mixed modulation.

[0016] Furthermore, when the inverter modulation mode is unipolar modulation, the inductor current boundary value correction formula is as follows:

[0017]

[0018] When the inverter modulation mode is bipolar modulation, the inductor current boundary value correction formula is as follows:

[0019]

[0020] Where, L is the inductance of the inverter power inductor, I b is the lower boundary value of the inductor current that meets the soft switching condition.

[0021] Furthermore, the inverter is a single-phase inverter or a three-phase inverter.

[0022] Beneficial effects of the present invention:

[0023] (1) By updating the inductor current boundary value set in the digital controller in real time, the control error caused by the parasitic output capacitance of the switching device is corrected, the control accuracy of the inductor current is improved, and the THD of the inverter output current is optimized;

[0024] (2) It only needs to modify the control parameters in real time from the digital control level, without the need for additional control circuits, which is easy to implement and low cost;

[0025] (3) Applicable to a variety of topologies, such as half-bridge inverter, full-bridge inverter, T-type three-level inverter or any other inverter topology that can implement TCM control;

[0026] (4) Applicable to a variety of modulation modes, such as unipolar modulation, bipolar modulation, unipolar and bipolar mixed modulation or any other modulation mode that can achieve TCM control. BRIEF DESCRIPTION OF THE DRAWINGS

[0027] The present invention will be further described in detail below in conjunction with the accompanying drawings and specific embodiments.

[0028] Figure 1 It is a schematic diagram of the topological structure of the full-bridge inverter grid-connected operation;

[0029] Figure 2 It is the ideal waveform of the inductor current of the full-bridge inverter based on TCM control;

[0030] Figure 3 This is the actual waveform of the inductor current of the full-bridge inverter based on TCM control;

[0031] Figure 4 It is the key waveform under unipolar modulation of full-bridge inverter based on TCM control;

[0032] Figure 5 is the parasitic output capacitance of the switching device (C oss ) and its drain-source voltage (v ds )’s typical relationship curve;

[0033] Figure 6 It is the key waveform under bipolar modulation of full-bridge inverter based on TCM control;

[0034] Figure 7 This is a flow chart of the implementation of the inverter output current THD optimization method based on TCM control; DETAILED DESCRIPTION

[0035] The present invention provides an inverter output current THD optimization method based on TCM control, such as Figure 7 As shown, the specific implementation steps are as follows:

[0036] Step 1: Sample the input voltage V on the DC side of the inverter dc , the output voltage v on the AC side o ; Obtain the parasitic output capacitance C of the switch tube according to the switch tube data sheet oss With its drain-source voltage v ds The nonlinear relationship curve of the parasitic output capacitance C is calculated. oss Charge from 0 volts to v b Accumulated charge Q c ;

[0037] Step 2: Calculate the lower boundary value of the inductor current I to be set in the digital controller in real time based on the sampled data and the inductor current boundary value correction formula bs ;

[0038] Step 3: Digital Controller Based on I bs A shutdown signal is sent to the corresponding switch device to correct the inductor current boundary and optimize the inverter output current THD.

[0039] The technical solution of the present invention is further described below in conjunction with the accompanying drawings and specific embodiments, but the present invention is not limited to these embodiments.

[0040] Embodiment: The inverter working mode is grid-connected mode

[0041] Figure 1 This is a schematic diagram of the topological structure of the full-bridge inverter grid-connected operation; Q1-Q4 are power switch tubes, D1-D4 are body diodes, and C oss1 -C oss4is the parasitic output capacitance of the corresponding switch tube, L is the inverter inductor, L o is the output filter inductor, C o is the output filter capacitor, V dc is the DC input voltage, v g is the grid voltage (in grid-connected mode, the grid voltage is the output voltage v on the AC side o ),i L is the inductor current, i g is the grid current (in the grid-connected mode, the grid current is the output current). Since the positive and negative half-cycles of the inverter operate symmetrically within the power frequency cycle, the following analysis takes the positive half-cycle as an example.

[0042] Figure 2 It is the ideal waveform of the inductor current of the full-bridge inverter based on TCM control; the upper boundary value of the inductor current is controlled by calculating the conduction time of the corresponding switch device through software, and the lower boundary value of the inductor current is controlled by hardware reset. By controlling the bidirectional flow of the inductor current, the zero voltage turn-on of the power switch tube is realized, and at the same time, the average value of the inductor current in each switching cycle is always equal to the reference value i of the grid-connected current. ref .

[0043] Figure 3 The actual waveform of the inductor current of the full-bridge inverter based on TCM control. After the inductor current drops to the set boundary value, the inductor current further drops due to the resonance between the power inductor and the parasitic output capacitance of the switching device during the dead zone. The actual inductor current boundary value (I ba ) and the boundary value (I bs ) has a large deviation, which affects the overall control of the inductor current. The average value of the inductor current is not equal to the reference value of the grid-connected current, which eventually leads to distortion of the output current and a larger THD of the grid-connected current. Therefore, the present invention takes into account the influence of the parasitic output capacitance of the power switching device and the nonlinearity of its capacitance on the control of the inverter output current, and proposes an adaptive correction method for the boundary value of the inductor current on the basis of TCM control, which corrects the control error caused by the parasitic output capacitance of the switching device in real time, improves the control accuracy of the inductor current, and optimizes the THD of the inverter output current.

[0044] When the full-bridge inverter adopts a unipolar modulation strategy, Q1 and Q3 operate at high frequencies, and Q2 and Q4 operate at low frequencies. In the positive half cycle, Q2 is normally off and Q4 is normally on. When Q1 and Q4 are turned on, the inductor current rises linearly. After reaching the upper boundary value of the inductor current, Q1 is turned off and Q2 is turned on. The inductor current decreases linearly to the set lower boundary value of the inductor current I bs After reset, each switching cycle is cycled according to this principle. In actual application, the driving signal (v gs_Q1 ) and Q3’s drive signal (vgs_Q3 ) between the key waveforms such as Figure 4 As shown in the figure, in the [t0-t1] stage, the inductor current is large, and the parasitic output capacitor charging and discharging process is very short, which has a very small impact on the overall control of the inductor current and can be ignored; in the [t2-t4] stage, the inductor current is small, and the parasitic output capacitor charging and discharging process is long, which has a large impact on the overall control of the inductor current and cannot be ignored. During the [t2-t4] period, the parasitic output capacitor of Q1 is discharged, and its drain-source voltage (v ds_Q1 ) by V dc Gradually drops to 0; the parasitic output capacitance of Q3 is charged, and its drain-source voltage gradually rises from 0 to V dc .

[0045] Figure 5 is the parasitic output capacitance of the switching device (C oss ) and its drain-source voltage (v ds ) is a typical relationship curve. Among them, v b Defined as the boundary voltage; when v ds <v b , the parasitic output capacitance C of the switch tube oss With its drain and source voltage v ds decreases and increases sharply, which is recorded as area I; when v ds >v b , the parasitic output capacitance C of the switch tube oss With its drain and source voltage v ds The capacitance decreases slowly with the increase of capacitance, and the capacitance can be regarded as a constant value, which is recorded as region II.

[0046] Furthermore, considering the impact of the nonlinearity of the parasitic output capacitor on the inverter output current control, Figure 4 In the [t2-t3] stage, when the inductor current drops to the set boundary current I bs After that, due to the low drain and source voltage of Q3, the corresponding Figure 5 In region I, the parasitic output capacitor has a large capacitance, which results in a longer charging time and a lower bridge arm midpoint voltage V AB The voltage across the inverter inductor is maintained at -v g , the inductor current will continue to be approximately equal to -v g The slope of / L decreases linearly to time t3, causing the actual boundary current to deviate from the set value I bs ; [t3-t4] stage corresponds to Figure 5 In region II, the capacitance of the parasitic output capacitor is small, the charging time is very short, and the influence on the control of the inductor current is small and can be ignored.

[0047] Combination Figure 4 From the above analysis, we can get the actual boundary value of the inductor current Iba With the set boundary value I bs The relationship between:

[0048]

[0049] Where, Δt = t3 - t2;

[0050] The inductor current charges the parasitic output capacitance of Q3, and its drain-source voltage increases from 0 to v b , according to the charge conservation law, we can get

[0051]

[0052] remember Combining equations (1) and (2), we can get:

[0053]

[0054] In actual control, the actual inductor current boundary value I is required ba Equal to the ideal inductor current boundary value I b ; Therefore, I ba =I b Substituting into formula (3), we get:

[0055]

[0056] According to formula (4), the set value of the inductor current boundary can be updated in real time following the grid voltage, so that the actual inductor current boundary value is close to the ideal value, the control error caused by the parasitic output capacitance of the switching device is corrected in real time, the control accuracy of the inductor current is improved, and the THD of the inverter output current is optimized.

[0057] When the full-bridge inverter adopts a bipolar modulation strategy, Q1 and Q4 operate synchronously at high frequency, and Q2 and Q3 operate synchronously at high frequency. In the positive half cycle, when Q1 and Q4 are turned on, the inductor current rises linearly. After reaching the upper boundary value of the inductor current, Q1 and Q4 are turned off, and Q2 and Q3 are turned on. The inductor current decreases linearly to the set lower boundary value of the inductor current I bs After reset, each switching cycle is cycled according to this principle. In actual application, the driving signal (v gs_Q1 ) and Q3’s drive signal (v gs_Q3 ) between the key waveforms such as Figure 6As shown. In the [t0-t1] stage, the inductor current is large, and the parasitic output capacitor charging and discharging process is very short, which has a very small impact on the overall control of the inductor current and can be ignored; in the [t2-t4] stage, the inductor current is small, and the parasitic output capacitor charging and discharging process is long, which has a large impact on the overall control of the inductor current and cannot be ignored. During the [t2-t4] period, the parasitic output capacitors of Q1 and Q4 are discharged, and their drain-source voltage (v ds_Q1 ) by V dc Gradually drops to 0; the parasitic output capacitance of Q2 and Q3 is charged, and the drain and source voltages gradually rise from 0 to V dc .

[0058] Furthermore, considering the impact of the nonlinearity of the parasitic output capacitor on the inverter output current control, Figure 6 In the [t2-t3] stage, when the inductor current drops to the set boundary current I bs After that, due to the low drain and source voltages of Q2 and Q3, the corresponding Figure 5 In region I, the parasitic output capacitor has a large capacitance, which results in a longer charging time and a lower bridge arm midpoint voltage V AB The voltage across the inverter inductor is maintained at -(V dc +v g ), the inductor current will continue to be approximately equal to -(V dc +v g ) / L decreases linearly to time t3, causing the actual boundary current to deviate from the set value I bs ; [t3-t4] stage corresponds to Figure 5 In region II, the capacitance of the parasitic output capacitor is small, the charging time is very short, and the influence on the control of the inductor current is small and can be ignored.

[0059] Combination Figure 6 From the above analysis, we can get the actual boundary value of the inductor current I ba With the set boundary value I bs The relationship between:

[0060]

[0061] Where, Δt = t3 - t2;

[0062] The inductor current charges the parasitic output capacitance of Q2 and Q3, and the drain and source voltages of both increase from 0 to v b , according to the charge conservation law, we can get

[0063]

[0064] remember Combining equations (5) and (6), we can get:

[0065]

[0066] Similarly, in actual control, the actual inductor current boundary value I ba Equal to the ideal inductor current boundary value I b ; Therefore, I ba =I b Substituting into formula (7), we get:

[0067]

[0068] According to formula (8), the set value of the inductor current boundary under bipolar modulation can be updated in real time following the grid voltage, so that the actual inductor current boundary value is close to the ideal value, the control error caused by the parasitic output capacitance of the switching device is corrected in real time, the control accuracy of the inductor current is improved, and the THD of the inverter output current is optimized.

[0069] The above descriptions are only some implementation cases of the present invention and are not intended to limit the present invention in other forms. Technicians in the relevant technical field can make various modifications or supplements to the specific embodiments described or replace them in a similar manner. However, any simple modifications and equivalent changes made to the above embodiments based on the technical essence of the present invention without departing from the content of the technical solution of the present invention still fall within the protection scope of the technical solution of the present invention.

Claims

1. A method for optimizing inverter output current THD based on TCM control, characterized in that: include: Sample the input voltage V on the DC side of the inverter dc , the output voltage v on the AC side o ; and according to the parasitic output capacitance C of the switching device oss With its drain-source voltage v ds The nonlinear relationship curve is used to calculate the charge Q c ; Based on the sampling data and the inductor current boundary value correction formula, the lower boundary value I of the inductor current to be set in the digital controller is calculated in real time bs ; Digital controller according to I bs A shutdown signal is sent to the corresponding switch device to correct the inductor current boundary and optimize the inverter output current THD.

2. The method according to claim 1, characterized in that The charge Q c Equal to the parasitic output capacitance C of the switching device oss Charge from 0 volts to v b Accumulated charge Q c , the calculation formula is as follows: Among them, v b is the boundary voltage, C oss (v ds ) is the parasitic output capacitance C of the switching device oss With its drain-source voltage v ds The non-linear relationship curve.

3. The method according to claim 1, characterized in that The inverter main power circuit topology is a half-bridge inverter, a full-bridge inverter or a T-type three-level inverter.

4. The method according to claim 1, characterized in that The inverter working mode is a grid-connected mode or an off-grid mode.

5. The method according to claim 1, characterized in that The inverter modulation mode is unipolar modulation, bipolar modulation or unipolar and bipolar mixed modulation.

6. The method according to claim 5, characterized in that When the inverter modulation mode is unipolar modulation, the inductor current boundary value correction formula is as follows: When the inverter modulation mode is bipolar modulation, the inductor current boundary value correction formula is as follows: Where, L is the inductance of the inverter power inductor, I b is the lower boundary value of the inductor current that meets the soft switching condition.

7. The method according to claim 1, characterized in that The inverter is a single-phase inverter or a three-phase inverter.

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