Electric motor control device, electric vehicle and hybrid electric vehicle

Abstract

Sideband noise is reduced, and furthermore, electrical power loss is suppressed. By providing an inverter 19m, which is arranged between an electric motor 10m and a DC power source 18 to 23, and controlling the exchange of electrical power between the electric motor and the DC power source by switching the inverter via PWM pulses, a carrier frequency of the PWM pulses is determined as a high frequency k·fc of 7.5 kHz, at which sideband noise is low when a target torque T* and a speed ω of the electric motor are in a first range A and B to suppress sideband noise, whereas the carrier frequency is determined to be a low frequency fc of 5 kHz, which is lower than the high frequency, and reduces switching loss of the inverter when the target torque and speed are in a second range that is outside the first range.and the coil voltages of the electric motor are PWM-controlled, so that the output torque of the electric motor matches the target torque. The high frequency is kx fc, where 1 <k<2.,

Description

Technical field

[0001] The present invention relates to an electric motor control device that PWM controls the power supplied by an inverter to an electric motor, and in particular the control of a voltage control mode and a carrier frequency of PWM pulses. The electric motor control device of the present invention can be used, for example, in an electric vehicle (EV) that has an electric motor to drive wheels, and in a hybrid electric vehicle (HEV) that, in addition to the electric motor, is equipped with a combustion engine and a generator (also referred to as a motor or motor-generator) that is rotary-driven by the combustion engine. Background technology

[0002] Patent document 1 describes an electric motor drive that uses a two-phase modulation system in which one of the three phase voltages applied to an electric motor is fixed at a high level (H) or a neutral level (N), and the remaining two phases are PWM-controlled. The two-phase modulation introduces distortion into the waveform of the voltage applied to the electric motor, causing noise and a degradation of performance. To mitigate these problems, patent document 1 presents a control system for reducing the waveform distortion. Furthermore, since the modulation becomes equivalent to three-phase modulation when the displacement of a triangular wave is set to zero, a PWM waveform is generated within the two-phase modulation.Patent document 1 presents a switching control between two-phase and three-phase operation, wherein the modulation system is switched to the three-phase modulation system by specifying the displacement magnitude to 0. In a range where the motor speed is comparatively low, undesirable high-frequency noise is high when the carrier frequency of the PWM pulses is low, whereas switching losses in an inverter increase when the carrier frequency is increased. Consequently, to improve upon these problems, patent document 2 presents a motor control device that increases the carrier frequency in the range where the motor speed is comparatively low and decreases the carrier frequency in the range where the motor speed is high.Additionally, in the high-speed range, the motor is powered by square waves, and in the low-speed range, it is powered by sine waves, partial sine waves, or an overlapping sine wave. Patent document 3 describes an inverter control that uses three-phase modulation and a high carrier frequency (7 kHz) when high target torque is required at high speed, and two-phase modulation and a low carrier frequency (5 kHz) in other cases. Patent document 3 also describes a selection between two-phase and three-phase modulation, as well as a switching of the carrier frequency, and shows in the [references / references]. Fig. 4, Fig. 5, Fig. 6 to Fig. Seven voltage waveforms are applied to the electric motor, each in two-phase modulation with a carrier frequency of 5 kHz, in three-phase modulation with a carrier frequency of 5 kHz, in three-phase modulation with a carrier frequency of 7 kHz, and in two-phase modulation with a carrier frequency of 7 kHz. Patent document 3 shows in Fig. 8 also a hysteresis in a frequency switching between the two-phase modulation at 5 kHz and the three-phase modulation at 7 kHz. [Patent Document 1] Japanese Patent Application Publication No. JP H07-303302A [Patent document 2] Japanese Patent No. JP 3 837 986 B2 [Patent Document 3] Japanese Patent Application Publication No. JP 2004-289985A

[0003] US 2004 / 0207360 A1 concerns an inverter control in which a modulation mode is switched to two-phase modulation and a carrier frequency is set to 5 kHz when a motor speed command value is low, and a modulation mode is switched to three-phase modulation and the carrier frequency is set to 7 kHz to suppress harmonic components when the motor speed command value increases.

[0004] DE 11 2005 000 076 T5 relates to a method for managing a processor execution time in a motor controller wherein a motor speed range is determined on the basis of a comparison of received motor speed data with predetermined motor speed data.

[0005] US 2005 / 0194925 A1 relates to a device for controlling a three-phase AC motor using a two-phase modulation technique.

[0006] JP 2005 000 510 A relates to a method for reducing vibrations and noise caused by driving a motor.

[0007] JP 2004 048 885 A relates to a power converter for precise current control while suppressing switching losses in order to convert a DC voltage into an AC voltage using pulse width modulation.

[0008] JP 2007 110780 A relates to a motor control system in which heat generation from switching components can be compensated by switching a two-phase modulation control system to a three-phase modulation control system depending on the operating state of a motor.

[0009] JP 2005 117869 A relates to a two-phase modulation control inverter device.

[0010] JP H01-274669 A relates to a control method in which 3-phase modulation is used when a modulation rate is small, and 2-phase modulation is used when a modulation rate is large. Disclosure of the invention [The problem to be solved by the invention]

[0011] It is a problem that sideband noise is present in an audio frequency band while a vehicle is traveling at speeds ranging from 20 mph (miles per hour) to 50 mph (with an engine speed of 2000 rpm to 8000 rpm and a torque of -100 Nm to +100 Nm within a certain engine speed range). One known method for reducing the vehicle's sideband noise through inverter control is to increase the carrier frequency. For example, according to patent document 2, the noise is reduced by increasing the carrier frequency in the low-speed range. Generally, the modulation system (voltage control mode) is not changed when the carrier frequency is switched to suppress the sideband noise. However, according to patent document 3, the switching is performed between two-phase modulation at 5 kHz and three-phase modulation at 7 kHz ( Fig. 8 and Fig. 9, paragraph 0059), describes how switching is used to suppress a protection operation (shutdown) caused by a sharp drop in the inverter input voltage.

[0012] If the carrier frequency is increased to reduce sideband noise without changing the voltage modulation mode, the inverter's switching losses can increase, causing it to overheat. Furthermore, the power loss of the electric motor drive increases. In an inverter control system where switching is performed between two-phase modulation at 5 kHz and three-phase modulation at 7 kHz, as described in patent document 3, the two-phase modulation involves only a small number of switching operations compared to the total number of three-phase switching operations. Similarly, the 5 kHz carrier frequency requires only a small number of switching operations, consequently generating a small switching loss, while the sideband noise is high. Therefore, driving in three-phase modulation at 5 kHz is expected to generate a small power loss but a large amount of noise.Three-phase modulation involves a large number of switching operations, and the carrier frequency of 7 kHz also generates significant switching losses, even though the noise is low. Consequently, when driving a three-phase modulation signal at 7 kHz, low noise but high power loss is expected. Therefore, both noise reduction and power loss reduction are desirable.

[0013] One object of the present invention is to reduce sideband noise and also to suppress power loss.

[0014] The above problem is solved by an electric motor control device, a drive unit, and a hybrid drive unit according to the independent claims. Advantageous embodiments are specified in the dependent claims.

[0015] Specifically, when creating an inverter (19m) that is connected between an electric motor (10m) and a DC power source (18 to 23) and controls the exchange of electrical power between the electric motor and the DC power source by switching the inverter with PWM pulses, a carrier frequency of the PWM pulses is determined as a high frequency (7.5 kHz) at which the sideband noise is low when a target torque (TM*m) and a speed (ωm) of the electric motor are within a predetermined range (A, B) set to suppress sideband noise, whereas the carrier frequency is determined as a low frequency (5 kHz) that is lower than the high frequency (7.5 kHz) and reduces the switching loss of the inverter when the target torque and speed are outside the predetermined range, and coil voltages, i.e. phase voltages of the electric motor, are PWM-controlled.so that the output torque of the electric motor matches the target torque. An electric motor control device for performing the above function according to a first aspect of the present invention is described in the next section, which is numbered (1). (1) An electric motor control device comprises: a DC power source (18 to 23 and 40); an inverter (19m) connected between an electric motor (10m) and the DC power source to control an exchange of electrical power between them; an inverter control means (50 and 20m) generating PWM pulses with a relative duty cycle corresponding to voltage command signals (VU*, VV* and VW*) and sending the PWM pulses to the inverter to switch the inverter; a frequency change means (46) changing a carrier frequency of the PWM pulses generated by the inverter control means in a manner corresponding to a frequency control signal (FRf); and a motor control means which, when a target torque (T*) and speed (ω) of the electric motor are within a predetermined range,the frequency-changing device (46) provides the frequency control signal (FRf) for setting the carrier frequency to a high frequency (k·fc: 7.5 kHz), and during this switching, if a voltage control mode for controlling the three-phase voltages of the electric motor is a three-phase modulation mode in which each of the three-phase voltages is controlled by PWM, switches the mode to a two-phase modulation mode (A) in which two phases are controlled by PWM, while the application of PWM switching for the other phase is stopped, and if the target torque and speed are outside the predetermined range in a case where the first carrier frequency is the high frequency, frequency-changing device (46) provides the frequency control signal (FRf) for setting the carrier frequency to a low frequency (fc: 5 kHz) which is lower than the high frequency, and switches the modulation mode to the three-phase modulation mode (SVpwm),when a condition for switching to three-phase modulation mode is met, and the inverter control unit provides the voltage command signals (VU*, VV* and VW*) to match an output torque of the electric motor with the target torque.

[0016] Note that, for better understanding, reference symbols corresponding to elements or units of a later described embodiment and shown in the drawings are included in parentheses in the above illustrations for reference. The same applies below. [Effects of the invention]

[0017] According to the above, sideband noise can be suppressed by switching to the high frequency (7.5 kHz), and switching loss can also be suppressed due to switching to the two-phase modulation mode (A) at the same time as switching to the high frequency (7.5 kHz).

[0018] (2) The electric motor control device according to section (1) above, wherein, when the voltage control mode is the two-phase modulation mode (B), the motor control means (30m) continues the two-phase modulation mode even after switching to the high frequency when the carrier frequency is switched from the low frequency to the high frequency.

[0019] According to the above, since the switching loss is already small due to the continuation of the two-phase modulation mode, the switching loss will not be large even after switching to the high frequency to suppress sideband noise.

[0020] (3) The electric motor control device according to section (1) above, wherein the high frequency is k·fc, wherein 1 <k<2, wobei die Niederfrequenz als fc bezeichnet ist. Gemäß dem Obigen können Spulenströme des Elektromotors einen Kernverlust (Eisenverlust) und den Wechselrichterschaltverlust, der durch harmonische Verzerrung verursacht wird, zu kleinen Werten unterdrücken.

[0021] (4) Electric motor control device according to section (1) above, wherein the motor control means (30m) determines the voltage control mode based on a modulation rate (Mi = Vm* / Vuc) which is a ratio of an electric motor target voltage (Vm*) to a DC voltage (Vuc) supplied to the inverter by the DC power source, and based on a speed (ω) of the electric motor, and expands a range for the modulation rate and speed in which the two-phase modulation mode is carried out by changing the modulation mode from the three-phase modulation mode to the two-phase modulation mode when the carrier frequency is the high frequency in the three-phase modulation mode, or when the carrier frequency is the high frequency and the voltage control mode is determined as the three-phase modulation mode.

[0022] (5) Electric motor control device according to paragraph (1) above, wherein the predetermined area comprises a first area (A + B) and a second area (area that is in Fig. 7 enclosed by the dashed line), which contains the first region and is wider than the first region, and in which the motor control means (30m) switches the carrier frequency at the low frequency to the high frequency when the target torque and speed of the electric motor are designed to be in the first region, and switches the carrier frequency at the high frequency to the low frequency when the target torque and speed of the electric motor are designed to be outside the second region.

[0023] According to the above, a delay can be prevented if switching frequently occurs from outside the predetermined area into its interior, or vice versa.

[0024] (6) Electric motor control device according to paragraph (5) above, wherein, during a first switching operation in which the carrier frequency is switched from low frequency to high frequency and the voltage control mode is also switched from three-phase modulation to two-phase modulation, the motor control means (30m) corrects the target torque by adding a first torque correction value to the target torque to reduce one torque step between before and after the switching operation, whereas during a second switching operation in the opposite manner to the first switching operation, the motor control means (30m) corrects the target torque by adding a second torque correction value to the target torque to reduce one torque step between before and after the switching operation, and the motor control means (30m) provides the voltage command signals to the inverter control means.to bring the output torque of the electric motor into line with the corrected target torque.

[0025] According to the above, the torque step generated by switching from three-phase to two-phase modulation can be reduced. This reduction is determined by the hysteresis in the carrier frequency switching, which is assessed based on the first region (whether a low / high shift of the carrier frequency is necessary) and the second region (located outside the first region), which also determines whether a high / low shift of the carrier frequency is necessary. The torque step generated by switching from two-phase to three-phase modulation can also be reduced.

[0026] (7) Electric motor control device according to paragraph (1) above, wherein the predetermined range is a range for suppressing sideband noise and the high frequency is a frequency for reducing sideband noise.

[0027] (8) Electric motor control device according to paragraph (1) above, wherein the low frequency is a frequency to reduce the switching loss of the inverter.

[0028] (9) Drive unit comprising the electric motor control device according to any of paragraphs (1) to (8) above, and also comprising the electric motor (10m) which is powered by the inverter (19m) included in the electric motor control device and drives the wheels. According to the above, the operation and advantageous effects as described in paragraph (1) above can be obtained, for example, in the drive unit mounted on an electric vehicle.

[0029] (10) Hybrid drive unit comprising: A DC power source (18 to 23, and 40); a first electric motor (10m) driving wheels; a second electric motor (10g) driven by a combustion engine; a first inverter (19m) connected between the first electric motor and the DC power source to control an exchange of electrical power between them; a second inverter (19g) connected between the second electric motor and the DC power source to control an exchange of electrical power between them; a first inverter control device (50 and 20m) generating first PWM pulses having a duty cycle corresponding to the first voltage command signals (VU*, VV* and VW*) and sending the first PWM pulses to the first inverter to switch the first inverter; a second inverter control device generating second PWM pulses,which have a duty cycle corresponding to the second voltage command signals, and send the second PWM pulses to the second inverter to switch the second inverter; a first frequency-changing means (46) that modifies a first carrier frequency of the first PWM pulses generated by the first inverter control means in a manner corresponding to a first frequency control signal (FRf); a second frequency-changing means that modifies a second carrier frequency of the second PWM pulses generated by the second inverter control means in a manner corresponding to a second frequency control signal; a first motor control means (30m) that, when a target torque and speed of the first electric motor (10m) are within a first predetermined range,The first frequency control signal (FRf) is provided to the frequency-changing device (46) to set the first carrier frequency to a high frequency, and during this switching, if a voltage control mode for controlling three-phase voltages of the first electric motor (10m) is a three-phase modulation mode in which each of the three-phase voltages is controlled by PWM, the mode is switched to a two-phase modulation mode (A) in which two phases are controlled by PWM, while the application of PWM switching for the other phase is stopped if the target torque and speed are outside the predetermined range, in a case where the first carrier frequency is the high frequency, the first frequency-changing device (46) is provided with the first frequency control signal (FRf) to set the first carrier frequency to a low frequency which is less than the high frequency, and the modulation mode is switched to the three-phase modulation mode (SVpwm).when a condition for switching to the three-phase modulation mode is met, and provides the first inverter control means (50 and 20m) with the first voltage command signals (VU*, VV* and VW*) to bring an output torque of the first electric motor (10m) into accordance with the target torque; and a second motor control means (30g) which, when a target torque and speed of the second electric motor (10g) are within a predetermined range, provides the frequency change means with the second frequency control signal to set the second carrier frequency to a high frequency, and during this switching, if a voltage control mode for controlling the three-phase voltages of the second electric motor is a three-phase modulation mode in which each of the three-phase voltages is controlled by PWM, switches the mode to a two-phase modulation mode in which two phases are controlled by PWM,While the application of PWM control to the other phase is stopped if the target torque and speed are outside the predetermined range, in the case that the second carrier frequency is the high frequency, the second frequency-changing device provides the second frequency control signal to set the second carrier frequency to a low frequency lower than the high frequency, and switches the modulation mode to the three-phase modulation mode when a condition for switching to the three-phase modulation mode is met, and provides the second voltage command signals to the second inverter control device to bring an output torque of the second electric motor into conformity with the target torque. According to the above, the operation and advantageous effects as described in section (1) above can be obtained, for example, in the drive unit mounted on a hybrid electric vehicle. Brief description of the drawings [ Fig. 1] Fig. Figure 1 shows a block diagram that provides a simplified representation of a structure according to a first embodiment of the present invention. [ Fig. 2] Fig. Figure 2 shows a block diagram that provides a brief overview of the functional structure of a 30m motor control device, as shown in Fig. 1 shown, shows. [ Fig. 3] Fig. Figure 3 shows a flowchart that provides a brief overview of engine control by a microprocessor MPU, as in Fig. 2 shown, shows. [ Fig. 4] Fig. Figure 4 shows a flowchart detailing the “modulation control” section 9, as described in Fig. 3 shown, shows. [ Fig. 5] Fig. Figure 5 shows a flowchart illustrating the first half of the details of the “Determination of the Carrier Frequency and Modulation Mode” 22, as described in Fig. 4 shown, shows. [ Fig. 6] Fig. Figure 6 shows a flowchart illustrating the second half of the details of the “Determination of the Carrier Frequency and Modulation Mode” 22, as described in Fig. 4 shown, shows. [ Fig. 7] Fig. Figure 7 shows a graph with coordinate axes representing the torque and rotational speed of an electric motor 10m, where the graph shows a first region (A + B) where a high carrier frequency is used, a boundary (dashed line) of a region where the high carrier frequency is deleted, and modulation mode regions. Fig. 7 is applied to a case where a secondary-side voltage Vuc of a converter 40 is equal to 220 V. [ Fig. 8] Fig. Figure 8 shows a graph that shows the first region (A + B) where the high carrier frequency is used, the boundary (dashed line) of the region where the high carrier frequency is deleted, and modulation mode regions. Fig. 8 is applied to a case where the secondary-side voltage Vuc of converter 40 is equal to 300 V. [ Fig. 9] Fig. Figure 9 shows a graph that shows the first region (A + B) where the high carrier frequency is used, the boundary (dashed line) of the region where the high carrier frequency is deleted, and modulation mode regions. Fig. 9 is applied to a case where the secondary-side voltage Vuc of converter 40 is equal to 400V. [ Fig. 10] Fig. Figure 10 shows a graph illustrating the combined power losses of the electric motor (10m) and an inverter (19m), as shown in Fig. 1 shown, shows. [Description of reference symbols] 10m and 10g electric motors 11 to 13 three-phase stator coils 14m to 16m current sensors 17m and 17g rotary indicator 18 batteries that are mounted on the vehicle 21 primary-side voltage sensor 22 primary side capacitor 23 secondary-side capacitor 24 secondary-side voltage sensor 25 secondary-side current sensor 34 Subtraction 35 Addition 41 Throttle or throttle coil 42 Switching element (for voltage step upwards) 43 Switching element (for downward voltage step) 44 and 45 diodes ωm and ωg rotational speeds Vdc primary side voltage (battery voltage) Vuc secondary-side voltage (upward-stepped voltage) Best embodiments of the invention

[0030] Other purposes and features of the present invention will become clear from the description of the following embodiment with reference to the drawings.

[0031] Fig. Figure 1 shows a simplified representation of a first embodiment of the present invention. In this embodiment, the electric motor 10m to be controlled is a permanent magnet synchronous motor mounted on a vehicle for driving the wheels. It has a rotor with a permanent magnet embedded within it and a stator with three phase coils 11 to 13: a U-phase, a V-phase, and a W-phase, respectively. A voltage-type inverter 19m supplies power from a battery 18, mounted on the vehicle, to the electric motor 10m. A rotor of a rotary encoder 17m for detecting the magnetic pole position of the rotor is connected to the rotor of the electric motor 10m. The rotary encoder 17m generates an analog voltage (rotation angle signal) SGθm, representing a rotation angle of the rotor, and supplies this analog voltage to a motor control device 30m.

[0032] When power is switched on for a vehicle electrical system, a primary-side capacitor 22 is connected to the battery 18, which serves as an electric accumulator in the vehicle, thus forming a primary-side DC power source together with the battery 18. A voltage sensor 21 provides the motor control devices 30m and 30g with a voltage detection signal Vdc, which represents the voltage of the primary-side capacitor 22 (voltage of the battery 18 in the vehicle). In the present embodiment, the current sensor 21 uses a voltage divider resistor. One end of an inductor 41 of a converter 40 is connected to a positive terminal (positive lead) of the primary-side DC power source.

[0033] The converter 40 additionally has a semiconductor switch 42 for an upward voltage step, which serves as a switching element for an upward voltage step to switch on and off between the other end of the inductor 41 and a negative terminal (negative line) of the primary-side DC power source, a semiconductor switch 43 for regeneration, which serves as a switching element for a downward voltage step to switch on and off between a positive terminal of a secondary-side capacitor 23 and the other end of the inductor 41, and diodes 44 and 45, which are each connected in parallel to the semiconductor switches 42 and 43.

[0034] When the semiconductor switch 42 is turned on (conducting) for a voltage step upwards, a current flows from the primary-side DC power source 18, 22 through the inductor 41 to the switch 42, resulting in the charging of the inductor 41; whereas when the switch 42 is turned off (non-conducting), the inductor 41 discharges a high voltage to the secondary-side capacitor 23 via the diode 45. Thus, a higher voltage than the voltage of the primary-side DC power source is induced to charge the secondary-side capacitor 23. The high-voltage charging of the secondary-side capacitor 23 continues by repeatedly turning the switch 42 on and off. The secondary-side capacitor 23 is therefore charged at the high voltage.Since the energy stored by the choke coil 41 increases according to the duration that the switch is on, while the switching on and off is repeated with a constant time period, it is possible to adjust the rate (energy supply rate for power operation) at which the power from the primary-side DC power source 18, 22 is supplied through the converter 40 to the secondary-side capacitor 23 by adjusting the time that the switch is on within the constant time period (on-time: ratio of the time that the switch is on to the constant time period), i.e., by PWM control.

[0035] When the semiconductor switch 43 is switched on (conducting) for regeneration, the power stored in the secondary-side capacitor 23 is supplied to the primary-side DC power source 18, 22 via the switch 43 and the inductor 41 (reverse power supply: regeneration). In this case as well, it is possible to adjust the rate (regeneration power supply rate) at which the power is supplied in reverse from the secondary-side capacitor 23 through the converter 40 to the primary-side DC power source 18, 22 by adjusting the time during which the switch 43 is switched on within the constant time period, i.e., by PWM control.

[0036] The voltage-type inverter 19m is provided with six switching transistors Tr1 to Tr6, which are switched on (conducting) to be driven by six sequences of control signals generated in parallel by a driver circuit 20m. These signals convert the DC voltage of the secondary-side capacitor 23 (output voltage of the converter 40, i.e., the secondary-side voltage) into three sequences of AC voltage with phase differences of 2π / 3 between the sequences, i.e., into a three-phase AC voltage. The three sequences are then applied to the stator coils 11 to 13 of the three phases (U-phase, V-phase, and W-phase) of the electric motor 10m. As a result, the phase currents iUm, iVm, and iWm flow into the stator coils 11 to 13 of the electric motor 10m, and the rotor of the electric motor 10m rotates.To increase the capacity of the power supply for switching transistors Tr1 to Tr6 on and off with PWM pulses, and to suppress surge voltage, the large-capacitance secondary-side capacitor 23 is connected to a secondary-side output line of converter 40, which serves as an input line of converter 19m. On the other hand, the primary-side capacitor 22, which forms the primary-side DC power source, is small, inexpensive, and has a low capacitance, significantly smaller than that of the secondary-side capacitor 23. Current sensors 14m to 16m are mounted on the power supply lines connected to the stator coils 11 to 13 of the electric motor 10m, using hole ICs that detect the phase currents iUm, iVm and iWm respectively to generate current detection signals (alologous voltages) which are provided to the motor control device 30m.

[0037] Fig. Figure 2 shows a functional structure of the motor control device 30m. In the present embodiment, the motor control device 30m is an electronic control device, mainly comprising a microcomputer MPU and the drive circuit 20m, current sensors 14m to 16m, the rotary encoder 17m, the primary-side voltage sensor 21, and an interface (signal processing circuit, not shown) between a secondary-side voltage sensor 24 and a secondary-side current sensor 25. The motor control device 30m further includes an interface (communication circuit, not shown) between the microcomputer and a main controller of a vehicle drive control system (not shown) on the vehicle. Note that the secondary-side voltage sensor 24, as shown in Fig. Figure 1 shows that a secondary-side voltage Vuc (of the secondary-side capacitor 23) is detected, and the corresponding voltage signal Vuc is provided to the motor control circuits 30m and 30g.

[0038] Referring to Fig. 2 can be seen that the microcomputer in the motor control device 30m calculates a rotation angle (magnetic pole position) θm and a rotational speed (angular velocity) ωm of the rotor of the electric motor 10m based on the rotation angle signal SGθm provided by the rotation detector 17m.

[0039] Note that although the rotation angle of the 10m electric motor's rotor is not exactly the same as the magnetic pole position, they are proportionally related, and the coefficient of proportionality is determined by the number of magnetic poles p of the 10m electric motor. Furthermore, although the rotational speed is not equal to the angular velocity, they are also proportionally related, and the coefficient of proportionality is determined by the number of magnetic poles p of the 10m electric motor. In this document, the rotation angle θm refers to the magnetic pole position. The rotational speed ωm refers to the angular velocity, but sometimes also to the rotational speed.

[0040] The microcomputer of the motor control device 30m performs feedback control by carrying out a vector control calculation in the "output calculation" 35 for a known dq-axis model, in which the d-axis corresponds to the direction of a pair of magnetic poles in the rotor of the electric motor 10m, and the q-axis is set to be perpendicular to the d-axis. The microcomputer converts to digital and reads the current detection signals iUm, iVm, and iWm from the current sensors 14m to 16m, and in the current feedback calculation converts the three-phase current values ​​iUm, iVm, and iWm on the fixed coordinates into two-phase current values ​​idm and iqm on the d-axis and q-axis of the rotational coordinates, respectively, using a three-phase / two-phase conversion, which is a known fixed / rotational coordinate transformation.

[0041] A main controller (not shown) of the vehicle propulsion control system provides a target motor torque TM*m to the microcomputer of the motor control device 30m. Note that the main controller calculates a required vehicle torque TO*m based on the vehicle speed and the vehicle's accelerator pedal input, and generates the target motor torque TM*m according to the required vehicle torque TO*m, which is then provided to the microcomputer. The microcomputer provides the rotational speed ω rpm of the electric motor 10m as an output signal to the main controller.

[0042] The microcomputer of the motor control device 30m adds a torque correction value, described later, to the motor target torque TM*m at a summation point 33. Then, in a torque command limit 34, the microcomputer of the motor control device 30m reads a limit torque TM*mmax, corresponding to the secondary-side voltage Vuc and the rotational speed ωm, from a limit torque table (lookup table) and sets TM*mmax as the target torque T*m if the target torque TM*m, corrected by the summation point 33, exceeds TM*mmax. The motor target torque TM*m, corrected by the summation point 33, is set as the target torque T*m if the corrected motor target torque TM*m is less than or equal to TM*mmax. The motor target torque T*m generated by the limit described above is then provided to the output calculation 35.

[0043] Note that the limit torque table is a memory area in which the values ​​of the voltage Vuc and the rotational speed ωm within the variable range of the secondary-side voltage Vuc and the rotational speed range ωm, respectively, are used as addresses, and at each of the address values, the maximum torque that can be generated by the electric motor 10m is written as the limit torque TM*mmax. Note also that the memory area represents a memory area of ​​a RAM (not shown) in the microcomputer, according to the present embodiment. The larger the secondary-side voltage Vuc, the larger the limit torque TM*mmax, whereas the smaller the secondary-side voltage Vuc, the smaller the limit torque TM*mmax. Likewise, the smaller the rotational speed ωm, the larger the limit torque TM*mmax, whereas the larger the rotational speed ωm, the smaller the limit torque TM*mmax.

[0044] The microcomputer described above contains a non-volatile memory in which the data TM*mmax of the limit torque table are written, and the microcomputer reads the data from the non-volatile memory to write it into the RAM in the process in which an operating voltage is applied to the microcomputer to start it and a motor drive system, as in Fig. Figure 1 shows how to initialize the microcomputer. It contains a number of other similar lookup tables, which, like the limit torque table, represent memory areas in the RAM where the reference data stored in the non-volatile memory is written, as will be described later.

[0045] The initial calculation 35 contains a first high-efficiency torque curve table A, which serves as a reference table. The first high-efficiency torque curve table A is described with d-axis current values ​​id, each corresponding to the motor speed ωm and the motor target torque T*m for generating each of the target torques T*m at each corresponding motor speed.

[0046] The output torque of the electric motor is determined according to each value of the d-axis current id and the q-axis current iq. There are infinitely many combinations of id and iq that can generate the same output torque for a single speed value, i.e., at the same motor speed, and these combinations lie on a constant torque curve. There is one combination of id and iq with maximum power utilization efficiency (minimal power consumption) on the constant torque curve, and this combination represents a high-efficiency torque point. A curve that connects high-efficiency torque points on a plurality of torque curves is a high-efficiency torque curve, which exists for every speed.By setting the target current values ​​to the d-axis current id and the q-axis current iq according to the position of a given value of the motor target torque T*m on the high-efficiency torque curve for a motor speed, and energizing the electric motor 10m based on the target current, the electric motor 10m generates an output torque of the target value T*m with a high-performance utilization efficiency for motor energizing.

[0047] In the present embodiment, the high-efficiency torque curve is divided into two curves: a first high-efficiency torque curve A, representing the d-axis value, and a second high-efficiency torque curve B, representing the q-axis value. The first high-efficiency torque curve A contains a pair of curves applied to a power operating range and a regeneration range, respectively; both represent the d-axis target current with respect to the engine speed and the target torque.

[0048] The first high-efficiency torque curve table A is a storage area in which the d-axis target current is inscribed according to the target torque T*m for generating the target torque with minimal power consumption. It is formed from a pair of a power run table A1 for a power run and a regeneration table A2 for a regeneration. Based on the electric motor speed ωm and the given target torque T*m, the table to be used is determined as to whether the situation is a power run or a regeneration.

[0049] Note that as the rotational speed ωm of the electric motor 10m increases, the back electromotive forces generated in the stator coils 11 to 13 increase, leading to higher terminal voltages in the coils 11 to 13. Consequently, it becomes difficult to deliver the target current from the inverter 19m to the coils 11 to 13, resulting in the failure to achieve the target output torque. In this case, the target torque T*m can be achieved by reducing the d-axis current id and the q-axis current iq by Δid and Δiq, respectively, along the constant torque curve for the given motor target torque T*m, although this reduces power utilization efficiency. This is known as field weakening control. The d-axis field weakening current Δid is generated by a field setting parameter calculation to be used for calculating a d-axis current command and a q-axis current command.The d-axis weakening magnetic field current Δid is calculated by a field weakening current calculation 41. Details of this will be described later.

[0050] In the calculation of the d-axis current command in the “output calculation” 35, the microcomputer MPA calculates a d-axis target current id* by subtracting the d-axis field weakening current Δid from the d-axis current value id, which is read from the first high-efficiency torque curve table A, according to the target torque T*m, which is determined by the torque command limitation. id*=−id−Δid

[0051] In the calculation of the q-axis current command, a second high-efficiency torque curve table B is used in the output calculation 35. The second high-efficiency torque curve B, which represents the q-axis value of the high-efficiency torque curve, is corrected into a curve representing a q-axis target current. This target current is obtained by subtracting the q-axis field weakening current Δiq, which is paired with the d-axis field weakening current Δid. The data of the second high-efficiency torque curve B are stored in the second high-efficiency torque curve table B after the correction.

[0052] The second high-efficiency torque curve table B is a storage area in which the d-axis target current, corresponding to the target torque T*m, and the d-axis field weakening current Δid for generating the target torque with minimal power consumption—i.e., the target current on the second high-efficiency torque curve B after correction—have been inscribed. It also consists of a pair of tables: a power-run table B1 for power operation and a regeneration table B2 for regeneration. Based on the electric motor speed ωm and the target torque T*m, the table to be used is determined as to whether power operation or regeneration is required.

[0053] When calculating the q-axis current command, the q-axis target current iq*, which corresponds to the target torque T*m, and the d-axis field weakening current Δid are read from the second high-efficiency torque curve table B and used as the q-axis current command.

[0054] In the initial calculation 35, the microcomputer of the motor control device 30m calculates a current deviation δid between the d-axis target current id* and the d-axis current value id, and a current deviation δiq between the q-axis target current iq* and the q-axis current value iq, and then performs proportional and integral control based on the current deviations δid and δiq (PI calculation of the feedback control). Thus, the microcomputer of the motor control device 30m calculates a voltage reduction Vzdp, which represents a voltage command value, for a proportional component, and a voltage reduction Vzdi, which represents a voltage command value, for an integral component, based on the current deviation δid, and calculates a voltage reduction Vzd by summing the voltage reductions Vzdp and Vzdi as follows. Vzd=Vzdp+Vzdi

[0055] Furthermore, the output calculation 35 reads the rotational speed ω and the q-axis current iq, and calculates, based on the rotational speed ω, the q-axis current iq and the q-axis inductance Lq, an induced voltage ed induced by the q-axis current iq, as follows. ed=ωm⋅Lq⋅iq

[0056] By subtracting the induced voltage ed from the voltage reduction Vzd, the output calculation 35 also calculates a d-axis voltage command value vd*, which serves as an output voltage, as follows: vd*=Vzd−ed=Vzd−ωm⋅Lq⋅iq

[0057] The initial calculation 35 also calculates a voltage reduction Vzqp, which represents a voltage command value for a proportional component, and a voltage reduction Vzqi, which represents a voltage command value for an integral component, based on the current deviation δig and calculates a voltage reduction Vzq by summing the voltage reductions Vzqp and Vzqi as follows. Vzq=Vzqp+Vzqi

[0058] Based on the rotational speed ω, a counter-electromotive force constant MIf, the d-axis current id and the inductance Ld on the d-axis, the output calculation 35 further calculates an induced voltage eq that is induced by the d-axis current id, as follows. eq=ωm(MIf+Ld⋅id)

[0059] By adding the induced voltage eq to the voltage reduction Vzq, the output calculation 35 also calculates a q-axis voltage command value vq*, which serves as the output voltage, as follows. vq*=Vzq+eq=Vzq+ωm(MIf+Ld⋅id)

[0060] Next, a two-phase / three-phase converter 36, acting as a rotation / fixed coordinate transformation, converts target voltages vd* and vq* in the rotational coordinate system into target voltages VU*, VV*, and VW* for the respective phases in the fixed coordinate system, according to the two-phase / three-phase conversion. The target voltages VU*, VV*, and VW* for the respective phases are sent to a PWM pulse generator 50 by a modulation 37 when the voltage control mode is three-phase modulation mode. When the voltage control mode is three-phase modulation mode, the target voltages VU*, VV*, and VW* for the respective phases are converted in three-phase modulation mode into two-phase modulation voltages by a two-phase modulation 38 of the modulation 37 and sent to the PWM pulse generator 50.If the voltage mode is a 1-pulse mode in which all phases are powered by square waves, target voltages VU*, VV* and VW* for respective phases in the three-phase modulation mode are converted into phase voltages powered by square waves by a 1-pulse conversion in the modulation 37, and provided to the PWM pulse generator 50.

[0061] The PWM pulse generator 50, to which the three-phase target voltages VU*, VV* and VW* are supplied, converts these into PWM pulses MUm, MVm and MWm respectively for output of their voltage values. These pulses have a frequency (carrier frequency) synchronized with a low-frequency (5 kHz) or high-frequency (7.5 kHz) clock generated by a carrier clock generator 47, and provides the PWM pulses MUm, MVm and MWm as output signals to the driver circuit 20m, as shown in Fig. As shown in Figure 1, the driver circuit 20m generates six sequences of driver signals in parallel based on the PWM pulses MUm, MVm, and MWm, and switches transistors Tr1 to Tr6 of the voltage-type inverter 19m on and off using the driver signals of the respective sequences. As a result, VU*, VV*, and VW* are each applied to the stator coils 11 to 13 of the electric motor 10m, causing the phase currents iUm, iVm, and iWm to flow. The PWM pulse generator, provided with the target voltages for each phase in two-phase modulation mode, generates two phases of PWM pulses and the remaining one phase of an on or off signal (constant voltage output signal). If provided with target voltages for each phase in single-pulse modulation mode, the PWM pulse generator outputs energizing interval signals that supply energy to the respective phases with square waves.

[0062] The field weakening current calculation 41 calculates a voltage saturation indicator m, which is a parameter for field weakening control. Based on the d-axis voltage command value vd* and the q-axis voltage command value vq*, the field weakening current calculation 41 calculates a voltage saturation assessment indicator mi as a value representing a degree of voltage saturation, as follows. mi=√(vd*2+vq*2) / Vuc

[0063] Then, the field weakening current calculation 41 subtracts a constant kv from the voltage saturation assessment indicator mi to obtain a calculated voltage saturation value ΔV, where, if a threshold value representing the maximum output voltage of the inverter 19m is referred to as a reference value Vmax, the constant kv is given by the following equation. Vmax=kv⋅Vuc

[0064] The calculated voltage saturation value ΔV is given as follows. ΔV=mi−kv

[0065] Then the field weakening current calculation 41 calculates a field setting parameter.

[0066] When calculating the field setting value, ΔV is summed to obtain a sum ΣΔV. If the sum ΣΔV has a positive value, it is multiplied by a proportionality constant to calculate the d-axis field weakening current Δid as a positive setting value for implementing field weakening control. If the calculated voltage saturation value ΔV or the sum ΣΔV has a negative value, the setting value Δid and the sum ΣΔV are set to zero. The setting value Δid is used in calculating the d-axis current command and in calculating the q-axis current command.

[0067] The “two-phase / three-phase conversion” 36 calculates a target electric motor voltage Vm* in the two-phase / three-phase conversion process. The calculation formula is Vm* = √(Vd* 2 + Vq* 2 ). From the electric motor target voltage Vm* and the voltage Vuc of the secondary-side capacitor 23 (voltage value detected by the voltage sensor 24), a modulation rate circuit 43 in a modulation controller 42 calculates a modulation rate Mi as follows. Mi=Vm* / Vuc

[0068] A carrier frequency and modulation mode determiner 44 determines the carrier frequency and modulation mode based on the target torque T* of the electric motor 10m, the rotational speed ω, and the modulation rate Mi. The carrier frequency and modulation mode determiner 44 instructs the carrier clock generator 47 to generate an output signal of the determined carrier frequency and, depending on the determined modulation mode, instructs a selection 40 in the modulation 37 to generate a target voltage output signal for the modulation mode. The carrier frequency and modulation mode determiner 44 also provides the carrier frequency and modulation mode to a torque error correction 46.

[0069] During a first transition, in which the carrier frequency is switched from a low frequency fc to a high frequency k·fc, and the voltage control mode is switched from the three-phase modulation mode to the two-phase modulation mode, the torque error correction 46 reads a first torque correction value, which is used to reduce a torque step between before and after the transition, and which is associated with the current target torque T* and the speed ω of the electric motor 10m, from a first transition lookup table (for Dpwm, k·fc), and adds the first torque correction value to the torque command value TM*m at the summing point 33.During a second switchover in the opposite direction to the one above (switching from k·fc and two-phase modulation to fc and three-phase modulation), the torque error correction 46 reads a second torque correction value, used to reduce the torque step between before and after the switchover, from a lookup table for the second switchover (for SVpwm, fc), which is associated with the current target torque T* and the speed ω of the electric motor 10m, and adds the second torque correction value to the torque command value TM*m at summation point 33. After the correction described above has been performed, the process is repeated from the torque command limiter 34 to the two-phase / three-phase conversion 36, and then the modulation 37 outputs the target voltages for the respective phases in the modulation mode determined by the carrier frequency and modulation mode determination 44.Note that the switching of the target voltages for each phase is suspended from the time when the first change or the second change has occurred until the recalculation process from the torque command limitation 34 to the two-phase / three-phase conversion 36 is completed.

[0070] The MPU microcomputer, as in Fig. Figure 2 shows that the system is not only equipped with a CPU, but also with RAM, ROM, or flash memory for storing data and various programs. The programs, reference data, and lookup tables stored in the ROM or flash memory are written to the RAM, and based on these programs, input processing, computation, and output processing are performed in the block enclosed by the double-dashed line.

[0071] Fig. Figure 3 shows a simplified representation of a motor drive control (MDC) system, which is executed by the microprocessor (MPU) based on the following programs. When an operating voltage is applied, the MPU initializes itself, the PWM pulse generator (50), the carrier clock generator (47), and the driver circuit (20m), and sets the inverter (19m) to the stop and standby state to drive the electric motor (10m). The MPU then waits for a motor drive start command from a main controller of the vehicle drive control system (not shown). When the motor drive start command is given, the MPU sets initial values ​​for the electric motor control in internal registers to "Processing Start" (Step 1) and reads input signals or data into "Read Input" (Step 2).The microcomputer MPU converts to digital and reads the first target torque TM*m, which is given by the main control, the respective phase currents iU, iV and iW, which are detected by the current sensors 14m to 16m, the rotation angle signal SGθm of the rotary encoder 17m, and the voltages Vdc and Vuc, which are detected by the voltage sensors 21 and 24.

[0072] Note that only the step number is shown in parentheses, with the word "step" omitted from the following description.

[0073] Next, the MPU microcomputer calculates the rotation angle θ and the rotational speed ω based on the rotation angle signal SGθ (rotation angle data SGθ) that has been read (3). This function is in Fig. 2 shown as an angle and velocity calculation 32. Next, the microcomputer MPU reads the limit torque TM*max from the limit torque table, which corresponds to the read motor target torque TM*, the read DC voltage Vuc, and the calculated speed ω, and determines TM*max as the target torque T* if the read motor target torque TM* exceeds TM*max. If the motor target torque TM* is equal to or less than TM*max, the read motor target torque TM* is determined as the target torque T* (4). This function is in Fig. 2 shown as torque command limit 34. Next, the microcomputer MPU converts the detected three-phase currents iU, iV, and iW, which have been read, into the two-phase d-axis current value id and the q-axis current value using the three-phase / two-phase conversion (5). This function is in Fig. 2 as current feedback 31. Next, the microcomputer MPU calculates the d-axis field weakening current Δid to perform the d-axis field weakening control (6). This function is shown in Fig. 2 as the field weakening current calculation 41 shown.

[0074] The contents of “Output calculation” (7) are the same as the contents of the output calculation 35 described above, as in Fig. Figure 2 shows the voltage target values ​​Vd* and Vq* on the d- and q-axes, which were calculated in "Output Calculation" (7), are converted into target voltages VU*, VV*, and VW* for the respective phases in the three-phase modulation mode (8). At this point, the electric motor target voltage Vm* is also calculated. A subsequent step, "Control Modulation" (9), calculates the modulation rate Mi and determines the carrier frequency and the modulation mode based on the modulation rate Mi, the target torque T*, and the rotational speed ω.

[0075] Fig. Figure 4 shows details of “controlling the modulation” (9). This step calculates the modulation rate Mi = Vm* / Vuc (21) and determines the carrier frequency and modulation mode based on the target torque T* and the rotational speed ω of the electric motor 10m, and the modulation rate Mi (22). Details of the “determination of carrier frequency and modulation mode” (22) are given later with reference to the Fig. 6, Fig. 7, Fig. 8 to Fig. 9 described. If the specified carrier frequency differs from the carrier frequency currently being output, such that the frequency must be switched from the low frequency fc (5 kHz) to the high frequency k·fc (7.5 kHz, i.e., k = 1.5 in the present embodiment), the correction value used to reduce the output torque step caused by the switching, and assigned to the current target torque and speed, is read from the two-phase modulation lookup table, high frequency (for Dpwm, k·fc) (24). Then the torque command value TM* is corrected by the magnitude of the correction value that has been read (26), and based on the corrected torque command value, steps 6 to 8 are repeated to recalculate the target voltages (instantaneous voltages) VU*, VV*, and VW* for the respective phases in the three-phase modulation mode (27).If the specified carrier frequency differs from the carrier frequency currently being output, such that the frequency is to be switched from the high frequency k·fc (7.5 kHz) to the low frequency fc (5 kHz), the correction value used to reduce the output torque step caused by the switching, and assigned to the current target torque and speed, is read from the three-phase modulation lookup table, low frequency (for SVpwm, fc) (25). Then the torque command value TM* is corrected by the magnitude of the correction value that has been read (26), and based on the corrected torque command value, steps 6 to 8 are repeated to recalculate the target voltages VU*, VV*, and VW* for the respective phases in the three-phase modulation mode (27).

[0076] Reference is made again to Fig. 3. The next step, "Output Update" (10), sends the target voltages for each phase in the modulation mode determined in the modulation control (9) as an output signal to the PWM pulse generator 50 and instructs the carrier clock generator 47 to generate an output signal at the specified carrier frequency. Next, after a period of waiting for the timing for the next iteration (11), the process proceeds to perform the "Read Input" (2) step again. Then, the "Read Input" step (2) described above and the subsequent processes are executed. If a stop command is issued by a control unit while waiting for the timing for the next iteration, the microprocessor MPU stops the output for energizing the motor rotation (13) at that time.

[0077] It will be on Fig. 5 Reference is made. In “Determining the carrier frequency and modulation mode” (22), if the modulation mode in question is SVpwm (three-phase modulation mode, low frequency fc), a lookup is performed to determine whether the target torque and rotational speed are in A ( Fig. 7, Fig. 8 to Fig. 9) in a first region (high-frequency switching region), in which the carrier frequency must be the high frequency k·fc (42). If the target torque and speed in A are in the first region, the carrier frequency is determined to be the high frequency k·fc and, accordingly, the voltage control mode is switched from three-phase modulation (SVpwm) to two-phase modulation mode (43). If the target torque and speed are outside of A, it is checked whether a condition is met for switching from three-phase modulation to two-phase modulation while maintaining the low frequency fc (44). If the condition is met, the modulation mode is switched from three-phase modulation to two-phase modulation (Dpwm) while maintaining the low frequency fc (45).

[0078] If the current modulation mode is Dpwm (two-phase modulation mode, low frequency fc), a lookup is performed to determine whether the target torque and speed are in B ( Fig. 7, Fig. 8 to Fig. 9) in the first region, in which the carrier frequency is to be the high frequency k·fc (46). If the target torque and speed are in B, the carrier frequency is determined to be the high frequency k·fc (47). The voltage control mode continues as the two-phase modulation mode. If the target torque and speed are outside of B, a lookup is performed to determine if a condition is met for switching from two-phase modulation to three-phase modulation (SVpwm) while maintaining the low frequency fc (48). If the condition is met, the modulation mode is switched from two-phase modulation (Dpwm) to three-phase modulation (SVpwm) while maintaining the low frequency fc (49).If the condition for switching to three-phase modulation (SVpwm) is not met, a lookup is performed to determine if a condition is met to make the voltage control mode the 1-pulse mode (50). If the condition is met, the voltage control mode is switched to the 1-pulse mode (51).

[0079] Next, we will also be looking at Fig. 6. If the current modulation mode is that of A or B (two-phase modulation mode, high frequency k·fc), a lookup is performed to determine if a condition is met to switch the voltage control mode to the 1-pulse mode (52). If the condition is met, the voltage control mode is switched to the 1-pulse mode (53). If the condition for the 1-pulse mode is not met, a lookup is performed to determine if the target torque and speed are outside a second range (range for switching to the low frequency, i.e., range for aborting the high frequency), which is indicated by the dashed line outside the first range, as shown in the Fig. 7, Fig. 8 to Fig. 9 shown (54). If the target torque and speed are outside the second range, the high frequency must be aborted. Consequently, a lookup is performed to determine whether the target torque and speed are within the three-phase modulation range (55), and if the result is confirmation, the three-phase modulation mode and the low frequency fc (SVpwm) are set (56). If the target torque and speed are not within the three-phase modulation range, the two-phase modulation mode and the low frequency (Dpwm) are set (57).

[0080] If the current modulation mode is the 1-pulse mode (58), a lookup is performed to determine if a condition is met to make the carrier frequency the high frequency (59). If the condition is met, the carrier frequency is determined to be the high frequency k·fc, and in conjunction with this, the two-phase modulation mode (A or B) is set as the voltage control mode (60). If the high frequency condition is not met, a lookup is performed to determine if a condition is met to make the modulation mode and carrier frequency the two-phase modulation mode and the low frequency fc (Dpwm) (61). If the condition is met, the two-phase modulation and the low frequency fc (Dpwm) are set (62).

[0081] The regional sections “SVpwm” (three-phase modulation, low frequency fc), “A, B” (two-phase modulation, high frequency k·fc), “Dpwm” (two-phase modulation, low frequency fc) and “1 Pulse” (square wave energization of all phases), as described above, are in Fig. 7, Fig. 8 and Fig. Figure 9 shows the cases where the secondary-side voltage Vuc of the converter 40, detected by the voltage sensor 25, is 220 V, 300 V, and 400 V, respectively. Region "A" is a region where three-phase modulation is used when the carrier frequency is held at the same frequency (fc), but two-phase modulation is used to reduce switching losses caused by switching to the high frequency in conjunction with switching the carrier frequency from the low frequency fc to the high frequency k·fc for the purpose of reducing sideband noise. Region "B" is a region where two-phase modulation is used to reduce switching losses, and where the carrier frequency is switched from the low frequency fc to the high frequency k·fc to reduce sideband noise.

[0082] The MPU microcomputer's memory stores the reference values, as shown in Fig. 5 and Fig. Figure 6 shows the reference values ​​(thresholds) for low / high switching and high / low switching of the carrier frequency, and the reference values ​​for modulation switching, in each division of the range of the secondary-side voltage Vuc, to form a reference data table (LUT, i.e., a lookup table), and the microcomputer MPU uses the above reference values ​​by reading them from the LUT according to the Vuc values ​​at the time when the “determination of the carrier frequency and modulation mode” is performed (22), as shown in the Fig. 5 and Fig. 6 shown.

[0083] The value of k of the high frequency k·fc, which is used to suppress sideband noise in regions A and B, is preferably such that 1 <k<2 ist. Wenn k≤1 ist, erhält man keine Wirkung zur Unterdrückung des Seitenbandrauschens, und wenn k> If k = 2, the switching loss is large. The present embodiment uses a value of k = 1.5 to obtain a high frequency of 7.5 kHz.

[0084] If the high frequency is maintained by setting k = 1.5, in regions A and B where the high frequency k·fc is used to suppress sideband noise, the sum of the power losses of the electric motor is 10 mΩ and the inverter is 19 mΩ (similar to 10 g and 19 g), so the total loss is approximately equal to that of the reference example, which does not use a high frequency but uses the low frequency fc (5 kHz) entirely, as in Fig. Figure 10 shows that in region "A", although the iron loss (core loss) of the electric motor 10m increases at a comparatively low speed due to the frequency being switched to the high frequency k·fc, the switching loss of the inverter 19m is reduced by the change from three-phase to two-phase modulation. Consequently, the overall loss is not significantly increased. In region "B", the carrier frequency is switched to the high frequency while the two-phase modulation is maintained. Consequently, although the switching loss of the inverter 19m increases, the overall loss does not increase significantly because the iron loss is reduced.

[0085] As described above, the overall loss is not significantly increased according to the present invention. Since the high frequency k·fc is used in both region A and region B, sideband noise is suppressed. Therefore, the present invention can suppress sideband noise without significantly increasing the power loss of the electric motor drive.

[0086] The control function of the motor control device 30m for controlling the operation of the electric motor 10m for rotating the wheels has been described above.

[0087] Reference is made again to Fig.1. The electric motor 10g, which is driven by a machine in the vehicle, can be described as a generator or a motor-generator. In the present embodiment, the electric motor 10g serves as an electric motor (power run) for driving the engine for starting when the engine is started, and as a generator (regeneration) for generating electrical power by being driven by the engine after starting. The function and operation of the motor control device 30g for controlling the electric motor 10g are the same as those of the motor control device 30m. Furthermore, the structure and function of the inverter 19g for supplying power to the electric motor 10g are the same as those of the inverter 19m. The structure and function of the motor control device 30g are the same as those of the motor control device 30m.

[0088] When the motor is started, a target torque TM*g of a positive value is provided by a main controller (not shown) of the motor control device 30g, and the motor control device 30g performs a control operation similar to the control operation of the motor control device 30m described above. After the motor has started and its output torque has increased, the main controller switches the target torque TM*g to a negative value for generation (regeneration). The motor control device 30g then controls the inverter 19g so that the output torque of the electric motor 10g matches the target torque (target load of the motor) of the negative value. The contents of this calculation (output control calculation) are also similar to the output control calculation of the motor control device 30m described above.

[0089] As described above, the secondary-side voltage Vuc (the voltage of the secondary-side capacitor 23), which serves as an output voltage of the converter 40, is used to calculate the torque command limit in the motor control devices 30m and 30g, and likewise to calculate the field weakening currents Δid and Δiq. Within the maximum value of the secondary-side voltage achievable with the power capacity of the primary-side DC power source 18, 22, the secondary-side voltage Vuc is preferably set to correspond to the target torques TM*m, TM*g and rotational speeds, i.e., such that it is higher the higher the target torques and higher the rotational speeds. A converter control device 30v sets the secondary-side voltage Vuc.

[0090] In the present embodiment, the converter control device 30v is also an electronic control device, primarily formed by a microcomputer, and includes the microcomputer, an interface (signal processing circuit) not shown, and a PWM pulse generator. The converter control circuit 30v further includes an interface (communication circuit) not shown between the microcomputer and the main control unit of the vehicle drive control system (not shown) on the vehicle.

[0091] The converter control device 30v reads a secondary-side target voltage Vuc*m (first secondary-side target voltage Vuc*m) provided by the motor control device 30m, and a secondary-side target voltage Vuc*g (second secondary-side target voltage Vuc*g) provided by the motor control device 30g, and converts digitally and reads the battery voltage Vdc and the secondary-side voltage Vuc detected by sensors 21 and 24, respectively.Next, the higher voltage from the first secondary-side target voltage Vuc*m or the second secondary-side target voltage Vuc*g is determined as a target voltage Vuc*, and a PWM signal Pvf for switching element 42 on / off for an upward voltage step and a PWM signal Pvr for switching element 43 on / off for a downward voltage step are generated and provided to a 20V driver circuit so that the voltage Vuc detected by the voltage sensor 24 matches the target voltage Vuc*. The 20V driver circuit switches the semiconductor switches 42 and 43 on and off based on the PWM signals Pvf and Pvr.The on and off of switching element 42 for an upward voltage step of converter 40 is PWM-controlled when the voltage needs to step upwards, whereas the on and off of switching element 43 for the regeneration of converter 40 is PWM-controlled when the voltage needs to step downwards. The semiconductor switch 42 for an upward voltage step and the semiconductor switch 43 for regeneration are switched in a complementary manner, such that the latter is kept off while the former is on, and the latter is kept on while the former is off.

Claims

[1] Electric motor control device with a DC power source (18); an inverter (19m, 19g) connected between an electric motor (10m, 10g) and the DC power source (18) to control an exchange of electrical power between them; an inverter control device (20m, 20g) that generates PWM pulses which have a duty cycle corresponding to voltage command signals, and that sends the PWM pulses to the inverter (19m, 19g) to switch the inverter; a frequency-changing means (46) that modifies a carrier frequency of the PWM pulses generated by the inverter control means in a manner corresponding to a frequency control signal; and a motor control means (30m, 30g) that determines whether a target torque and speed of the electric motor are within a predetermined range, which is a range for suppressing sideband noise, when a current modulation mode is a three-phase modulation mode in which three-phase voltages are controlled by PWM and a current carrier frequency is a low frequency, which, when the target torque and speed of the electric motor are within the predetermined range, as a result of the determination, provides the frequency control means with the frequency control signal to set the carrier frequency to a high frequency that is higher than the low frequency, and in accordance with this, switches the voltage control mode from the three-phase modulation mode to a two-phase modulation mode in which two phases are controlled by PWM, while the application of PWM control to the other phase is stopped, which,If the target torque and speed are outside the predetermined range, as a result of the determination, the carrier frequency is maintained as the low frequency, which determines whether a condition is met to switch from three-phase modulation to two-phase modulation, which, if the condition is met, switches the voltage control mode from three-phase modulation to two-phase modulation while maintaining the low frequency, and which provides voltage command signals to the inverter control means to bring an output torque of the electric motor into conformity with the target torque, wherein , During a first switch, in which the carrier frequency is switched from low frequency to high frequency, and the voltage control mode is also switched from three-phase modulation to two-phase modulation, the motor control device (30m, 30g) corrects the target torque by adding a first torque correction value to the target torque to reduce the torque step between before and after the switch. During a second switch, which is a switch in the opposite manner to the first, the motor control device corrects the target torque by adding a second torque correction value to the target torque to reduce the torque step between before and after the switch. The motor control device also provides voltage command signals to the inverter control device to adjust the output torque of the electric motor (10m,10g) to bring into accordance with the corrected target torque. [2] Electric motor control device according to claim 1, wherein, when the voltage control mode is the two-phase modulation mode, the motor control means (30m, 30g) continues the two-phase modulation mode even after switching to the high frequency when the carrier frequency is switched from the low frequency to the high frequency. [3] Electric motor control device according to claim 1, wherein, where the low frequency is designated as fc, the high frequency is equal to k·fc, wherein 1 <k<2, ist. [4] Electric motor control device according to claim 1, wherein the motor control means (30m, 30g) determines the voltage control mode based on a modulation rate which is a ratio of an electric motor target voltage to a DC voltage supplied to the inverter by the DC power source, and based on the speed of the electric motor, and widens a range for the modulation rate and speed in which the two-phase modulation mode is performed by changing the modulation mode from the three-phase modulation mode to the two-phase modulation mode when the carrier frequency is the high frequency in the three-phase modulation mode, or when the carrier frequency is the high frequency and the voltage control mode is determined as three-phase modulation mode. [5] Electric motor control device according to claim 1, wherein the predetermined area includes a first area and a second area which contains the first area and is wider than the first area, and wherein the motor control means switches the carrier frequency at the low frequency to the high frequency when the target torque and speed of the electric motor (10m, 10g) are in the first area, and switches the carrier frequency at the high frequency to the low frequency when the target torque and speed of the electric motor are outside the second area. [6] Electric motor control device according to claim 1, wherein the high frequency is a frequency for reducing sideband noise. [7] Electric motor control device according to claim 1, wherein the low frequency is a frequency for reducing switching losses of the inverter, [8] Drive unit with the electric motor control device according to any one of claims 1 to 7; and an electric motor which is supplied with power by the inverter (19m, 19g) contained in the electric motor control device and drives wheels. [9] Hybrid drive unit with a DC power source (18); a first electric motor (10m) that drives wheels; a second electric motor (10g) which is driven by a fuel engine; a first inverter (19m) which is arranged between the first electric motor and the DC power source to control an exchange of electrical power between them; a second inverter (19g) arranged between the second electric motor and the DC power source to control an exchange of electrical power between them; a first inverter control device (20m) that generates first PWM pulses which have a duty cycle corresponding to the first voltage command signals, and sends the first PWM signals to the first inverter to switch the first inverter; a second inverter control device (20g) that generates second PWM pulses which have a duty cycle corresponding to the second voltage command signals, and sends the second PWM pulses to the second inverter to switch the second inverter; a first frequency-changing means (46) that modifies a first carrier frequency of the first PWM pulses generated by the first inverter control means in a manner corresponding to a first frequency control signal; a second frequency-changing means (46) that modifies a second carrier frequency of the second PWM pulses generated by the second inverter control means (20g) in a manner corresponding to a second frequency control signal; a first motor control means (30m) that determines whether a target torque and speed of the first electric motor are within a predetermined range, which is a range for suppressing sideband noise, when a current modulation mode is a three-phase modulation mode in which three-phase voltages are controlled by PWM and a current carrier frequency is a low frequency, which, when the target torque and speed of the first electric motor (10m) are within the first predetermined range, as a result of the determination, provides the frequency-changing means with the first frequency control signal to set the first carrier frequency to a high frequency that is higher than the low frequency, and in accordance with this, switches the voltage control mode from the three-phase modulation mode to a two-phase modulation mode in which two phases are controlled by PWM,while the application of PWM switching to the other phase is stopped, which, if the target torque and speed are outside the first predetermined range, as a result of the determination, maintains the first carrier frequency as the low frequency, which determines whether a condition is met to switch from three-phase modulation to two-phase modulation, which, if the condition is met, switches the voltage control mode from three-phase modulation to two-phase modulation while maintaining the low frequency, and which provides first voltage command signals to the first inverter control means to bring an output torque of the first electric motor into conformity with the target torque; and , a second motor control means (30g) that determines whether a target torque and speed of the second electric motor (10g) are within a second predetermined range, which is a range for suppressing sideband noise, when a current modulation mode is a three-phase modulation mode in which three-phase voltages are controlled by PWM and a current carrier frequency is a low frequency, which, when the target torque and speed of the second electric motor (10g) are within the second predetermined range as a result of the determination, provides the frequency-changing means with the second frequency control signal to set the second carrier frequency to a high frequency higher than the low frequency, and accordingly switches a voltage control mode from the three-phase modulation mode to a two-phase modulation mode in which two phases are PWM-controlled,While the application of PWM switching to the other phase is stopped, if the target torque and speed are outside the second predetermined range, as a result of the determination, the second carrier frequency is maintained as the low frequency, which determines whether a condition is met to switch from three-phase modulation to two-phase modulation, which, if the condition is met, switches the voltage control mode from three-phase modulation to two-phase modulation while maintaining the low frequency, and which provides second voltage command signals to the second inverter control means to match an output torque of the second electric motor to the target torque, wherein, During a first switch, in which the carrier frequency is switched from low frequency to high frequency, and the voltage control mode is also switched from three-phase modulation to two-phase modulation, the first and second motor control means (30m, 30g) correct the target torque by adding a first torque correction value to the target torque to reduce the torque step between before and after the switch, whereas during a second switch, which is a switch in the opposite manner to the first switch, the first and second motor control means correct the target torque by adding a second torque correction value to the target torque to reduce the torque step between before and after the switch, and the first and second motor control means provide voltage command signals to the first and second inverter control means.to bring the output torque of the first and second electric motors (10m, 10g) into line with the corrected target torque.

Images