Frequency conversion and phase shift dual-mode switching control strategy
The frequency conversion and phase shift dual-mode switching control strategy addresses inefficiencies in AC/DC converters by optimizing soft switching and reducing reverse currents, enhancing efficiency and reducing converter size and cost in hybrid microgrids.
Patent Information
- Application Number
- CN202310706566.5
- Authority / Receiving Office
- CN · China
- Patent Type
- Applications(China)
- Current Assignee / Owner
- Filing Date
- 2023-06-14
- Publication Date
- 2024-12-17
AI Technical Summary
Traditional AC/DC converters in hybrid microgrids face issues such as low efficiency, high cost, low power density, and high turn-off losses due to reverse currents, especially in single-stage converters, and they require complex two-stage conversions with electrolytic capacitors that reduce reliability.
A frequency conversion and phase shift dual-mode switching control strategy that optimizes soft switching by varying the phase shift and frequency modulation based on real-time power conditions, using a single-stage dual active bridge converter to minimize reverse currents and improve efficiency.
The strategy enhances efficiency, reduces converter size and cost, and minimizes turn-off losses by optimizing soft switching across full voltage and load ranges, ensuring zero voltage turn-on and reducing reverse currents.
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Figure CN119154629A_ABST
Abstract
Description
Technical Field
[0001] The present invention relates to the technical field of electric energy conversion, and in particular to a frequency conversion and phase shift dual-mode switching control strategy. Background Art
[0002] The AC / DC hybrid microgrid integrates the characteristics of a single AC microgrid and a DC microgrid, and has the advantages of flexible power supply and low system loss. Among them, the AC / DC converter between the AC subgrid and the DC subgrid serves as a bridge connecting the two networks, realizing energy conversion and support, and is responsible for regulating the energy balance of the entire microgrid. It is one of the key technologies of the AC / DC hybrid microgrid.
[0003] The traditional two-stage AC / DC topology can provide inherent bidirectional power transmission capability, electrical isolation, and a simple method to achieve soft switching. However, in order to maintain the stability of DC voltage and instantaneous power flow, a huge electrolytic capacitor is inevitably required, which reduces the reliability and life of the two-stage AC / DC converter. In addition, its energy transfer requires two-stage conversion, and it also has many disadvantages such as low overall efficiency, high cost, and low power density. Due to these shortcomings of the two-stage converter, single-stage isolated AC / DC converters with higher power density and higher reliability have been widely studied by scholars in recent years.
[0004] A common single-stage isolated AC / DC converter is Figure 1 As shown in the figure, a hybrid modulation strategy combining phase shift and frequency modulation is often used. By adjusting the phase shift angle, the current at the time when all switches are turned on is set to a constant reset value, creating the conditions for zero voltage turn-on of all switches, and achieving zero voltage turn-on of switches within the full input range, which greatly improves the efficiency of the converter. At the same time, since the input current needs to be sinusoidal, the PFC function is achieved by frequency modulation. However, since the current on the AC side is limited to a fixed reset value at a certain moment, a reverse current will be generated on the secondary output side. At the same time, due to the existence of the transformer ratio, the actual current reset value after the smaller reverse reset current is folded back to the secondary side will be very large, resulting in a higher reverse current on the secondary output, and then a higher turn-off loss of the secondary switch.
[0005] Another single-phase single-stage AC / DC converter based on a semi-dual active bridge (semi-DAB) DC / DC converter is shown in Figure 2, the switching moment between the dual phase shift (DPS) and single phase shift (SPS) modes is selected according to the instantaneous value of the input side AC current, using only two control variables, namely the primary side phase shift and the phase shift between the two bridges. This modulation strategy enables the converter to achieve power factor correction without an internal current tracking loop. In addition, the modulation strategy of multi-mode switching increases the operating range of the zero voltage switch. However, under light load conditions, some switches will lose soft switching, there will be harmonic distortion near the voltage zero crossing point, and only one-way transmission is possible. Summary of the invention
[0006] The present invention aims to provide a frequency conversion and phase shift dual-mode switching control strategy.
[0007] To achieve the above object, the technical solution of the present invention is:
[0008] A variable frequency phase shift dual mode switching control strategy, comprising:
[0009] Step S1, calculating the switching frequency f under frequency conversion plus extended phase shift modulation;
[0010] Step S2, determining whether it is in rectification mode or inversion mode;
[0011] Step S3, in the rectification mode, the magnitude of the phase shift ratio D1 and the phase shift ratio D2 are calculated in real time and compared;
[0012] Step S4, if the phase shift ratio D1 is smaller than the phase shift ratio D2, then continue to use frequency conversion plus extended phase shift modulation;
[0013] Step S5, if the phase shift ratio D1 is not less than the phase shift ratio D2, switching to frequency conversion plus single phase shift modulation;
[0014] Step S6, in the inverter mode, calculate the shift ratio D2 in real time and compare it with 1;
[0015] Step S7, if the phase shift ratio D2 is less than 1, continue to use frequency conversion plus extended phase shift modulation;
[0016] Step S8: If the phase shift ratio D2 is not less than 1, switch to frequency conversion plus single phase shift modulation.
[0017] Under frequency conversion plus extended phase shift modulation, the expression of the phase shift ratio D1 and the phase shift ratio D2 is:
[0018]
[0019] Among them, the expressions of parameter A and parameter C are as follows:
[0020]
[0021] Among them, I B1 ,IB2 is the reset current, n is the transformer turns ratio, V ac is the AC voltage, V dc is the DC voltage, L is the phase-shift inductance, where the phase shift ratio is the proportion of the phase shift angle to half a cycle, and f is the switching frequency.
[0022] Under variable frequency plus extended phase shift modulation, the expression of the above switching frequency f is as follows:
[0023]
[0024] in,
[0025]
[0026] Among them, I ref is the current reference value.
[0027] Under variable frequency plus extended phase shift modulation, the average input current I ac It can be expressed as:
[0028]
[0029] Under frequency conversion plus single phase shift modulation, the expression of the phase shift ratio D is as follows:
[0030]
[0031] Among them, I B is the reset current, L is the phase shift inductor, V ac is the AC voltage, V dc is the DC voltage, n is the transformer turns ratio, and f is the switching frequency.
[0032] Under variable frequency plus single phase shift modulation, the expression of the above switching frequency f is as follows:
[0033]
[0034] In the above formula,
[0035]
[0036] Among them, I ref is the current reference value.
[0037] Under variable frequency plus single phase shift modulation, the average input current I ac It can be expressed as:
[0038]
[0039] In a specific embodiment, the step S2 further includes measuring power P, where if the power P is greater than 0, it is a rectification mode; if the power P is less than 0, it is an inverter mode.
[0040] The present invention also provides a single-stage dual active bridge converter, which applies the above-mentioned frequency conversion and phase shift dual-mode switching control strategy.
[0041] The present invention also provides a three-phase Swiss-type modular single-stage AC / DC converter, which applies the above-mentioned frequency conversion and phase shift dual-mode switching control strategy.
[0042] Beneficial effect: The variable frequency phase-shift dual-mode switching control strategy of the present invention optimizes the soft switching range, realizes zero voltage turn-on in the full voltage / full load range, and reduces the reverse current on the secondary side, thereby reducing the turn-off loss, thereby optimizing the design of the single-stage bidirectional AC / DC converter, maximizing efficiency, and reducing its size and cost.
[0043] In order to make the above features and advantages of the invention more obvious and easy to understand, embodiments are given below and described in detail with reference to the accompanying drawings. BRIEF DESCRIPTION OF THE DRAWINGS
[0044] Figure 1 The schematic diagram of a circuit of a single-stage isolated AC / DC converter is shown in FIG.
[0045] Figure 2 The schematic diagram of a circuit of a single-phase single-stage AC / DC converter based on a semi-dual active bridge DC / DC converter is shown.
[0046] Figure 3 It is a circuit diagram of a single-stage dual active bridge converter.
[0047] Figure 4 In rectification mode Figure 3 Schematic diagram of the switch drive signal and voltage and current waveforms of the circuit under the single-phase shift plus variable frequency modulation control strategy.
[0048] Figure 5 In rectification mode Figure 3 Schematic diagram of the output current waveform of the circuit in the middle under the single-phase-shifted additive variable frequency modulation control strategy.
[0049] Figure 6 In rectification mode Figure 3 Schematic diagram of the switch drive signal and voltage and current waveforms of the circuit under the extended phase-shift plus variable frequency modulation control strategy.
[0050] Figure 7 In rectification mode Figure 3 Schematic diagram of the output current waveform of the circuit in the middle under the extended phase-shift plus variable frequency modulation control strategy.
[0051] Figure 8 In rectification mode Figure 3Schematic diagram of the phase shift ratio of the circuit in the middle under the extended phase shift plus frequency conversion modulation control strategy.
[0052] Fig. 9 Inverter mode Figure 3 Schematic diagram of the phase shift ratio of the circuit in the middle under the extended phase shift plus frequency conversion modulation control strategy.
[0053] Fig.10 The present invention provides a flow chart of a frequency conversion and phase shift dual-mode switching control strategy.
[0054] Fig.11 for Figure 3 Medium Circuit Application Fig.10 The control strategy in the rectification mode outputs the current I o Waveform diagram of .
[0055] Fig.12 for Figure 3 Medium Circuit Application Fig.10 The control strategy outputs the current I in the inverter mode. o Waveform diagram of .
[0056] Fig.13 It is a circuit diagram of a three-phase Swiss-type modular single-stage AC / DC converter.
[0057] Fig.14 The voltage and current waveforms of a three-phase Swiss modular single-stage AC / DC converter.
[0058] Fig.15 This is the output current waveform of a three-phase Swiss modular single-stage AC / DC converter. DETAILED DESCRIPTION
[0059] In order to make the purpose and technical solution of the embodiment of the present invention clearer, the technical solution of the embodiment of the present invention will be clearly and completely described below in conjunction with the drawings of the embodiment of the present invention. Obviously, the described embodiment is a part of the embodiment of the present invention, not all of the embodiments. Based on the described embodiment of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.
[0060] The working principle of a frequency conversion and phase shift dual-mode switching control strategy of the present invention will be described below in a specific embodiment.
[0061] Figure 3 Figure 1 is a circuit diagram of a single-stage dual active bridge converter. Figure 3 As shown, switch S 31 With switch S 32 Connect in series in the same direction, switch S 33 With switch S 34Connect in series in the same direction, switch S 31 With switch S 32 The two ends of the series connection and the switch S 33 With switch S 34 The two ends of the series connection are connected in parallel, and the switch S 31 With switch S 32 The series midpoint connected inductor L 31 The first end of the inductor L 31 The second end is connected to the AC power supply u ac The first end, AC power supply u ac The second end of the switch S 33 With switch S 34 The series midpoint of the capacitor C 31 With capacitor C 32 Connect in series in the same direction, switch S 35 With switch S 36 Connected in series in the same direction, capacitor C 31 With capacitor C 32 The two ends of the series connection and the switch S 35 With switch S 36 The two ends of the series connection are connected in parallel, and the switch S 35 With switch S 36 The series midpoint connected inductor L 32 The first end of the inductor L 32 The second end of the transformer T31 is connected to the first end of the primary winding of the transformer T31. 31 The second end of the primary winding is connected to capacitor C 31 With capacitor C 32 The series midpoint of the switch Q 31 With switch Q 32 Connect in series in the same direction, switch Q 33 With switch Q 34 Connect in series in the same direction, switch Q 31 With switch Q 32 The two ends of the series connection and the switch Q 33 With switch Q 34 The two ends of the series connection are connected in parallel, and the switch Q 31 With switch Q 32 The series midpoint is connected to the first end of the secondary winding of the transformer T31, and the second end of the secondary winding of the transformer T31 is connected to the switch Q 33 With switch Q 34 The series midpoint, capacitor C 33 With switch Q 33 With switch Q 34 The two ends of the series connection are connected in parallel, and the DC power supply V dc Connected in parallel with capacitor C 33 Both ends.
[0062] The following describes the rectification mode Figure 3The working process of the circuit in the single phase shift plus variable frequency modulation control strategy, the switch drive signal and voltage and current waveforms are as follows Figure 4 As shown. In each cycle, the output current I o There will be some reverse current, such as Figure 5 As shown in Figure 2, the magnitude of the reverse current at time t1 and t2 is related to the absolute value of the inductor current at this time |I L | is proportional to the current. However, due to the fixed reset current I B The modulation measures are not used to meet the soft switching conditions, resulting in the current at time t1 not being limited, and due to the high transformation ratio of the transformer, the current value returned to the secondary side will be very large.
[0063] Under the control strategy of single phase shift plus variable frequency modulation, the average input current I ac It can be expressed as:
[0064]
[0065] Where n is the transformer turns ratio, f is the switching frequency, and L is the phase shift inductor. Figure 3 In the figure, L is the inductance L 32 .
[0066] The expression of the shift ratio D is as follows:
[0067]
[0068] Among them, I B is the reset current, V ac is an AC voltage, such as Figure 5 As shown in the figure, the phase shift ratio is the ratio of the phase shift angle to half a period, and its value range is 0-1.
[0069] The expression of switching frequency f is as follows:
[0070]
[0071] In the above formula,
[0072]
[0073] Among them, I ref is the current reference value. Figure 3 Topology in I ref is the alternating current I ac The absolute value of the magnitude.
[0074] Under the control strategy of single phase shift plus variable frequency modulation, Figure 3 The operation status of the intermediate circuit in the inverter mode is similar to that in the rectifier mode, which will not be described here.
[0075] The following describes the rectification mode Figure 3 The working process of the circuit in the extended phase shift plus frequency conversion modulation control strategy, the switch drive signal and voltage and current waveforms are as follows Figure 6 As shown. In each cycle, the output current I o There will be some reverse current, such as Figure 7 As shown, its magnitude is related to the inductor current I at this moment. L Proportional to. Similar to single-phase shift, the reset current I B1 and I B2 The reverse current that affects the secondary output current. In order to meet the soft switching conditions on the AC side and the soft switching at the maximum voltage, the reset current I B1 ,I B2 It is set to 4-5A, but due to the transformer ratio, the secondary reverse current will have a peak value of 16-20A, which will cause large turn-off losses.
[0076] Under the control strategy of extended phase shift plus variable frequency modulation, the average input current I ac It can be expressed as:
[0077]
[0078] Where n is the transformer turns ratio, f is the switching frequency, and L is the phase shift inductor. Figure 3 Where L is the inductance L 32 , the expression of the shift ratio D1 and the shift ratio D2 is
[0079]
[0080] Among them, the expressions of parameter A and parameter C are as follows:
[0081]
[0082] Among them, I B1 ,I B2 is the reset current, V ac is the AC voltage. The phase shift ratio is the ratio of the phase shift angle to half a period, and its value range is 0-1.
[0083] More specifically, the switching frequency f is expressed as follows:
[0084]
[0085] in,
[0086]
[0087] Among them, I ref is the reference current value.
[0088] Under the control strategy of extended phase shift plus variable frequency modulation, Figure 3 The operation status of the intermediate circuit in the inverter mode is similar to that in the rectifier mode, which will not be described here.
[0089] For the control strategy of extended phase shift plus frequency conversion modulation, such as Figure 7 As shown, the maximum reverse current on the secondary side is proportional to the inductor current I L It is related to the current peak at time t0, therefore, it is only necessary to change the current at time t0 to improve the reverse current situation of the secondary side.
[0090] In the rectification mode, due to the limitation of extended phase shift modulation, there are different power expressions when the phase shift ratio D2 is greater than the phase shift ratio D1 or the phase shift ratio D2 is less than the phase shift ratio D1. Figure 8 As shown in the figure, when reducing the reset current at time t0, the phase shift ratio D1 is greater than the phase shift ratio D2, which will cause the current to be distorted near the zero crossing point. Therefore, the single phase shift modulation strategy can be used for the partial moments when the phase shift ratio D2 is not greater than the phase shift ratio D1. Although the disadvantage of using single phase shift modulation in the full cycle is that the reset current is too large, near the zero crossing point, since the input voltage is very low during this period, only a very low reset current is required, and the disadvantage of single phase shift modulation is reduced.
[0091] In inverter mode, Fig. 9 As shown, the phase shift D2 value exceeds 1 (due to simulation limitations Fig. 9 The highest value is 1), so we can also consider using single phase shift modulation near the zero crossing point.
[0092] Therefore, the flow chart of a frequency conversion phase shift dual mode switching control strategy of the present invention is as follows: Fig.10 As shown, including,
[0093] Step S1, calculating the switching frequency f under frequency conversion plus extended phase shift modulation;
[0094] Step S2, determining whether the mode is rectification mode or inverter mode, more specifically, measuring power P, if power P is greater than 0, it is rectification mode; if power P is less than 0, it is inverter mode;
[0095] Step S3, in the rectification mode, the magnitude of the phase shift ratio D1 and the phase shift ratio D2 are calculated in real time and compared;
[0096] Step S4, if the phase shift ratio D1 is smaller than the phase shift ratio D2, then continue to use frequency conversion plus extended phase shift modulation;
[0097] Step S5, if the phase shift ratio D1 is not less than the phase shift ratio D2, switching to frequency conversion plus single phase shift modulation;
[0098] Step S6, in the inverter mode, calculate the shift ratio D2 in real time and compare it with 1;
[0099] Step S7, if the phase shift ratio D2 is less than 1, continue to use frequency conversion plus extended phase shift modulation;
[0100] Step S8: If the phase shift ratio D2 is not less than 1, switch to frequency conversion plus single phase shift modulation.
[0101] Furthermore, the phase shift ratio D or the phase shift ratio D1, the phase shift ratio D2, and the switching frequency f are input into the PWM signal generator to obtain the driving signal of each switch.
[0102] Fig.11 The output current I in the rectifier mode is shown o The waveform of Fig.12 The output current I in inverter mode is shown o From the waveform, we can see that in most of the time period, the reverse current maintains a very low peak value, which greatly reduces the turn-off loss; near the phase zero point, the reverse current value is large, but the voltage is low and the loss is small.
[0103] In another specific embodiment, Fig.10 The frequency conversion phase shift dual mode switching control strategy of the present invention can also be used for Fig.13 The three-phase Swiss modular single-stage AC / DC converter shown includes a low-frequency rectifier module 131, a DC / DC dual active half-bridge module 132, a DC / DC dual active half-bridge module 133, and a DC / DC dual active half-bridge module 134. AC power supply u A The first end of the low-frequency rectifier module 131 is connected to the first input end of the AC power supply u B The first end is connected to the second input end of the low-frequency rectifier module 131, and the AC power supply u C The first end is connected to the third input end of the low-frequency rectifier module 131, and the AC power supply u A The second end of the AC power supply u B The second end of the AC power supply u C The second end of the low-frequency rectifier module 131 is connected to the first output end P and the second output end Y of the low-frequency rectifier module 131 in parallel with the input end of the DC / DC dual active half-bridge module 132. The second output end Y and the third output end N of the low-frequency rectifier module 131 are connected in parallel with the input end of the DC / DC dual active half-bridge module 133. The first output end P and the third output end N of the low-frequency rectifier module 131 are connected in parallel with the input end of the DC / DC dual active half-bridge module 134. The output end of the DC / DC dual active half-bridge module 132 is connected in series with the output end of the DC / DC dual active half-bridge module 133, and then connected in parallel with the output end of the DC / DC dual active half-bridge module 134. The output end of the DC / DC dual active half-bridge module 134 is connected in parallel with the capacitor C 47 With resistor R 41 .
[0104] The low frequency rectifier module 131 includes a diode D ap 、Diode D an 、Diode D bp 、Diode D bn 、Diode D cp 、Diode D cn , Bidirectional switch S ay , Bidirectional switch S by , Bidirectional switch S cy , diode D ap The anode of the diode D is connected an The cathode of diode D bp The anode of the diode D is connected bn The cathode of diode D cp The anode of the diode D is connected cn The cathode of diode D ap The cathode of diode D bp The cathode of diode D cp The cathode of the diode D is connected an The anode of diode D bn The anode of diode D cn Anode connection, AC power supply u A The first end of the diode D ap Anode, AC power supply u B The first end of the diode D bp Anode, AC power supply u c The first end of the diode D cp Anode, AC power supply u A The second end of the AC power supply u B The second end of the AC power supply u C The second end of the diode D ap The anode of the bidirectional switch S ay The first end of the diode D bp The anode of the bidirectional switch S by The first end of the diode D cp The anode of the bidirectional switch S cy The first end of the bidirectional switch S ay The second end of the bidirectional switch S by The second end of the bidirectional switch S cy The second end of the diode D is connected to the second output terminal Y. cp The cathode of the diode D is the first output terminal P. cn The anode of is the third output terminal N.
[0105] The DC / DC dual active half-bridge module 132 includes a capacitor C 41 , capacitor C 42, switch S 41 , switch S 42 , switch Q 41 , switch Q 42 , switch Q 43 , switch Q 44 , capacitor C 41 The second end of the capacitor C 42 The first end of the switch S 41 With switch S 42 Connected in series in the same direction, capacitor C 41 The first end of the switch S 41 The drain capacitor C 42 The second end of the switch S 42 The source of the switch S 41 The source of 44 Connect the first terminal of the primary side of transformer T41, and the second terminal of the primary side of transformer T41 is connected to capacitor C 42 The first end of the switch tube Q 41 With switch Q 42 Connect in series in the same direction, switch Q 43 With switch Q 44 Connect in series in the same direction, switch Q 41 The drain of the switch Q 43 The drain of the switch Q 42 The source of the connected switch Q 44 The source of the transformer T41 is connected to the switch Q 41 The source of the transformer T41 is connected to the switch Q 44 The drain of the first output terminal P passes through the inductor L 41 Connect capacitor C 41 The first end.
[0106] The DC / DC dual active half-bridge module 133 includes a capacitor C 43 , capacitor C 44 , switch S 43 , switch S 44 , switch Q 45 , switch Q 46 , switch Q 47 , switch Q 48 , capacitor C 43 The second end of the capacitor C 42 The first end of the switch S 45 With switch S 46 Connected in series in the same direction, capacitor C 43 The first end of the switch S 43 The drain capacitor C 44 The second end of the switch S 44 The source of the switch S 43 The source of45 Connect the first terminal of the primary side of transformer T42, and the second terminal of the primary side of transformer T42 is connected to capacitor C 44 The first end of the switch tube Q 45 With switch Q 46 Connect in series in the same direction, switch Q 47 With switch Q 48 Connect in series in the same direction, switch Q 45 The drain of the switch Q 47 The drain of the switch Q 46 The source of the connected switch Q 48 The source of the transformer T42 is connected to the switch Q 45 The source of the transformer T42 is connected to the switch Q 48 The drain of the second output terminal Y passes through the inductor L 42 Connect capacitor C 43 The first end.
[0107] The DC / DC dual active half-bridge module 134 includes a capacitor C 45 , capacitor C 46 , switch S 45 , switch S 46 , switch Q 49 , switch Q 410 , switch Q 411 , switch Q 412 , capacitor C 45 The second end of the capacitor C 46 The first end of the switch S 45 With switch S 46 Connected in series in the same direction, capacitor C 45 The first end of the switch S 45 The drain capacitor C 46 The second end of the switch S 46 The source of the switch S 45 The source of 46 Connect the first terminal of the primary side of transformer T43, and the second terminal of the primary side of transformer T43 is connected to capacitor C 45 The first end of the switch tube Q 49 With switch Q 410 Connect in series in the same direction, switch Q 411 With switch Q 412 Connect in series in the same direction, switch Q 49 The drain of the switch Q 411 The drain of the switch Q 410 The source of the connected switch Q 48 The source of the transformer T43 is connected to the switch Q 49 The source of the transformer T43 is connected to the switch Q 412The drain of the first output terminal P passes through the inductor L 43 Connect capacitor C 45 The first end of the switch Q 44 The source of the connected switch Q 47 The drain of the switch Q 43 The drain of the switch Q 411 The drain of the switch Q 48 The source of the connected switch Q 412 The source.
[0108] The bidirectional switch S in the low-frequency rectifier module 131 ay , Bidirectional switch S by , Bidirectional switch S cy At the moment when the two-phase voltages intersect, the low-frequency switch is turned on, and the bidirectional switch of the phase with the larger absolute value of the voltage amplitude is turned off, while the bidirectional switch of the phase with the smaller amplitude is turned on. In this way, the phase with the larger amplitude of the forward voltage passes through the diode D ap or diode D bp or diode D cp Connect to the first output terminal P, the phase with the larger negative voltage amplitude passes through the diode D an or diode D bn or diode D cn The third output terminal N is connected, and the remaining phase is connected through the conducting bidirectional switch S ay Or bidirectional switch S by Or bidirectional switch S cy Connected to the second output terminal Y, the voltage u A 、Voltage u B 、Voltage u C becomes voltage u P 、Voltage u Y 、Voltage u N , we get three positive line voltages u PN 、u PY and u YN , where u PN It is always the largest one (regardless of whether the input voltage is normally supplied or distorted or has three-phase imbalance). The other two voltages u PY and u YN Alternating changes, such as Fig.14 shown.
[0109] The input voltages of the DC / DC dual active half-bridge module 132, the DC / DC dual active half-bridge module 133, and the DC / DC dual active half-bridge module 134 are three line voltages u PN 、u PY and u YN , high frequency switching of each switch, the voltage u PN 、Voltage u PY、Voltage u YN The 0 level is fed into the dual active half-bridge, and then the dual active half-bridge provides soft switching through phase shifting and frequency conversion of high-frequency switching.
[0110] Among them, the voltage u PN 、Voltage u PY 、Voltage u YN Satisfies the following relationship: PN =u PY +u YN , voltage u PY and voltage u YN The corresponding DC / DC dual active half-bridge modules are connected in series so that the output voltage of the two is equal to u PN output voltage, and only one load is connected in parallel at voltage u PN The output end of the corresponding DC / DC dual active half-bridge module has a voltage u PY and voltage u YN The output end of the capacitor is not connected in parallel. The output load and voltage u PN Since the purpose is to achieve PFC, the waveform of the input current of the three DC / DC dual active half-bridge modules should be consistent with the waveform of the input voltage, such as Fig.14 shown.
[0111] Fig.13 The average input current I of the circuit under the single phase shift plus variable frequency modulation control strategy ac It can be expressed as:
[0112]
[0113] Where n is the transformer turns ratio, f is the switching frequency, and L is the phase shift inductor. Fig.13 In the example, for each phase L is the inductance L 44 、Inductance L 45 、Inductance L 46 .
[0114] The expression of the shift ratio D is as follows:
[0115]
[0116] Among them, I B is the reset current, V ac is the AC voltage. The phase shift ratio is the ratio of the phase shift angle to half a period, and its value range is 0-1.
[0117] The expression of switching frequency f is as follows:
[0118]
[0119] In the above formula,
[0120]
[0121] Among them, I ref is the current reference value. Fig.13 For the circuit, the current reference value of each phase is different, namely, current i PN 、Current i PY 、Current i YN , where the current i PN The effective value is equal to the current i PY The effective value of the current i YN Valid values.
[0122] In order to meet the soft switching conditions on the AC side and the soft switching at the maximum voltage, the current is set to 4-5A. However, due to the transformer ratio, the secondary reverse current will have a peak value of 16-20A, which will cause a large turn-off loss. In order to reduce the switching loss, we add a new phase shift angle based on the single phase shift plus variable frequency modulation to become extended phase shift plus variable frequency modulation. In each cycle, there will be some reverse current at the DC output port, such as Fig.15 As shown, its magnitude is proportional to the inductor current at this moment. Similar to single-phase shift, the reset current I B1 ,I B2 Affects the reverse current size of the secondary output current.
[0123] Under the control strategy of extended phase shift plus variable frequency modulation, the average input current I ac It can be expressed as:
[0124]
[0125] Where n is the transformer turns ratio, f is the switching frequency, and L is the converter phase shift inductance. Fig.13 In the example, for each phase L is the inductance L 44 、Inductance L 45 、Inductance L 46 , the expression of the shift ratio D1 and the shift ratio D2 is
[0126]
[0127] Among them, the expressions of parameter A and parameter C are as follows:
[0128]
[0129] Among them, I B1 is the reset current, I B2 is the reset current, V acis the AC voltage. The phase shift ratio is the ratio of the phase shift angle to half a period, and its value range is 0-1.
[0130] More specifically, the switching frequency f is expressed as follows:
[0131]
[0132] in,
[0133]
[0134] Among them, I ref is the current reference value. Fig.13 For the circuit, the current reference value of each phase is different, namely, current i PN 、Current i PY 、Current i YN , where the current i PN The effective value is equal to the current i PY The effective value of the current i YN Valid values.
[0135] Under the control strategy of extended phase shift plus variable frequency modulation, there will be a situation where the phase shift ratio D2 is smaller than the phase shift ratio D1 near the voltage zero crossing point, which will lead to a reduction in converter gain and current distortion, and a frequency surge will occur near the voltage zero crossing point. The smaller the reset current, the higher the frequency. To solve this problem, the Fig.10 The present invention provides a variable frequency phase shift dual mode switching control strategy to control each phase of the three phases, so that on the one hand current distortion can be avoided, and on the other hand the reset current can be very small when the phase shift is extended, thereby greatly reducing the loss.
[0136] It should be noted that the variable frequency phase shift dual-mode switching control strategy of the present invention is not limited to the above two topologies.
[0137] Although the present invention has been disclosed as above by way of embodiments, it is not intended to limit the present invention. Any person having ordinary knowledge in the technical field may make some changes and modifications without departing from the spirit and scope of the present invention. Therefore, the protection scope of the present invention shall be determined by the scope of the attached patent application.
Claims
1. A frequency conversion phase shift dual mode switching control strategy, characterized in that: include, Step S1, calculating the switching frequency f under frequency conversion plus extended phase shift modulation; Step S2, determining whether it is in rectification mode or inversion mode; Step S3, in the rectification mode, the magnitude of the phase shift ratio D1 and the phase shift ratio D2 are calculated in real time and compared; Step S4, if the phase shift ratio D1 is smaller than the phase shift ratio D2, then continue to use frequency conversion plus extended phase shift modulation; Step S5, if the phase shift ratio D1 is not less than the phase shift ratio D2, switching to frequency conversion plus single phase shift modulation; Step S6, in the inverter mode, calculate the shift ratio D2 in real time and compare it with 1; Step S7, if the phase shift ratio D2 is less than 1, continue to use frequency conversion plus extended phase shift modulation; Step S8: If the phase shift ratio D2 is not less than 1, switch to frequency conversion plus single phase shift modulation.
2. A frequency conversion phase shift dual mode switching control strategy as claimed in claim 1, characterized in that: Under frequency conversion plus extended phase shift modulation, the expression of the phase shift ratio D1 and the phase shift ratio D2 is: Among them, the expressions of parameter A and parameter C are as follows: Among them, I B1 ,I B2 is the reset current, n is the transformer turns ratio, V ac is the AC voltage, V dc is the DC voltage, L is the phase-shift inductance, where the phase shift ratio is the proportion of the phase shift angle in half a cycle, and f is the switching frequency.
3. A frequency conversion phase shift dual mode switching control strategy as claimed in claim 2, characterized in that: The expression of the switching frequency f is as follows: in, Among them, I ref is the current reference value.
4. A frequency conversion phase shift dual mode switching control strategy as claimed in claim 3, characterized in that: Average input current I ac It can be expressed as:
5. A frequency conversion phase shift dual mode switching control strategy as claimed in claim 1, characterized in that: Under frequency conversion plus single phase shift modulation, the expression of the phase shift ratio D is as follows: Among them, I B is the reset current, L is the phase shift inductor, V ac is the AC voltage, V dc is the DC voltage, n is the transformer turns ratio, and f is the switching frequency.
6. A frequency conversion phase shift dual mode switching control strategy as claimed in claim 5, characterized in that: The expression of the switching frequency f is as follows: In the above formula, Among them, I ref is the current reference value.
7. A frequency conversion phase shift dual mode switching control strategy as claimed in claim 6, characterized in that: Average input current I ac It can be expressed as:
8. A frequency conversion and phase shift dual-mode switching control strategy as claimed in claim 1, characterized in that: The step S2 also includes measuring power P. If the power P is greater than 0, it is a rectification mode; if the power P is less than 0, it is an inverter mode.
9. A single-stage dual active bridge converter, characterized in that: A variable frequency phase shift dual mode switching control strategy as described in any one of claims 1 to 8 is applied.
10. A three-phase Swiss type modular single-stage AC / DC converter, characterized in that: A variable frequency phase shift dual mode switching control strategy as described in any one of claims 1 to 8 is applied.
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