Electric vehicle charging pile control method based on generalized second-order differential

Abstract

The invention discloses an electric vehicle charging pile control method based on generalized second-order differential, and belongs to the technical field of electrics. The charging pile topology comprises a three-phase alternating current power grid, a filter inductor L, a three-phase VIENNA rectifier, a full-bridge three-level DC / DC converter, an LC filter circuit and a load resistor R. The control method comprises the steps of sampling, rectifier control and converter control. Wherein the three-phase VIENNA rectifier adopts common PI control based on voltage and current double closed loops, and the full-bridge three-level DC / DC converter adopts double-loop control based on a generalized second-order differential PID (Proportion Integration Differentiation) controller. The generalized second-order differential function not only retains the phase lead compensation differential effect on a specific frequency range, but also eliminates the inherent high-frequency noise amplification effect of the traditional differential function. According to the control method, the dynamic performance of the system can be effectively improved, the output signal ripple is small, and the control speed is high.

Description

Technical Field

[0001] The present invention belongs to the field of electrical technology, and in particular relates to a method for controlling an electric vehicle charging pile based on generalized second-order differential. Background Art

[0002] In recent years, the research and development of electric vehicles has become a global hotspot because of their prospect of achieving zero carbon emissions compared to fuel vehicles. Charging piles are the most critical supporting equipment for electric vehicles. In order to better meet the increasing charging needs of electric vehicle users, the research and development of charging piles is also crucial.

[0003] Existing charging piles can be mainly divided into AC charging piles and DC charging piles. DC charging piles have a faster charging speed and are therefore more promising. DC charging piles generally adopt a topological structure of front-end AC / DC and rear-end DC / DC. The front-end has problems such as low power factor, high current harmonic content, and low power utilization. The rear-end has a high switching frequency, which will generate a lot of switching losses and reduce efficiency. In addition, since the charging pile module has a high power level and a wide output voltage range, the dynamic requirements are also high.

[0004] Therefore, experts and scholars at home and abroad have proposed some control methods for the control technology of electric vehicle charging piles, mainly including:

[0005] The paper entitled "Research on Control Strategy of DC Charging Module Based on Three-Level Structure" (Zhao Dongqi. Research on Control Strategy of DC Charging Module Based on Three-Level Structure [D]. Northern Minzu University, 2024.), the front stage of the paper is a three-phase VIENNA rectifier topology, and adopts a dual closed-loop control method based on SVPWM, which effectively improves the power factor and current waveform quality. The back-stage DC converter adopts a half-bridge three-level LLC resonant converter, adopts a hybrid control strategy of PSM and PFM, and proposes a variable frequency burst modulation method based on this control strategy to maintain the dead time and the minimum effective duty cycle, thereby improving the output voltage range for achieving zero voltage switching. However, the front- and rear-stage control methods in the paper are relatively complex, and under high power conditions, the resonant inductor and filter capacitor in the LLC resonant converter are easy to affect the dynamic performance of the system; entitled "Research on Cooperative Control of Three-Level DC Charging Module and Its Front- and Rear-stage Cooperative Control" (Xu Chengwu. Cooperative Control of Three-Level DC Charging Module and Its Front- and Rear-stage Cooperative Control Research [D]. Zhejiang University, 2022.), which also uses a front-stage three-phase VIENNA rectifier and a rear-stage half-bridge three-level LLC resonant converter, and on this basis proposes two control methods for the coordinated work of the front and rear stages to expand the output voltage adjustment range, but the dynamic performance of the DC output voltage in the article is poor; entitled "Design of a high-power charging pile for electric vehicles based on VIENNA rectifier" (Xiong Zilong. Design of a high-power charging pile for electric vehicles based on VIENNA rectifier [D]. Shandong University, 2022.), this article proposes a main circuit structure of a high-power charging pile with an energy storage battery, which is different from the structure of a bidirectional DC / DC converter. The charging and discharging functions of the energy storage battery in this structure are realized by a single-phase DC / DC converter, but the single-phase DC / DC converter in this article uses a two-level full-bridge converter. Compared with the three-level converter, the two-level full-bridge converter has higher voltage withstand requirements for the switch tube and worse reliability in high-voltage and high-power systems.

[0006] In summary, the existing research results have shortcomings, specifically:

[0007] 1. The control method for the front-stage rectifier and the rear-stage DC converter is slightly complicated;

[0008] 2. Most existing charging pile modules have poor system reliability at high power and it is difficult to achieve output in a wide voltage range;

[0009] 3. There is little research on the dynamic stability performance and improvement measures of the output voltage of the charging pile module. Summary of the invention

[0010] The technical problem to be solved by the present invention is the deficiency of the prior art. The present invention provides an electric vehicle charging pile control method based on generalized second-order differential. Compared with the traditional differential function, the generalized second-order differentiator not only retains the phase lead compensation differential effect for a specific frequency range, but also eliminates the high-frequency noise amplification effect inherent in the traditional differential function. It is controlled by a PID controller based on generalized second-order differential, and has good dynamic performance, small output signal ripple and fast control speed.

[0011] The object of the present invention is achieved in this way. The present invention provides an electric vehicle charging pile control method based on generalized second-order differential, wherein the electric vehicle charging pile topology includes a three-phase AC power grid, a three-phase filter inductor, a three-phase VIENNA rectifier, a full-bridge three-level DC / DC converter, an LC filter circuit, and a load resistor; the LC filter circuit includes a filter inductor L o And the voltage stabilizing capacitor C o ;

[0012] The full-bridge three-level DC / DC converter includes a primary side H bridge, a high-frequency transformer T and a secondary side H bridge, wherein the primary side H bridge includes two bridge arms connected in parallel between a DC positive bus and a DC negative bus, each bridge arm includes four switch tubes G with parallel parasitic capacitance and anti-parallel diodes. mj and 2 freewheeling diodes F mi , m is the phase sequence, m = a, b; j and i are the numbers of the switch tube and the freewheeling diode, j = 1, 2, 3, 4, i = 1, 2; the switch tube G m1 The collector is connected to the DC positive bus, and the emitter is connected to the switch tube G m2 The collector of the freewheeling diode F m1 The negative pole of the switch tube G m2 The emitter is connected to the switch tube G m3 The collector of m , m=a,b; switch tube G m3 The emitter of the switch tube G m4 The collector of the freewheeling diode F m2 The positive pole of the freewheeling diode F m2 The cathode of the freewheeling diode F m1 The positive pole of the switch tube G m4 The emitter of the high-frequency transformer T is connected to the negative DC bus; the positive electrode of the input end of the high-frequency transformer T is connected in series with a primary side DC blocking capacitor C r , a primary side resonant inductor L r Rear contact P a , the negative contact P at the input end of the high-frequency transformer T b The secondary side H bridge includes 4 diodes, among which diode D 5 The positive electrode of the diode D 6The negative pole of the diode D is connected to the positive pole of the output terminal of the high-frequency transformer T. 7 The positive electrode of the diode D 8 The negative pole of the high frequency transformer T is connected to the negative pole of the output terminal;

[0013] The three-phase AC power grid is connected in series with a three-phase filter inductor, a three-phase VIENNA rectifier, and a full-bridge three-level DC / DC converter in sequence. The diode D 5 The cathode of diode D 7 The negative pole of the filter inductor L o One end of the filter inductor L o The other end is connected to the voltage stabilizing capacitor C o The positive pole of diode D 6 The positive electrode of diode D 8 The positive pole is connected to the voltage stabilizing capacitor C o The negative pole, load resistor and voltage stabilizing capacitor C o in parallel;

[0014] The control method comprises the following steps:

[0015] Step 1: Data sampling

[0016] Sample the output voltage of the three-phase VIENNA rectifier and record it as the rectifier output voltage U 1 , sample the three-phase filter inductor current and record it as the input current I a ,I b ,I c , sample the three-phase AC grid voltage and record it as the input voltage E a ,E b ,E c , sample the voltage of the load resistor and record it as the output voltage U 2 , sample the current flowing through the load resistor and record it as the output current I 2 ;

[0017] Step 2, Control of three-phase VIENNA rectifier

[0018] Step 2.1, set the input voltage E a ,E b ,E c The grid phase angle θ is obtained through the phase-locked loop output. Combined with the grid phase angle θ, the input current I a ,I b ,I c And the input voltage E a ,E b ,E c Perform single synchronous rotating coordinate transformation to obtain the input current dq axis component I d ,Iq And the input voltage dq axis component E d ,E q ;

[0019] Step 2.2, give the three-phase VIENNA rectifier output voltage command value and record it as the rectifier output voltage command value U 1 * , the rectifier input voltage command value U 1 * With the rectifier output voltage U 1 The first voltage error signal is obtained by subtracting, and the first voltage error signal is used as the input of the first PI controller, and the output is the input current d-axis component command value I d * ;

[0020] Step 2.3: Give the input current q-axis component command value I q * , and according to the input current d-axis component command value I d * And the input current q-axis component command value I q * , the control signal U is obtained through the second PI controller d ,U q ;

[0021] Step 2.4, combined with the grid phase angle θ, the obtained control signal U d ,U q The control signal U in the stationary coordinate system is obtained by single synchronous rotating coordinate inverse transformation. a ,U b ,U c , and then the PWM generator generates the PWM drive signal PWMQ of the three-phase VIENNA rectifier A ,PWMQ B ,PWMQ C , driving signal PWMQ A ,PWMQ B ,PWMQ C The phase difference is 120°, and the frequency is f s1 ;

[0022] Step 3: Control of the full-bridge three-level DC / DC converter

[0023] Step 3.1, set the duty cycle command value D * ;

[0024] Step 3.2, set the voltage command value of the load resistance and record it as the output voltage command value U 2 * , the output voltage command value U2 * And the output voltage U 2 The second voltage error signal is obtained by subtracting, and the second voltage error signal is used as the input of the third PI controller based on generalized second-order differential, and the output is the output current command value I 2 * ;

[0025] Step 3.3: Set the output current command value I 2 * and output current I 2 The current error signal is obtained by subtracting, and then the current error signal is input into the fourth PI controller, and the output of the fourth PI controller and the duty cycle command value D are used to obtain the current error signal. * , the phase shift angle γ is calculated by the phase shift equation;

[0026] Step 3.4: Set the duty cycle command value D * The phase shift angle γ is used to generate the drive signal PWMG through the PWM generator. 1 and the drive signal PWMG 2 , where the drive signal PWMG 1 and the drive signal PWMG 2 The phase difference is a phase shift angle γ; the drive signal PWMG 1 The driving signal PWMG is generated by an inverse generator 3 , driving signal PWMG 2 The driving signal PWMG is generated by an inverse generator 4 ; Switching tube G a1 , switch tube G b4 Both are driven by the drive signal PWMG 1 Driver, switch tube G a4 , switch tube G b1 Both are driven by the drive signal PWMG 3 Driver, switch tube G a2 , switch tube G b3 Both are driven by the drive signal PWMG 2 Driver, switch tube G a3 , switch tube G b2 Both are driven by the drive signal PWMG 4 Drive, the drive signal frequency is f s2 .

[0027] Preferably, the input current d-axis component command value I in step 2.2 is d * The expression is:

[0028]

[0029] In the formula, K p1is the proportional coefficient of the first PI controller, K i1 is the integral coefficient of the first PI controller, s is the Laplace operator;

[0030] Step 2.3: Control signal U d ,U q Its expression is:

[0031]

[0032] In the formula, K p2 is the proportional coefficient of the second PI controller, K i2 is the integral coefficient of the second PI controller.

[0033] Preferably, the output current command value I in step 3.2 is 2 * The expression is:

[0034]

[0035] Where s is the Laplace operator, K p3 is the proportional coefficient of the third PI controller based on generalized second-order differential, K i3 is the integral coefficient of the third PI controller based on generalized second-order differential, G(s) is the generalized second-order differential transfer function, K h is the proportional coefficient of the generalized second-order differential transfer function, Q h is the quality factor of the generalized second-order differential transfer function, ω h is the angular frequency of the generalized second-order differential transfer function;

[0036] The phase shift equation in step 3.3 is:

[0037]

[0038] In the formula, K p4 is the proportional coefficient of the fourth PI controller, K i4 is the integral coefficient of the fourth PI controller.

[0039] Compared with the prior art, the present invention has the following beneficial effects:

[0040] 1. Compared with the PI controller, the PID controller based on generalized second-order differential has a phase lead compensation function in a specific frequency range, which can make up for the phase lag caused by the integral link, thereby increasing the phase margin and improving the dynamic stability performance of the system;

[0041] 2. Compared with the ordinary differential link, the generalized second-order differential function retains the phase lead compensation differential effect in a specific frequency range, and there is no amplification characteristic in the high-frequency band, thereby reducing the interference of high-frequency noise on the system and improving the reliability of the system. The generalized second-order differential function can be adjusted by adjusting the coefficient K h and the angular frequency ω h The phase angle compensation can be controlled near the bandwidth frequency;

[0042] 3. The three-level full-bridge conversion circuit adopted in the present invention is controlled by a PID controller based on generalized second-order differential, and has good output dynamic performance, fast control speed, and small ripple. BRIEF DESCRIPTION OF THE DRAWINGS

[0043] Figure 1 It is a topological structure diagram of the electric vehicle charging pile in the present invention.

[0044] Figure 2 It is a control block diagram of the three-phase VIENNA rectifier in the present invention.

[0045] Figure 3 It is a control block diagram of the full-bridge three-level DC / DC converter in the present invention.

[0046] Figure 4 is the output voltage U of the three-phase VIENNA rectifier in the simulation of the embodiment of the present invention 1 Schematic diagram of the changes.

[0047] Figure 5 The output voltage U on the load R when the PI control method and the proposed control method are used in the simulation of the embodiment of the present invention is 2 Schematic diagram of changes.

[0048] Figure 6 The output voltage U on the load R when the PI control method and the proposed control method are used in the simulation of the embodiment of the present invention is 2 Schematic diagram of steady-state waveform.

[0049] Figure 7 When the PID control method and the proposed control method are used in the simulation of the embodiment of the present invention, the output voltage U on the load R 2 Schematic diagram of changes. DETAILED DESCRIPTION

[0050] The present invention is described in detail below in conjunction with the accompanying drawings of the present invention.

[0051] Figure 1 This is a topological diagram of the electric vehicle charging pile used in the present invention. Figure 1It can be seen that the electric vehicle charging pile topology of the present invention includes a three-phase AC power grid, a three-phase filter inductor, a three-phase VIENNA rectifier, a full-bridge three-level DC / DC converter, an LC filter circuit, and a load resistor; the LC filter circuit includes a filter inductor L o And the voltage stabilizing capacitor C o .

[0052] The full-bridge three-level DC / DC converter includes a primary side H bridge, a high-frequency transformer T and a secondary side H bridge, wherein the primary side H bridge includes two bridge arms connected in parallel between a DC positive bus and a DC negative bus, each bridge arm includes four switch tubes G with parallel parasitic capacitance and anti-parallel diodes. mj and 2 freewheeling diodes F mi , m is the phase sequence, m = a, b; j and i are the numbers of the switch tube and the freewheeling diode, j = 1, 2, 3, 4, i = 1, 2; the switch tube G m1 The collector is connected to the DC positive bus, and the emitter is connected to the switch tube G m2 The collector of the freewheeling diode F m1 The negative pole of the switch tube G m2 The emitter is connected to the switch tube G m3 The collector of m , m=a,b; switch tube G m3 The emitter of the switch tube G m4 The collector of the freewheeling diode F m2 The positive pole of the freewheeling diode F m2 The cathode of the freewheeling diode F m1 The positive pole of the switch tube G m4 The emitter of the high-frequency transformer T is connected to the negative DC bus; the positive electrode of the input end of the high-frequency transformer T is connected in series with a primary side DC blocking capacitor C r , a primary side resonant inductor L r Rear contact P a , the negative contact P at the input end of the high-frequency transformer T b The secondary side H bridge includes 4 diodes, among which diode D 5 The positive electrode of the diode D 6 The negative pole of the diode D is connected to the positive pole of the output terminal of the high-frequency transformer T. 7 The positive electrode of the diode D 8 The negative pole of the high-frequency transformer T is connected to the negative pole of the output terminal.

[0053] The three-phase AC power grid is connected in series with a three-phase filter inductor, a three-phase VIENNA rectifier, and a full-bridge three-level DC / DC converter in sequence. The diode D 5 The cathode of diode D7 The negative pole of the filter inductor L o One end of the filter inductor L o The other end is connected to the voltage stabilizing capacitor C o The positive pole of diode D 6 The positive electrode of diode D 8 The positive pole is connected to the voltage stabilizing capacitor C o The negative pole, load resistor and voltage stabilizing capacitor C o in parallel.

[0054] exist Figure 1 On the top, L is the three-phase filter inductor, E a ,E b ,E c is a three-phase AC power grid. Figure 1 It can be seen that the three-phase VIENNA rectifier in the present invention includes 6 switch tubes with reverse parallel diodes, 6 freewheeling diodes and two rectifier stabilizing capacitors. The rectifier stabilizing capacitor C 1 The negative electrode and the rectifier voltage stabilizing capacitor C 2 The positive pole of the rectifier is connected to the voltage stabilizing capacitor C 1 The positive electrode and the rectifier voltage stabilizing capacitor C 2 The negative electrode of the rectifier constitutes a DC output port of the VIENNA rectifier, and the DC positive bus and the DC negative bus are led out from the DC output port. Figure 1 Up, Q a1 , Q a2 , Q b1 , Q b2 , Q c1 , Q c2 Represents 6 switch tubes with reverse parallel diodes, D a1 , D a2 , D b1 , D b2 , D c1 , D c2 Represents 6 freewheeling diodes.

[0055] In this embodiment, the three-phase AC power grid provides a three-phase AC voltage with a frequency of 50 Hz and an effective value of 220 V. The inductance of the three-phase filter inductor L is 240 uH, and the rectifier voltage stabilizing capacitor C 1 , C 2 The capacitance value is 1680uF, and the primary side resonant inductor L r The inductance is 4.2uH, and the primary side DC blocking capacitor C r The capacitance is 10uF, the single-phase transformer ratio is 1:1, and the filter inductor L o The inductance is 227uH, the voltage stabilizing capacitor C o The capacitance is 170uF and the load R is 6.25Ω.

[0056] Figure 2 is a control block diagram of the three-phase VIENNA rectifier of the present invention, Figure 3 is a control block diagram of a full-bridge three-level DC / DC converter of the present invention, Figure 2 , Figure 3 It can be seen that the control method comprises the following steps:

[0057] Step 1: Data sampling

[0058] Sample the output voltage of the three-phase VIENNA rectifier and record it as the rectifier output voltage U 1 , sample the three-phase filter inductor current and record it as the input current I a ,I b ,I c , sample the three-phase AC grid voltage and record it as the input voltage E a ,E b ,E c , sample the voltage of the load resistor and record it as the output voltage U 2 , sample the current flowing through the load resistor and record it as the output current I 2 .

[0059] Step 2, Control of three-phase VIENNA rectifier

[0060] Step 2.1, set the input voltage E a ,E b ,E c The grid phase angle θ is obtained through the phase-locked loop output. Combined with the grid phase angle θ, the input current I a ,I b ,I c And the input voltage E a ,E b ,E c Perform single synchronous rotating coordinate transformation to obtain the input current dq axis component I d ,I q And the input voltage dq axis component E d ,E q ;

[0061] Step 2.2, give the three-phase VIENNA rectifier output voltage command value and record it as the rectifier output voltage command value U 1 * , the rectifier output voltage command value U 1 * With the rectifier output voltage U 1 The first voltage error signal is obtained by subtracting, and the first voltage error signal is used as the input of the first PI controller, and the output is the input current d-axis component command value I d * ;

[0062] Step 2.3: Give the input current q-axis component command value I q * , and according to the input current d-axis component command value I d * And the input current q-axis component command value I q * , the control signal U is obtained through the second PI controller d ,U q ;

[0063] Step 2.4, combined with the grid phase angle θ, the obtained control signal U d ,U q The control signal U in the stationary coordinate system is obtained by single synchronous rotating coordinate inverse transformation. a ,U b ,U c , and then the PWM generator generates the PWM drive signal PWMQ of the three-phase VIENNA rectifier A ,PWMQ B ,PWMQ C , driving signal PWMQ A ,PWMQ B ,PWMQ C The phase difference is 120°, and the frequency is f s1 .

[0064] In this embodiment, the input current d-axis component command value I d * The expression is:

[0065]

[0066] In the formula, K p1 is the proportional coefficient of the first PI controller, K i1 is the integral coefficient of the first PI controller, and s is the Laplace operator.

[0067] Step 2.3: Control signal U d ,U q Its expression is:

[0068]

[0069] In the formula, K p2 is the proportional coefficient of the second PI controller, K i2 is the integral coefficient of the second PI controller.

[0070] In this embodiment, U 1 * =700V, I q* =0, K p1 =0.65, K i1 =65,K p2 =0.01, K i2 =0.4, frequency f s1 =16kHz.

[0071] Step 3: Control of the full-bridge three-level DC / DC converter

[0072] Step 3.1, set the duty cycle command value D * ;

[0073] Step 3.2, set the voltage command value of the load resistance and record it as the output voltage command value U 2 * , the output voltage command value U 2 * And the output voltage U 2 The second voltage error signal is obtained by subtracting, and the second voltage error signal is used as the input of the third PI controller based on generalized second-order differential, and the output is the output current command value I 2 * ;

[0074] Step 3.3: Set the output current command value I 2 * and output current I 2 The current error signal is obtained by subtracting, and then the current error signal is input into the fourth PI controller, and the output of the fourth PI controller and the duty cycle command value D are used to obtain the current error signal. * , the phase shift angle γ is calculated by the phase shift equation;

[0075] Step 3.4: Set the duty cycle command value D * The phase shift angle γ is used to generate the drive signal PWMG through the PWM generator. 1 and the drive signal PWMG 2 , where the drive signal PWMG 1 and the drive signal PWMG 2 The phase difference is a phase shift angle γ; the drive signal PWMG 1 The driving signal PWMG is generated by an inverse generator 3 , driving signal PWMG 2 The driving signal PWMG is generated by an inverse generator 4 ; Switching tube G a1 , switch tube G b4 Both are driven by the drive signal PWMG 1 Driver, switch tube G a4 , switch tube G b1 Both are driven by the drive signal PWMG3, the switch tube Ga2 , switch tube G b3 Both are driven by the drive signal PWMG2, and the switch tube G a3 , switch tube G b2 Both are driven by the drive signal PWMG 4 Drive, the drive signal frequency is f s2 .

[0076] In this embodiment, the output current command value I 2 * The expression is:

[0077]

[0078] Where s is the Laplace operator, K p3 is the proportional coefficient of the third PI controller based on generalized second-order differential, K i3 is the integral coefficient of the third PI controller based on generalized second-order differential, G(s) is the generalized second-order differential transfer function, K h is the proportional coefficient of the generalized second-order differential transfer function, Q h is the quality factor of the generalized second-order differential transfer function, ω h is the angular frequency of the generalized second-order differential transfer function.

[0079] The phase shift equation is:

[0080]

[0081] In the formula, K p4 is the proportional coefficient of the fourth PI controller, K i4 is the integral coefficient of the fourth PI controller.

[0082] In this embodiment, the output voltage command value U 2 * is the step command value, before the step U 2 * =450V, after step U 2 * =500V, D * =0.5, K p3 =0.8, K i3 =800,K p4 =0.0025, K i4 =12.5, K h =8,Q h =1,ω h =6000π, f s2 =70kHz.

[0083] In order to verify the theory of the present invention, Matlab / Simulink simulation software is used to perform simulation.

[0084] Figure 4 The output voltage U of the three-phase VIENNA rectifier is given as 1 Waveform, by Figure 4 It can be seen that the output voltage U 1 Can maintain stable operation, at t = 0.1s, U 2 * Generate a step, output voltage U 1 The waveform can always maintain stable operation at 700V and has good dynamic response characteristics.

[0085] In order to verify the advantages of the control method based on generalized second-order differential, the PID controller based on generalized second-order differential is compared with the ordinary PI controller, and the PI parameters and DC converter topology are kept consistent. Figure 5 The output voltage U of the load resistor R under the PID control based on generalized second-order differential and the ordinary PI control are given. 2 By comparing the waveforms, it can be found that when the system output voltage command value U 2 * When a step occurs at t=0.1s, compared with the PID controller based on generalized second-order differential, the transient waveform under the ordinary PI controller shows a relatively large oscillation phenomenon, and the oscillation decays more slowly, reaching a steady state at t=0.11s. The PID control based on generalized second-order differential can effectively reduce the oscillation process and make the waveform reach stability faster. In addition, according to Figure 6 It can be seen that in terms of voltage ripple, PID control based on generalized second-order differential also makes the output steady-state voltage ripple smaller. Figure 7 The output voltage U of the load resistor R under the PID control based on generalized second-order differential and the ordinary PID control is given. 2 In order to reflect the characteristic that the traditional differential link will amplify the high-frequency noise, Gaussian white noise is introduced into the feedback loop of the control system. It can be found that under the ordinary PID control method, the output voltage waveform is obviously interfered by the noise and the quality is poor.

Claims

1. A control method for an electric vehicle charging pile based on generalized second-order differential, wherein the electric vehicle charging pile topology includes a three-phase AC power grid, a three-phase filter inductor, a three-phase VIENNA rectifier, a full-bridge three-level DC / DC converter, an LC filter circuit, and a load resistor; the LC filter circuit includes a filter inductor L o And the voltage stabilizing capacitor C o ; The full-bridge three-level DC / DC converter includes a primary side H bridge, a high-frequency transformer T and a secondary side H bridge, wherein: The primary side H bridge includes two bridge arms connected in parallel between the DC positive bus and the DC negative bus, each bridge arm includes four switch tubes G with parallel parasitic capacitance and anti-parallel diodes. mj and 2 freewheeling diodes F mi , m is the phase sequence, m = a, b; j and i are the numbers of the switch tube and the freewheeling diode, j = 1, 2, 3, 4, i = 1, 2; the switch tube G m1 The collector is connected to the DC positive bus, and the emitter is connected to the switch tube G m2 The collector of the freewheeling diode F m1 The negative pole of the switch tube G m2 The emitter is connected to the switch tube G m3 The collector of m , m=a,b; switch tube G m3 The emitter of the switch tube G m4 The collector of the freewheeling diode F m2 The positive pole of the freewheeling diode F m2 The cathode of the freewheeling diode F m1 The positive pole of the switch tube G m4 The emitter of the high-frequency transformer T is connected to the negative DC bus; the positive electrode of the input end of the high-frequency transformer T is connected in series with a primary side DC blocking capacitor C r , a primary side resonant inductor L r Rear contact P a , the negative contact P at the input end of the high-frequency transformer T b The secondary side H bridge includes 4 diodes, wherein the anode of diode D5 is connected to the cathode of diode D6, and the connection point thereof is connected to the anode of the output terminal of the high-frequency transformer T, and the anode of diode D7 is connected to the cathode of diode D8, and the connection point thereof is connected to the cathode of the output terminal of the high-frequency transformer T; The three-phase AC power grid is connected in series with the three-phase filter inductor, the three-phase VIENNA rectifier, and the full-bridge three-level DC / DC converter in sequence. The cathode of the diode D5 and the cathode of the diode D7 on the secondary side of the full-bridge three-level DC / DC converter are connected to the filter inductor L o One end of the filter inductor L o The other end is connected to the voltage stabilizing capacitor C o The positive electrode of the diode D6 and the positive electrode of the diode D8 are connected to the voltage stabilizing capacitor C o The negative pole, load resistor and voltage stabilizing capacitor C o in parallel; Characterized in that the control method comprises the following steps: Step 1: Data sampling The output voltage of the three-phase VIENNA rectifier is sampled and recorded as the rectifier output voltage U1, and the three-phase filter inductor current is sampled and recorded as the input current I a ,I b ,I c , sample the three-phase AC grid voltage and record it as the input voltage E a ,E b ,E c , sample the voltage of the load resistor and record it as the output voltage U2, sample the current flowing through the load resistor and record it as the output current I2; Step 2, Control of three-phase VIENNA rectifier Step 2.1, set the input voltage E a ,E b ,E c The grid phase angle θ is obtained through the phase-locked loop output. Combined with the grid phase angle θ, the input current I a ,I b ,I c And the input voltage E a ,E b ,E c Perform single synchronous rotating coordinate transformation to obtain the input current dq axis component I d ,I q And the input voltage dq axis component E d ,E q ; Step 2.2, give the three-phase VIENNA rectifier output voltage command value and record it as the rectifier output voltage command value U1 * , the rectifier output voltage command value U1 * The first voltage error signal is obtained by subtracting the rectifier output voltage U1, and the first voltage error signal is used as the input of the first PI controller, and the output is the input current d-axis component command value I d * ; Step 2.3: Give the input current q-axis component command value I q * , and according to the input current d-axis component command value I d * And the input current q-axis component command value I q * , the control signal U is obtained through the second PI controller d ,U q ; Step 2.4, combined with the grid phase angle θ, the obtained control signal U d ,U q The control signal U in the stationary coordinate system is obtained by single synchronous rotating coordinate inverse transformation. a ,U b ,U c , and then the PWM generator generates the PWM drive signal PWMQ of the three-phase VIENNA rectifier A ,PWMQ B ,PWMQ C , driving signal PWMQ A ,PWMQ B ,PWMQ C The phase difference is 120°, and the frequency is f s1 ; Step 3: Control of the full-bridge three-level DC / DC converter Step 3.1, set the duty cycle command value D * ; Step 3.2, set the voltage command value of the load resistance and record it as the output voltage command value U2 * , the output voltage command value U2 * The second voltage error signal is obtained by subtracting the output voltage U2, and the second voltage error signal is used as the input of the third PI controller based on generalized second-order differential, and the output is the output current command value I2 * ; Step 3.3, set the output current command value I2 * The current error signal is obtained by subtracting the output current I2, and then the current error signal is input into the fourth PI controller, and the duty cycle command value D is calculated according to the output of the fourth PI controller. * , the phase shift angle γ is calculated by the phase shift equation; Step 3.4: Set the duty cycle command value D * The phase shift angle γ is passed through the PWM generator to generate a drive signal PWMG1 and a drive signal PWMG2, wherein the drive signal PWMG1 and the drive signal PWMG2 differ in phase by a phase shift angle γ; the drive signal PWMG1 is passed through an inverse generator to generate a drive signal PWMG3, and the drive signal PWMG2 is passed through an inverse generator to generate a drive signal PWMG4; the switch tube G a1 , switch tube G b4 Both are driven by the drive signal PWMG1, the switch tube G a4 , switch tube G b1 Both are driven by the drive signal PWMG3, the switch tube G a2 , switch tube G b3 Both are driven by the drive signal PWMG2, the switch tube G a3 , switch tube G b2 They are all driven by the drive signal PWMG4, and the drive signal frequency is f s2 .

2. The electric vehicle charging pile control method based on generalized second-order differential according to claim 1, characterized in that: Step 2.2: Input current d-axis component command value I d * The expression is: In the formula, K p1 is the proportional coefficient of the first PI controller, K i1 is the integral coefficient of the first PI controller, s is the Laplace operator; Step 2.3: Control signal U d ,U q Its expression is: In the formula, K p2 is the proportional coefficient of the second PI controller, K i2 is the integral coefficient of the second PI controller.

3. The electric vehicle charging pile control method based on generalized second-order differential according to claim 1 is characterized in that: Step 3.2 Output current command value I2 * The expression is: Where s is the Laplace operator, K p3 is the proportional coefficient of the third PI controller based on generalized second-order differential, K i3 is the integral coefficient of the third PI controller based on generalized second-order differential, G(s) is the generalized second-order differential transfer function, K h is the proportional coefficient of the generalized second-order differential transfer function, Q h is the quality factor of the generalized second-order differential transfer function, ω h is the angular frequency of the generalized second-order differential transfer function; The phase shift equation in step 3.3 is: In the formula, K p4 is the proportional coefficient of the fourth PI controller, K i4 is the integral coefficient of the fourth PI controller.

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