A motor weak magnetic field model predictive control method and device with integrated modulation function

Abstract

The present invention discloses a motor weak magnetic field model predictive control method with integrated modulation function, comprising the following steps: step 1, establishing a current prediction model; step 2, respectively establishing a first cost function for solving the optimal duty ratio and a second cost function for solving the suboptimal duty ratio based on the current prediction model, and determining the final duty ratio based on the solution result of the first cost function; step 3, generating a three-phase switching signal based on the final duty ratio, and constructing a weak magnetic field regulator with a value of 1 and the maximum value of the optimal duty ratio; step 4, applying the obtained three-phase switching signal to a two-level voltage source three-phase inverter, and adjusting the inverter using a weak magnetic field regulator until the voltage limit is extended to a voltage hexagon. The present invention also provides a motor weak magnetic field model predictive control device. The method provided by the present invention can effectively improve the operating efficiency of a permanent magnet synchronous motor under weak magnetic field conditions.

Description

Technical Field

[0001] The present invention belongs to the technical field of permanent magnet synchronous motor control, and in particular relates to a motor weak magnetic field region model predictive control method and device with integrated modulation function. Background Art

[0002] Permanent magnet synchronous motors are widely used in electric vehicles, spindle drives and other fields due to their simple structure, high power density and wide speed regulation range. The above applications require the motor to have a wide speed regulation range. Nowadays, model predictive control has the advantages of flexible implementation methods, multivariable control and fast dynamic response speed. It can replace traditional controllers to handle complex nonlinearities and constraints, and has received more and more attention and extensive research in high-performance applications. With the continuous development of model predictive control technology, model predictive control methods have been extended to high-speed operation conditions of motors and have received more and more research. Since the back electromotive force of the permanent magnet synchronous motor increases accordingly with the increase of motor speed, in order to overcome the increasing back electromotive force, the output voltage of the inverter needs to be continuously increased, but the DC bus voltage of the voltage source inverter is fixed and limited. The commonly used space vector pulse width modulation technology is mainly aimed at the linear modulation area, that is, the voltage within the inscribed circle of the voltage hexagon. Therefore, when the motor speed rises to a certain level, the voltage output by the inverter under space vector pulse width modulation cannot meet the control requirements of the motor. To address the above problems, scholars have adopted two types of solutions: one uses a weak magnetic algorithm to reduce the motor drive's demand for the DC bus voltage, thereby expanding the motor drive system's speed range and improving torque output capabilities; the other uses overmodulation technology to improve the inverter's voltage output capability.

[0003] After the voltage limit is extended to the voltage hexagon, overmodulation technology is needed to improve the voltage output capability of the inverter. Overmodulation is generally divided into single-mode overmodulation and dual-mode overmodulation. Compared with the single-mode overmodulation algorithm, dual-mode overmodulation can improve the voltage output quality of the inverter. However, the calculation process of overmodulation technology is relatively complicated. In addition, the overmodulation algorithm will make the actual output voltage of the inverter inconsistent with the reference voltage obtained by the control algorithm, so that the actual control result of the motor deviates from the calculation result of the control strategy, reducing the dynamic response speed of the motor system.

[0004] Patent document CN112468043A discloses a voltage open-loop permanent magnet synchronous motor torque control method, wherein the method includes: receiving a torque command, and obtaining a voltage vector angle corresponding to the torque value according to the torque value in the torque command and a preset data table; obtaining control voltages in the d-axis and q-axis directions according to the voltage vector angle and the fundamental phase voltage amplitude; obtaining the phase voltages of phases A, B, and C according to the control voltage rotor position angle; judging the switching state of the switch tube of each bridge arm according to the positive and negative polarity of each phase voltage and determining the duration of the switching state; and controlling the opening and closing of the switch tube in each bridge arm according to the above-mentioned switching state and duration, thereby realizing the output torque control of the motor.

[0005] Patent document CN118508801A discloses a weak magnetic control method, device and controller for an asynchronous motor, wherein the weak magnetic control method comprises: obtaining a given value and an actual value of an excitation current, a given value and an actual value of a torque current, and performing control processing with a current regulator to obtain a given voltage; performing space vector modulation on the given voltage to obtain a control signal corresponding to an inverter; and controlling the inverter according to the control signal to drive the motor; wherein the given value of the excitation current is the rated value of the excitation current plus the voltage closed-loop output value, and the output value is the difference between the optimized reference voltage and the given voltage of the previous cycle, obtained through proportional-integral regulation and limiter processing, and the optimized reference voltage is a no-load voltage and a load voltage formed in an orthogonal mode. Summary of the invention

[0006] The object of the present invention is to provide a motor weak magnetic field region model predictive control method and device with integrated modulation function, which can effectively improve the operating efficiency of a permanent magnet synchronous motor under weak magnetic field conditions.

[0007] In order to achieve the first purpose of the present invention, the following technical solutions are provided: a motor weak magnetic zone model predictive control method with integrated modulation function, comprising the following steps: step 1, taking the duty cycle of the three-phase switch signal of the two-level voltage source three-phase inverter as the control quantity, taking the current output of the permanent magnet synchronous motor where the two-level voltage source three-phase inverter is located as the prediction quantity, and establishing a corresponding current prediction model; step 2, constructing a first cost function based on the current prediction model, the first cost function is used to solve the optimal duty cycle of the three-phase switch signal of the two-level voltage source three-phase inverter; constructing a second cost function based on the current prediction model and in combination with preset constraints, the first cost function is used to solve the optimal duty cycle of the three-phase switch signal of the two-level voltage source three-phase inverter; the second cost function is used to solve the optimal duty cycle of the three-phase switch signal of the two-level voltage source three-phase inverter; the second cost function is used to solve the optimal duty cycle of the three-phase switch signal of the two-level voltage source three-phase inverter; the first ... The second cost function is used to solve the suboptimal duty cycle of the three-phase switch signal of the two-level voltage source three-phase inverter; the final duty cycle is selected by judging the solution result of the first cost function: when any amplitude of the optimal duty cycle of the three-phase switch signal obtained by solving the first cost function is less than or equal to 1, the optimal duty cycle of the three-phase switch signal is used as the final duty cycle; conversely, when any amplitude is greater than 1, the suboptimal duty cycle of the three-phase switch signal obtained by solving the second cost function is used as the final duty cycle; step 3, converting the final duty cycle obtained in step 2, and generating the corresponding three-phase switch signal according to the conversion result; step 4, using the difference between the value 1 and the maximum value of the optimal duty cycle of the three-phase switch signal obtained in step 2 as the input to construct a weak magnetic regulator, the output of the weak magnetic regulator is used to calculate the d-axis reference current used in the first cost function and the second cost function in the next control cycle.

[0008] The present invention starts with improving the voltage output capacity of the inverter, and proposes a model predictive control algorithm that takes the duty cycle of the inverter's three-phase switching signal as the control quantity. The three-phase duty cycle of the inverter is directly calculated, eliminating the execution links of space vector modulation and overmodulation. Not only can the voltage output by the inverter cover the entire voltage hexagon, but also the output voltage calculated by the model predictive control can be consistent with the actual output voltage of the inverter.

[0009] Specifically, the expression of the current prediction model is as follows: In the formula, i d k+1 and i q k+1 Respectively represent ( k +1) T s The predicted values ​​of d and q axis currents at the time; i d k and i q k Respectively kT sThe d and q axis current measurement values ​​at the time; T s Indicates the control cycle; R s represents the stator resistance; L d , L q Respectively represent the d-axis and q-axis stator inductances; ω r k represents the angular velocity of the rotor; ψ f represents the flux linkage of permanent magnet; V dc Indicates the DC bus voltage of the inverter; d A k , d B k and d C k Respectively kT s The duty cycle of the inverter A phase switching signal, the duty cycle of the inverter B phase switching signal and the duty cycle of the inverter C phase switching signal at the moment; θ r Indicates the electrical angle of the motor rotor.

[0010] Specifically, the expression of the first cost function is as follows: In the formula, J 1 represents the first cost function; i d * and i q * are the reference values ​​of d-axis and q-axis currents respectively; i d k+2 and i q k+2 Respectively represent ( k +2) T s The predicted values ​​of d and q axis currents at the time; d A1 k+1 and d B1 k+1 Respectively represent ( k +1) T s The duty cycle of the inverter A phase switching signal and the duty cycle of the inverter B phase switching signal at the moment.

[0011] Specifically, the process of solving the optimal duty cycle of the three-phase switch signal is as follows: d A1 ≥0, d B1 ≥0, then ;if d A1 <0, d A1 < d B1，则 ;if d B1 <0, d B1 < d A1 ,but ; In the formula, d AO k+1 , d BO k+1 and d CO k+1 Respectively represent ( k +1) T s The optimal duty cycle of the inverter A phase switching signal, the optimal duty cycle of the inverter B phase switching signal and the optimal duty cycle of the inverter C phase switching signal at the moment.

[0012] Specifically, the expression of the second cost function is as follows: In the formula, J 2 represents the second cost function, d xw k+1 express( k +1) T s Inverter at the moment x Suboptimal duty cycle of the phase switching signals, d yw k+1 express( k +1) T s Inverter at the moment y Suboptimal duty cycle of the phase switching signals, d zw k+1 express( k +1) T s Inverter at the moment z The suboptimal duty cycle of the phase switching signal; d AO >1, dAO > d BO and d CO =0, d xw k+1 = d Bw k+1 , d yw k+1 = d Aw k+1 , d zw k+1 = d Cw k+1 ;when d AO >1, d AO >d CO and d BO =0, d xw k+1 = d Cw k+1 , d yw k+1 = d Aw k+1 , d zw k+1 = d Bw k+1 ;when d BO >1. d BO > d AO and d CO =0, d xw k+1 = d Aw k+1 , d yw k+1 = d Bw k+1 , d zw k+1 = d Cwk+1 ;when d BO >1. d BO > d CO and d AO =0, d xw k+1 = d Cw k+1 , d yw k+1 = d Bw k+1 , d zw k+1 = d Aw k+1 ;when d CO >1. d CO > d AO and d BO =0, d xw k+1 = d Aw k+1 , d yw k+1 = d Cw k+1 , d zw k+1 = d Bw k+1 ;when d CO >1. d CO > d BO and d AO =0, d xw k+1 = d Bw k+1 , d yw k+1 = d Cw k+1 ,d zw k+1 = d Aw k+1 .

[0013] Specifically, in step 3, the calculation expression of the three-phase switch signal is as follows: In the formula, d A , d B and d C They are respectively the actual output inverter A-phase switch signal duty cycle, B-phase switch signal duty cycle and C-phase switch signal duty cycle; d AO , d BO and d CO They respectively represent the optimal duty cycle of the inverter A-phase switching signal, the optimal duty cycle of the B-phase switching signal, and the optimal duty cycle of the C-phase switching signal calculated by the algorithm; d Aw , d Bw and d Cw They respectively represent the suboptimal duty cycle of the inverter A-phase switching signal, the suboptimal duty cycle of the B-phase switching signal, and the suboptimal duty cycle of the C-phase switching signal calculated by the algorithm.

[0014] Specifically, the expression of the weak magnetic regulator is as follows: In the formula, i dw The d-axis reference current weak magnetic compensation value output by the weak magnetic regulator is shown in the table. e w It represents the difference between 1 and the maximum value of the optimal duty cycle of the three-phase switching signal. k p Represents the proportional gain of the PI regulator; k i Represents the integral gain of the PI regulator; 1 / s represents the integral link.

[0015] Specifically, the e w The expression is as follows: In the formula, e w Represents the difference between 1 and the maximum value of the optimal duty cycle of the three-phase switching signal; d AO , d BO and d COThey represent the optimal duty cycle of the inverter A-phase switching signal, the optimal duty cycle of the B-phase switching signal and the optimal duty cycle of the C-phase switching signal respectively; max(•) represents the maximum value function.

[0016] In order to achieve the second purpose of the present invention, the following technical solution is provided: a motor weak magnetic field region model predictive control device, used to implement the steps of the motor weak magnetic field region model predictive control method with integrated modulation function.

[0017] Compared with the prior art, the present invention has the following beneficial effects:

[0018] Based on the solution of the first cost function, the duty cycle of the three-phase switch signal of the final output is selected to ensure all voltages within the hexagon of the inverter output current, so as to give full play to the voltage output capacity of the inverter and the utilization rate of the DC bus voltage and improve the speed regulation range of the motor;

[0019] In addition, extending the voltage limit of the field weakening regulator to the voltage hexagon can reduce the operating efficiency of the motor under field weakening conditions. BRIEF DESCRIPTION OF THE DRAWINGS

[0020] Figure 1 A flow chart of the motor field weakening region model predictive control method provided in this embodiment;

[0021] Figure 2 A schematic diagram of a two-level voltage source inverter provided in this embodiment;

[0022] Figure 3 A schematic diagram of voltage vector synthesis provided in this embodiment;

[0023] Figure 4 A schematic diagram of calculating the optimal duty cycle of a three-phase switch signal provided in this embodiment;

[0024] Figure 5 A schematic diagram of the relationship between the voltage vector and the duty cycle when the optimal duty cycle of the three-phase switch signal provided in this embodiment exceeds 1;

[0025] Figure 6 A control block diagram of a magnetic field weakening regulator provided in this embodiment;

[0026] Figure 7 A control block diagram of the motor field weakening region model predictive control device provided in this embodiment;

[0027] Figure 8 Simulation waveform diagram of the traditional predictive control method and the motor weak magnetic field area model predictive control method provided in this embodiment;

[0028] Fig. 9A comparison diagram of dynamic response waveforms of the conventional predictive control method and the motor field weakening region model predictive control method provided in this embodiment;

[0029] Fig.10 A data collection comparison diagram of the traditional predictive control method and the motor weak magnetic field area model predictive control method provided in this embodiment. DETAILED DESCRIPTION

[0030] In order to make the purpose, technical scheme and advantages of the embodiments of the present invention clearer, the technical scheme in the embodiments of the present invention will be clearly and completely described below in conjunction with the drawings in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. The components of the embodiments of the present invention generally described and shown in the drawings here can be arranged and designed in various different configurations. Therefore, the following detailed description of the embodiments of the present invention provided in the drawings is not intended to limit the scope of the claimed invention, but merely represents selected embodiments of the present invention. Based on the embodiments in the present invention, all other embodiments obtained by ordinary technicians in this field without making creative work are within the scope of protection of the present invention.

[0031] like Figure 1 As shown, a motor weak magnetic field model predictive control method with integrated modulation function provided in this embodiment includes the following steps:

[0032] Step 1: Taking the duty cycle of the three-phase switch signal of the two-level voltage source three-phase inverter as the control quantity and the current output of the permanent magnet synchronous motor where the two-level voltage source three-phase inverter is located as the prediction quantity, a corresponding current prediction model is established;

[0033] Furthermore, a two-phase rotating coordinate system oriented by the rotor flux is established, and the stator voltage equation of the permanent magnet synchronous motor on the d and q axes is: (1); where u d , u q , i d and i q They represent the stator voltage vector u s With the stator current vector i s Components on the d and q axes; R s represents the stator resistance; L d , L q represents the d and q axis stator inductance; ω rrepresents the angular velocity of the rotor; ψ f Represents the flux linkage of a permanent magnet.

[0034] Since the control cycle is short enough, the equation can be discretized to obtain the d and q axis stator currents in ( k +1) T s The predicted value at time is (2); where i s ( k +1)=[ i d k+1 , i q k+1 ] T , i d k+1 and i q k+1 Respectively represent ( k +1) T s The predicted values ​​of d and q axis currents at the time; i s ( k )=[ i d k , i q k ] T , i d k and i q k Respectively kT s The d and q axis current measurement values ​​at the time; u s ( k )=[ u d k , u q k ] T , u d k , u q k Respectively kT s The d and q axis voltages applied at the moment; ; ; .

[0035] like Figure 2 As shown in the figure, it is a two-level voltage source inverter provided by this embodiment. S x It represents the switching signal of the IGBT on the three-phase (A, B and C) bridge arm of the two-level voltage source inverter. x When the upper bridge arm IGBT is turned on and the lower bridge arm IGBT is turned off, S x =1", otherwise, S x =0", where x =A, B, C. There are 8 switching combinations corresponding to 8 basic vectors in the two-level voltage source inverter: U 0(000), U 1(100), U 2(110), U 3(010), U 4(011), U 5(001), U 6(101) and U 7(111), where the numbers in brackets from left to right are S A , S B and S C The value of .

[0036] Basic voltage vector of two-level voltage source inverter U 0(000)~ U The expression of 7(111) in the two-phase stationary coordinate system (α-β coordinate system) is: (3); In the formula, U dc Indicates the DC bus voltage; i =0, 1, …, 7.

[0037] In the traditional vector synthesis process, the voltage vector in the voltage plane is synthesized by two adjacent effective vectors and a zero vector.

[0038] like Figure 3 As shown in the figure, it is a schematic diagram of voltage vector synthesis provided in this embodiment. u s1 For example, u s1 can be U 1 and U 2 Synthesis: (4); whered 1. d 2 and d 0 respectively indicates U 1. U 2 and U 0 duty cycle, where 0≤ d 1≤1、0≤ d 2≤1、0≤ d 0≤1, d 1+ d 2+ d 0=1. Here, the zero vector only uses U 0, not used U 7.

[0039] Substituting equation (3) into equation (4) yields (5);wherein, ( S A1 , S B1 , S C1 )express U 1 corresponds to the three-phase switch signal; ( S A2 , S B2 , S C2 )express U 2 corresponds to the three-phase switch signal; U The three-phase switch signals corresponding to 0 are all 0 and can be ignored here. d A , d B and d C The definition of is shown in formula (5), which represents the duty cycle of the inverter three-phase switching signal corresponding to the synthetic voltage vector. d 1+ d 2≤1 and S A1 , S B1 , S C1 , S A2 , S B2 and S C2 The value of is only 0 or 1, so 0≤ d A ≤1, 0≤ d B ≤1, 0≤ d C ≤1, and dA , d B and d C There is no mutual restriction between them, that is, d A , d B ,and d C The values ​​of do not affect each other.

[0040] From the above derivation process, it can be seen that all voltage vectors in the voltage hexagon u s The expression in the α-β coordinate system can be uniformly expressed as follows (6); Formula (6) establishes the duty cycle of the three-phase switch signal ( d A , d B , d C ) and inverter output voltage u s The mathematical relationship between Figure 3 All voltage vectors within the medium voltage hexagon can be expressed by d A , d B and d C According to the expression in formula (6), the voltage on the hexagonal boundary is calculated, that is, Figure 3 In u s1 and u s3 shown.

[0041] Combined with the basic voltage vector U 0(000)~ U 6(101) corresponding to the switching signal, it can be seen that when using adjacent non-zero vectors and zero vectors U 0 synthesizes any vector, there is always one phase switch signal is 0. Figure 3 In u s1 ~ u s3 For example, by U 1. U 2 and U 0 Synthetic Vector u s1 When the C phase switch signal is 0; U 4. U 5 and U 0 Synthetic Vector u s2When the A phase switch signal is 0; U 5. U 6 and U 0 Synthetic Vector u s3 When , the B-phase switch signal is 0. Therefore, in equation (6) d A , d B and d C At least one of them is 0.

[0042] Substituting equation (6) into equation (2), we can obtain the motor current prediction model with the duty cycle of the three-phase switch signal as the control variable: (7); where ; .

[0043] In this embodiment, a cost function is constructed based on the current prediction model in equation (7) and a prediction control algorithm is designed. The duty cycle of the three-phase switch signal can be directly calculated by the control algorithm, thereby integrating the function of the modulation algorithm (including overmodulation) into the control algorithm without executing an additional modulation algorithm.

[0044] Step 2: constructing a first cost function based on the current prediction model, wherein the first cost function is used to solve the optimal duty ratio of the three-phase switch signals of the two-level voltage source three-phase inverter;

[0045] A second cost function is constructed based on the current prediction model and in combination with preset constraints, wherein the second cost function is used to solve a suboptimal duty cycle of a three-phase switch signal of a two-level voltage source three-phase inverter;

[0046] The final empty space ratio is selected by judging the solution of the first cost function:

[0047] When any amplitude of the optimal duty cycle of the three-phase switch signal obtained by solving the first cost function is less than or equal to 1, the optimal duty cycle of the three-phase switch signal is used as the final duty cycle. Conversely, when any amplitude is greater than 1, the suboptimal duty cycle of the three-phase switch signal obtained by solving the second cost function is used as the final duty cycle.

[0048] More specifically, the current prediction model in the above equation (7) has three control variables: d A , d B and d C If this model is used, the optimization process of model predictive control needs to solve three optimal control inputs, which will increase the computational complexity of the predictive control algorithm. Therefore, it is necessary to reduce the number of optimal control inputs to be solved, thereby reducing the computational complexity.

[0049] In each control cycle, d A , d B and d C There is always one phase with a duty cycle of 0. This embodiment first assumes d C =0, and then calculate the duty ratio of phase A and phase B under this assumption according to the preset target d A1 and d B1 .

[0050] Based on the above settings and considering delay compensation, use d A1 k+1 and d B1 k+1 In the substitution d A k and d B k , and order d C k+1 = 0, construct the cost function, and then design d A1 k+1 and d B1 k+1 The optimization problem for the control input is: (8); where J 1 represents the first cost function; i d * and i q * are the reference values ​​of the d-axis and q-axis currents respectively. The reference current is obtained by converting the preset reference torque (control target).

[0051] Based on formula (8), the optimal d A1 k+1 and d B1 k+1 It can be obtained by solving the system of equations in (9).

[0052] (9).

[0053] Using formula (8) we can get d A1k+1 and d B1 k+1 Finally, we analyze the optimal duty cycle of the actual three-phase switching signal ( d AO , d BO , d CO )and( d A1 , d B1 ) between them.

[0054] In formula (6) u s It can be considered as a combination of three vectors: (10); where ,and U A , U B and U C like Figure 4 As shown in a in .

[0055] Divide the voltage plane into three sectors, such as Figure 4 As shown in a in . d A1 and d B1 calculate d A , d B and d C The analysis is as follows: 1) When d A1 ≥0 and d B1 ≥0, U A and U B The corresponding resultant vector u s Located in sector 1, such as Figure 4 As shown in b in the figure. At this time, d A = d A1 , d B = d B1 , d C =0.

[0056] 2) When d A1 <0 andd A1 < d B1 hour, U A and U B Synthetic u s like Figure 4 The c in shown.

[0057] at this time, u s is located in sector 2 and should be U B and U C Synthesis, such as Figure 4 in the c and shown.

[0058] Figure 4 In c, triangle ONP is an equilateral triangle, then | d A1 U A |=| d C U C |; quadrilateral OPRQ is a parallelogram, triangle MRQ is an equilateral triangle, then |QM|=|QR|=| d A U A |, therefore, | d B U B |= | d A1 U A |+| d B1 U B |. Because | U A |=| U B | and d A1 <0, combined with the above analysis d A =0, d B = d B1 - d A1 , d C=- d A1 .

[0059] 3) When d B1 <0 and d A1 > d B1 hour, U A and U B Synthetic u s like Figure 4 in the d shown.

[0060] at this time u s is located in sector 3 and should be U A and U C Synthesis, such as Figure 4 in the d and shown.

[0061] Figure 4 In d, triangle ONQ is an equilateral triangle, then | d B1 U B |=| d C U C |; quadrilateral OPRQ is a parallelogram and triangle MRP is an equilateral triangle, then |PM|=|NR|=| d C U C |=| d B1 U B |,| d A U A |=| d B1 U B |-| d A1 U A |. Combining the above analysis, we can get d A = d A1 - d B1 ,d B =0, d C =- d B1 .

[0062] In summary, under the assumption d C = 0 and calculate d A1 and d B1 After that, the actual three-phase duty cycle d A , d B and d C The calculation results are summarized in Table 1. Combined with Table 1, by assuming d C =0, the three-phase duty ratio of all voltage vectors in the voltage hexagon can be calculated d A , d B and d C Therefore, when using the optimization algorithm to obtain d A1 and d B1 After that, the optimal three-phase duty ratio ( d AO , d BO , d CO ).

[0063] .

[0064] When the speed of the permanent magnet synchronous motor continues to increase and exceeds a certain speed value, due to the continuous increase of the motor back electromotive force, the stator voltage in the voltage hexagon can no longer meet the control requirements of the motor. Accordingly, the optimal value obtained by using the cost function is d AO , d BO and d CO will exceed its limit value, that is d AO , d BO and d CO At least one of them is greater than 1. At this time, it is necessary to find the suboptimal three-phase duty cycle corresponding to the suboptimal voltage vector on the hexagonal boundary.

[0065] When the optimal three-phase duty cycle exceeds its limit value, the corresponding optimal voltage vector exceeds the voltage hexagon. Taking sector 1 as an example, when the optimal three-phase duty cycle exists d AO >1 and d AO > d BO When , the corresponding optimal voltage vector is as follows Figure 5 At this time, it is necessary to find the suboptimal voltage vector on the voltage hexagonal boundary closest to it, that is, Figure 5 The corresponding suboptimal three-phase duty cycle is defined as d Aw , d Bw and d Cw .for Figure 5 In the example, d Aw =1, 0≤ d Bw ≤1, d Cw =0.

[0066] Will d Aw k =1 and d Cw k =0 is substituted into formula (7), and then the optimization problem in the weak magnetic region is established based on formula (7) as follows: (11); In the formula, J 2 represents the second cost function.

[0067] In the same way, when the optimal three-phase duty cycle exceeds its limit value, the suboptimal three-phase duty cycle optimization problem under different conditions is summarized as follows: (12); where x , y and z The values ​​of are shown in Table 2.

[0068] .

[0069] Based on formula (12), the optimal d A1 k+1 and d B1 k+1 It can be obtained by solving the system of equations in (13).

[0070] (13); According to Table 2 and formula (12), the suboptimal three-phase duty cycle that meets the maximum voltage output capacity of the inverter is calculated. This not only enables the model predictive control strategy proposed in this paper to have the functions of control and overmodulation at the same time, eliminating the execution of traditional overmodulation, but also makes the actual output voltage of the inverter consistent with the voltage calculated by the model predictive control algorithm.

[0071] Step 3: convert the final duty cycle obtained in step 2, and generate a corresponding three-phase switching signal according to the conversion result;

[0072] More specifically, when the motor runs in the high-speed area, the optimal three-phase duty cycle obtained by optimization exceeds its limit value, which will cause the motor current and other state quantities to be unable to track their reference values. Therefore, it is necessary to design a weak magnetic controller to adjust the reference current so that the optimal three-phase duty cycle returns to its limit value.

[0073] In order to make the output inverter switching sequence consistent with the switching sequence output by the seven-segment space vector modulation technology, the specific conversion process is as follows: (14); where d A , d B and d C They are respectively the actual output inverter A-phase switch signal duty cycle, B-phase switch signal duty cycle and C-phase switch signal duty cycle; d AO , d BO and d CO They respectively represent the optimal duty cycle of the inverter A-phase switching signal, the optimal duty cycle of the B-phase switching signal, and the optimal duty cycle of the C-phase switching signal calculated by the algorithm; d Aw , d Bw and d Cw They respectively represent the suboptimal duty cycle of the inverter A-phase switching signal, the suboptimal duty cycle of the B-phase switching signal, and the suboptimal duty cycle of the C-phase switching signal calculated by the algorithm.

[0074] Step 4: Use the difference between value 1 and the maximum value of the optimal duty cycle of the three-phase switch signal obtained in step 2 as input to construct a weakening magnetic field regulator, and the output of the weakening magnetic field regulator is used to calculate the d-axis reference current used in the first cost function and the second cost function in the next control cycle.

[0075] like Figure 6 As shown in the figure, the weak magnetic regulator provided in this embodiment converts the difference between 1 and the maximum value of the optimal duty cycle of the three-phase switch signal into ew =1-max( d A , d B , d C ) as the input of the PI weak magnetic field regulator, and then the output of the PI weak magnetic field regulator is used as the d-axis current compensation value i dw The compensation is applied to the d-axis current reference value to achieve magnetic field weakening control.

[0076] This embodiment also provides a motor field weakening region model predictive control device, such as Figure 7 The above-mentioned steps are used to implement the motor weakening region model predictive control method provided in the above-mentioned embodiment.

[0077] To ensure that all voltages within the inverter output current hexagon can be fully utilized, the inverter's voltage output capacity and DC bus voltage utilization rate can be fully utilized, and the actual inverter output voltage can be guaranteed to be consistent with the voltage calculated by the predictive control algorithm, so that the motor can obtain a better current dynamic response speed under transient conditions.

[0078] In order to better illustrate the technical effect of the method provided in this embodiment, feasibility verification is performed using simulation and experimental data.

[0079] The motor parameters of the 3.1kW permanent magnet synchronous motor system provided in this embodiment are shown in Table 3. The control algorithm of the experimental drive system is executed in a control board with DSP (TMS320F28379D) and FPGA (5CEFA7F27I7N) as the core. The sampling frequency and control frequency in the experiment are both 10kHz.

[0080] .

[0081] In order to verify the feasibility and effectiveness of the method proposed in this paper, Matlab / Simulink is used to monitor the state changes of the motor when the speed ramps up and enters high-speed working conditions under different model predictive control methods.

[0082] like Figure 8 As shown, Figure 8 a is the motor speed under the traditional model predictive control method n , d, q axis current i d , i q , A phase current i A and α, β axis voltage u α , u βThe simulation waveform.

[0083] Figure 8 b in the figure is the motor speed under the control of the method provided in this embodiment n , d, q axis current i d , i q , A phase current i A and α, β axis voltage u α , u β The simulation waveform.

[0084] And taking the q-axis current failing to track its reference value as the standard for judging the motor speed regulation limit, it can be seen that the method provided in this embodiment can make the inverter output all voltages within the entire voltage hexagon, while the traditional model predictive control method based on space vector modulation technology can only make the inverter output the voltage within the inscribed circle of the voltage hexagon. Therefore, the method provided in this embodiment can give full play to the voltage output capacity of the inverter.

[0085] At the same time, under the torque condition of 15 N·m, the speed regulation range of the traditional model predictive control is 0~2568r / min, and the speed regulation range of the method provided in this embodiment is 0~2752r / min. Therefore, compared with the traditional model predictive control method using the voltage hexagonal inscribed circle as the voltage limit, the method provided in this embodiment can further improve the speed regulation range of the motor.

[0086] like Fig. 9 As shown in the figure, in the experiment, when the motor runs at 200r / min and 2000r / min, the reference torque suddenly changes from 2N·m to 15N·m.

[0087] in, Fig. 9 a in the figure is the dynamic response waveform of the d and q axis currents when the reference torque suddenly changes under the traditional model predictive control method in the high-speed zone.

[0088] Fig. 9 b in the figure is the dynamic response waveform of the d-axis and q-axis currents when the reference torque suddenly changes according to the method provided in this embodiment.

[0089] It can be seen from the figure that when the speed is 200r / min, the q-axis current response speeds of the two model predictive control methods are the same, but when the speed is 2000r / min, the q-axis current response speed controlled by the method provided in this embodiment is faster than the q-axis current response speed of the traditional model predictive control method, and the d-axis current can better track the given value.

[0090] like Fig.10As shown in the figure, the experimental screenshot of monitoring the input power and output power of the inverter and motor when the motor speed is 3000r / min and the motor output torque is 10N·m. Based on the inverter output power and motor output power in the screenshot, the motor efficiency can be calculated. η .

[0091] in, Fig.10 Where a is the motor input and output power under traditional model predictive control.

[0092] Fig.10 Here, b represents the input and output power of the motor under the control of the method provided in this embodiment.

[0093] Depend on Fig.10 a and Fig.10 From b, it can be seen that the motor operating efficiency under the control of the traditional model predictive control method is 89.69%, and the motor operating efficiency under the control of the method provided in this embodiment is 91.60%. Therefore, compared with the traditional model predictive control method, the method provided in this embodiment can enable the motor to achieve higher efficiency when operating in a weak magnetic condition.

[0094] In addition, the terms "upper", "lower", "inner", "outer", "front", "rear" are used for descriptive purposes only and should not be understood as indicating or implying relative importance. Unless otherwise specifically stated, the relative steps, numerical expressions and values ​​of the components and steps described in these embodiments do not limit the scope of the present invention.

[0095] Of course, the above description is only a specific embodiment of the present invention and is not intended to limit the scope of implementation of the present invention. All equivalent changes or modifications made according to the structure, characteristics and principles described in the patent application scope of the present invention should be included in the patent application scope of the present invention.

[0096] Finally, it should be noted that the above-described embodiments are only specific implementations of the present invention, which are used to illustrate the technical solutions of the present invention, rather than to limit them. The protection scope of the present invention is not limited thereto. Although the present invention is described in detail with reference to the above-described embodiments, ordinary technicians in the field should understand that any technician familiar with the technical field can still modify the technical solutions recorded in the above-described embodiments within the technical scope disclosed by the present invention, or can easily think of changes, or make equivalent replacements for some of the technical features therein; and these modifications, changes or replacements do not make the essence of the corresponding technical solutions deviate from the spirit and scope of the technical solutions of the embodiments of the present invention, and should be included in the protection scope of the present invention. Therefore, the protection scope of the present invention shall be subject to the protection scope of the claims.

Claims

1. A motor weak magnetic field model predictive control method with integrated modulation function, characterized in that: The following steps are involved: Step 1: Taking the duty cycle of the three-phase switch signal of the two-level voltage source three-phase inverter as the control quantity and the current output of the permanent magnet synchronous motor where the two-level voltage source three-phase inverter is located as the prediction quantity, a corresponding current prediction model is established; Step 2: Constructing a first cost function based on the current prediction model, the first cost function is used to solve the optimal duty cycle of the three-phase switch signal of the two-level voltage source three-phase inverter; Constructing a second cost function based on the current prediction model and in combination with preset constraints, the second cost function is used to solve the three-phase switch signal of the two-level voltage source three-phase inverter Suboptimal duty cycle; by judging the solution result of the first cost function, the final duty cycle is selected: when any amplitude of the optimal duty cycle of the three-phase switch signal obtained by solving the first cost function is less than or equal to 1, the optimal duty cycle of the three-phase switch signal is used as the final duty cycle; conversely, when any amplitude is greater than 1, the suboptimal duty cycle of the three-phase switch signal obtained by solving the second cost function is used as the final duty cycle; step 3, converting the final duty cycle obtained in step 2, and generating a corresponding three-phase switch signal according to the conversion result; step 4, using the difference between the value 1 and the maximum value of the optimal duty cycle of the three-phase switch signal obtained in step 2 as the input to construct a weak magnetic regulator, the output of the weak magnetic regulator is used to calculate the d-axis reference current used in the first cost function and the second cost function in the next control cycle.

2. The motor field weakening region model predictive control method with integrated modulation function according to claim 1, characterized in that: The expression of the current prediction model is as follows: In the formula, i d k+1 and i q k+1 Respectively represent ( k +1) T s The predicted values ​​of d and q axis currents at time t, i d k and i q k Respectively kT s The d and q axis current measurement values ​​at the time; T s Indicates the control cycle; R s represents the stator resistance; L d , L q Respectively represent the d-axis and q-axis stator inductances; ω r k represents the angular velocity of the rotor; ψ f represents the flux linkage of permanent magnet; V dc Indicates the DC bus voltage of the inverter; d A k , d B k and d C k Respectively kT s The duty cycle of the inverter A phase switching signal, the duty cycle of the inverter B phase switching signal and the duty cycle of the inverter C phase switching signal at the moment; θ r Indicates the electrical angle of the motor rotor.

3. The motor field weakening region model predictive control method with integrated modulation function according to claim 1, characterized in that: The expression of the first cost function is as follows: In the formula, J 1 represents the first cost function; i d * and i q * are the reference values ​​of d-axis and q-axis currents respectively; i d k+2 and i q k+2 Respectively represent ( k +2) T s The predicted values ​​of d and q axis currents at the time; d A1 k+1 and d B1 k+1 Respectively represent ( k +1) T s The duty cycle of the inverter A phase switching signal and the duty cycle of the inverter B phase switching signal at the moment.

4. The motor field weakening region model predictive control method with integrated modulation function according to claim 1, characterized in that: The expression of the second cost function is as follows: In the formula, J 2 represents the second cost function, d xw k+1 express( k +1) T s Inverter at the moment x Suboptimal duty cycle of the phase switching signals, d yw k+1 express( k +1) T s Inverter at the moment y Suboptimal duty cycle of the phase switching signals, d zw k+1 express( k +1) T s Inverter at the moment z Suboptimal duty cycle of the phase switching signals.

5. The motor field weakening region model predictive control method with integrated modulation function according to claim 1, characterized in that: In step 3, the calculation expression of the three-phase switch signal is as follows: In the formula, d A , d B and d C They are respectively the actual output inverter A-phase switch signal duty cycle, B-phase switch signal duty cycle and C-phase switch signal duty cycle; d AO , d BO and d CO They respectively represent the optimal duty cycle of the inverter A-phase switching signal, the optimal duty cycle of the B-phase switching signal, and the optimal duty cycle of the C-phase switching signal calculated by the algorithm; d Aw , d Bw and d Cw They respectively represent the suboptimal duty cycle of the inverter A-phase switching signal, the suboptimal duty cycle of the B-phase switching signal, and the suboptimal duty cycle of the C-phase switching signal calculated by the algorithm.

6. The motor field weakening model predictive control method with integrated modulation function according to claim 1, characterized in that: The expression of the field weakening regulator is as follows: In the formula, i dw The d-axis reference current weak magnetic compensation value output by the weak magnetic regulator is shown in the table. e w It represents the difference between 1 and the maximum value of the optimal duty cycle of the three-phase switching signal. k p Represents the proportional gain of the PI regulator; k i Represents the integral gain of the PI regulator; 1 / s represents the integral link.

7. The motor field weakening region model predictive control method with integrated modulation function according to claim 6, characterized in that: Said e w The expression is as follows: In the formula, e w Represents the difference between 1 and the maximum value of the optimal duty cycle of the three-phase switching signal; d AO , d BO and d CO They represent the optimal duty cycle of the inverter A-phase switching signal, the optimal duty cycle of the B-phase switching signal and the optimal duty cycle of the C-phase switching signal respectively; max(•) represents the maximum value function.

8. A motor field weakening region model predictive control device, characterized in that: Steps for implementing the motor weak magnetic field model predictive control method with integrated modulation function as described in any one of claims 1 to 7.

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