DAB converter modulation method with optimal turn-off current

Abstract

The invention discloses a DAB converter modulation method with optimal turn-off current, and the method specifically comprises the steps: solving a TPS duty ratio with the optimal turn-off current through employing a carlo demand-Kuhn-tuck KKT condition in order to achieve the minimization of the switching loss when a DAB works, and when the given current is in different ranges, thereby obtaining an optimal phase shift duty ratio value. According to the method, the analytic solution of the TPS duty ratio in the DAB converter is solved according to the optimization target of the optimal turn-off current, the TPS duty ratio is calculated in real time according to the feedback state and the controller output in the dynamic operation process of the system, the turn-off current of the power device is minimized on the basis of ensuring that the power device achieves ZVS, low on-state loss is kept, and the power device is ensured to be stable. Therefore, the DAB converter can maintain low switching loss and high transmission efficiency in a full working range.

Description

Technical Field

[0001] The invention belongs to the technical field of power electronic device modulation, and in particular relates to a DAB converter modulation method with optimal off-current. Background Art

[0002] The Dual Active Bridge (DAB) converter is an advanced power electronic device that uses power semiconductor switches to control the flow of electrical energy, achieving bidirectional and efficient energy conversion between the DC ports on both sides. The DAB converter has the characteristics of high power density, simple structure, input and output isolation, and flexible control methods. It has been widely used in the fields of electric vehicle and power grid interconnection, energy storage systems, power electronic transformers, energy routers, etc.

[0003] In the DAB converter, the active full bridges on both sides apply switching voltage to the series inductor and transformer, and adjust the transmission power by setting the phase shift angle between the bridges. This is phase shift modulation; common modulation methods include single phase shift, double phase shift and triple phase shift (TPS) modulation and other technologies. Among them, the TPS modulation method stands out because it can control three phase shift angles, making other modulation technologies its subset. TPS modulation provides a strategy to improve conversion efficiency by calculating the best combination of three phase shift angles. Existing strategies usually take current stress or root mean square current as the optimization target on the basis of widening the zero voltage switching (ZeroVoltage Switching, ZVS) range as much as possible, that is, to achieve minimum conduction loss on the basis of ZVS. ZVS operation eliminates the turn-on loss of power devices and alleviates the voltage ringing problem caused by the parasitic capacitance of the switch tube. At this time, the power device loss mainly includes turn-off loss and conduction loss. However, with the increase of the switching frequency of the DAB converter, the turn-off loss has become one of the important reasons restricting the frequency increase and causing power loss. However, existing research has not conducted an in-depth analysis on this. The conversion efficiency of the DAB converter has not yet reached the theoretical optimal state, and its overall performance still has great potential for improvement. Summary of the invention

[0004] The object of the present invention is to provide a modulation method for a DAB converter with an optimal turn-off current, so as to achieve the operation goal of maintaining low switching loss and high transmission efficiency of the DAB converter in the full working range.

[0005] The technical solution adopted by the present invention is a modulation method for a DAB converter with an optimal turn-off current, specifically: in order to minimize the switching loss of the DAB during operation, solutions are sought for Region 1 and Region 4. In Region 1, the phase-shifted duty ratios satisfy the following relationships: -1 < D0 < 0; 0 < D1 < 1; 0 < D2 < 1; D1 < D0 + D2. In Region 4, the phase-shifted duty ratios satisfy the following relationships: 0 < D0 < D1 < 1; 0 < D2 < 1; 1 < D0 + D2. Define D0, D1, and D2 as phase-shift ratios. When the given current is within different ranges, the Karush-Kuhn-Tucker (KKT) conditions are used to solve for the TPS duty ratio with the optimal turn-off current, and then the optimal phase-shifted duty ratio values can be obtained.

[0006] The features of the present invention also lie in that

[0007] The KKT condition method has the following standard form, as shown in Equation (12);

[0008]

[0009] where L is the Lagrangian function; x is the independent variable; h is the equality constraint condition to be satisfied; g i (i = 1, 2,...) are the inequality condition to be satisfied.

[0010] To find the optimal solution of the phase-shift ratio, the KKT conditions within Region 1 are constructed as shown in the following Equation (13);

[0011]

[0012] In the formula, λ i , μ are KKT operators; f(D) is the optimization objective, with the minimum turn-off current as the target; h(D) is the equality constraint condition; g i(i＝1,2,3…) (D) is the inequality constraint condition; define the solution D * to represent the optimal duty ratio that satisfies the conditions; L(D, λ i , μ) is the Lagrangian function in the KKT conditions; calculate λ i , μ, and it should satisfy that the partial derivative of L with respect to D is equal to 0, h(D * ) = 0, g ii (D * ) ≤ 0, λ i > 0, λ i g i (D * ) = 0; K is a coefficient; D is the phase-shifted duty ratio matrix; m is the converter voltage gain; n is the turns ratio of the high-frequency transformer coil; V1 is the DC voltage on the primary side of the converter;

[0013] Solving the above formula gives a situation with only one solution, that is, when λ 1,3 ≥ 0; λ2,4,5 =0; μ≤0; g 1,3 (D * )=0, the coefficients are shown in the following formula (14);

[0014]

[0015] Substituting the constraint condition h(D) into equation (14), the optimal solution is shown in equation (15);

[0016]

[0017] At this time, the given output current value range is shown in formula (16);

[0018]

[0019] in, is the average current expected to be output by the converter during one switching cycle.

[0020] The KKT conditions in the construction working area 4 are shown in formula (17);

[0021]

[0022] Solving equation (17) yields four possible solutions:

[0023] Case 1: When λ i =0; μ≤0; g2(D * )≤0, the coefficients are shown in formula (18);

[0024]

[0025] Substituting the constraint condition h(D) into equation (18), the optimal solution is shown in equation (19);

[0026]

[0027] Combining equation (18) and equation (19), the range of the given output current can be obtained as shown in equation (20);

[0028]

[0029] Case 2: When λ 1,3,4 =0; λ2≥0; μ≤0; g2(D*)=0, the coefficients are shown in formula (21);

[0030]

[0031] Substituting the constraint condition h(D) into equation (21), the optimal solution is shown in equation (22);

[0032]

[0033] Satisfy the constraints: g1(D * )≤0, that is, m 3 +m 2 When +m-1≥0, combining equation (21) and equation (22), the range of the given output current value can be obtained as shown in equation (23);

[0034]

[0035] Satisfy the constraints: g3(D * )≤0, that is, m 3 +m 2 When +m-1<0, combining equation (21) and equation (22), the given output current value range can be obtained as shown in equation (24);

[0036]

[0037] Case 3: When λ 3,4,5… =0;λ 1,2 ≥0; μ≤0; g 1,2 (D * )=0;g3(D * )≤0, the coefficients are shown in formula (25);

[0038]

[0039] Substituting the constraint condition h(D) into equation (25), the optimal solution is shown in equation (26);

[0040]

[0041] In m 3 +m 2 Under the premise of +m-1≥0, combining equation (25) and equation (26), the range of the given output current value can be obtained as shown in equation (27);

[0042]

[0043] Case 4: When λ 1,4,5… =0;λ 2,3 ≥0; μ≤0; g 2,3 (D * )=0;D * When ∈D4, the coefficients are shown in formula (28);

[0044]

[0045] Substituting the constraint condition h(D) into equation (28), the optimal solution is shown in equation (29);

[0046]

[0047] In m 3 +m 2 Under the premise of +m-1<0, combining equation (28) and equation (29), the range of the given output current value can be obtained as shown in equation (30);

[0048]

[0049] From all the situations analyzed above, it can be seen that when the given current is within different ranges, the optimal phase shift duty cycle value can be obtained after KKT condition calculation.

[0050] The beneficial effects of the present invention are:

[0051] The modulation method of the DAB converter with optimal off-current of the present invention solves the optimization target of the optimal off-current to obtain the analytical solution of the TPS duty cycle in the DAB converter based on the analysis of the system model and the loss problem. During the dynamic operation of the system, the TPS duty cycle is calculated in real time according to the feedback state and the controller output. The off-current of the power device is minimized on the basis of ensuring that the power device achieves ZVS, and the on-state loss is maintained low, thereby achieving the operation goal of the DAB converter to maintain low switching loss and high transmission efficiency in the full working range. BRIEF DESCRIPTION OF THE DRAWINGS

[0052] Figure 1 A schematic diagram of the topological structure of the main circuit of a DAB converter disclosed in the prior art;

[0053] Figure 2 A voltage and current diagram of a DAB converter disclosed in the prior art when operating in region 1;

[0054] Figure 3 A voltage and current diagram of a DAB converter disclosed in the prior art when operating in region 4;

[0055] Figure 4 The figure is a flow chart of calculating the duty cycle of the TPS with the optimal off current of the present invention;

[0056] Figure 5 A block diagram of a DAB converter modulation method with an optimal off-current according to the present invention;

[0057] Figure 6 The simulation waveform diagram (I) of the DAB converter modulation method with optimal off current of the present invention;

[0058] Figure 7The simulation waveform diagram (II) of the DAB converter modulation method with optimal off-current of the present invention;

[0059] Figure 8 The simulation waveform diagram (III) of the DAB converter modulation method with optimal off current of the present invention;

[0060] Fig. 9 FIG4 is a simulation waveform diagram of the DAB converter modulation method with optimal off-current according to the present invention. DETAILED DESCRIPTION

[0061] The present invention is described in detail below with reference to the accompanying drawings and specific embodiments.

[0062] The schematic diagram of the main circuit topology of the DAB converter disclosed in the prior art is as follows: Figure 1 As shown in the figure, taking power MOSFET as the power switch tube as an example, the main circuit topology of the DAB converter consists of two groups of completely symmetrical bridge conversion units, an inductor and a high-frequency transformer. By adjusting the control method, the bidirectional flow of energy under different DC voltage levels can be achieved. In the main circuit, the inductor L r The inductance value is the sum of the leakage inductance of the high-frequency transformer and the inductance value of the external inductance. Taking the high-frequency transformer as the dividing line, the circuit composed of the various devices connected to the primary side of the transformer is called the primary side, and the circuit composed of the various devices connected to the secondary side of the transformer is called the secondary side. The primary side DC voltage V1 is connected in parallel with both ends of the filter capacitor C1, and the secondary side DC voltage V2 is connected in parallel with both ends of the filter capacitor C2. The first bridge is composed of power switch tubes Q1 and Q2, the second bridge is composed of power switch tubes Q3 and Q4, the third bridge is composed of power switch tubes Q5 and Q6, and the fourth bridge is composed of power switch tubes Q7 and Q8. The collector of the power switch tube Q1 is connected to the collector of the power switch tube Q3, and is also connected to the positive pole of the primary side DC voltage V1. The emitter of the power switch tube Q2 is connected to the emitter of the power switch tube Q4, and is also connected to the negative pole of the primary side DC voltage V1. The collector of the power switch tube Q5 is connected to the collector of the power switch tube Q7, and is also connected to the positive pole of the secondary side DC voltage V2. The emitter of the power switch tube Q6 is connected to the emitter of the power switch Q8, and is also connected to the negative electrode of the secondary DC voltage V2. The emitter of the power switch tube Q1 is connected to the collector of the power switch tube Q2, and is also connected to one end of the inductor, and the other end of the inductor is connected to one end of the primary side of the high-frequency transformer. The emitter of the power switch tube Q3 is connected to the collector of the power switch tube Q4, and is also connected to the other end of the primary side of the high-frequency transformer. The emitter of the power switch tube Q5 is connected to the collector of the power switch tube Q6, and the emitter of the power switch tube Q7 is connected to the collector of the power switch tube Q8. The same-name end of the secondary side of the high-frequency transformer is connected to the emitter of the power switch tube Q5, and the opposite-name end is connected to the emitter of the power switch tube Q7.

[0063] The duty cycle of the driving signals of all power switches Q1 to Q8 is 50%, and the frequency is fixed at f s The PWM signal of the two power switch tubes in each bridge arm is a complementary signal. In order to prevent the bridge arm from being directly connected, a certain dead time should be added to the complementary drive signal to prevent the bridge arm from being short-circuited or even causing damage to the converter. Define D0, D1, and D2 as shift ratios, where the shift ratio between the drive signals of the power switch tube Q1 and the power switch tube Q3 is D1, the shift ratio between the drive signals of the power switch tube Q1 and the power switch tube Q5 is D0, and the shift ratio between the drive signals of the power switch tube Q5 and the power switch tube Q7 is D2. Taking D1 as an example, taking the drive signal of the power switch tube Q1 as the reference, the lag time of the drive signal of the power switch tube Q3 is D1 multiplied by half of the switching period. The phase shift ratio D0 of the TPS modulation method ranges from -1 to 1, the value range of D1 is 0 to 1, and the value range of D2 is 0 to 1; D0, D1, and D2 are the phase shift duty ratios calculated after the optimal phase shift ratio; by changing the size of D0, D1, and D2, the bidirectional flow of power can be achieved and the size of the power output can be adjusted.

[0064] Taking the step-down mode from left to right as an example, according to the total voltage V of the primary side of the high-frequency transformer and the external inductor b1 and the secondary voltage of the high-frequency transformer V b2 The buck mode can work in various regions with different combinations of , and a large number of studies have shown that DAB converters can achieve better performance when working in regions 1, 3, and 4. The method used in the present invention has a solution only in regions 1 and 4. At this time, the relationship between the voltage at both ends of the high-frequency transformer and the primary current is as follows: Figure 2 and Figure 3 As shown. k0 、i k1 、i k2 、i k3 is the current value of each inflection point of the primary current of the high-frequency transformer, recorded as matrix i k ; The ratio of the time between two adjacent inflection point currents to half a switching period is τ0, τ1, τ2, τ3, respectively, recorded as matrix τ.

[0065] Example 1

[0066] The voltage and current diagram of the DAB converter disclosed in the prior art when working in region 1, in region 1, the phase shift duty ratio satisfies the following relationship: <D0<0;0<D1<1;0<D2<1;D1<D0+D2。

[0067] like Figure 2 As shown, according to Figure 2 The available displacement is compared with τ xThe matrix relationship between (x=0,1,2,3) is shown in formula (1);

[0068]

[0069] In the formula, the ratio of the time between two adjacent inflection point currents to half a switching period is τ0, τ1, τ2, τ3, respectively, recorded as matrix τ;

[0070] Reference Figure 1 , V1, V2 are the DC voltages on both sides of the converter, n is the turns ratio of the high-frequency transformer coil, and the converter voltage gain m is defined as shown in formula (2);

[0071]

[0072] Combining the voltage-current relationship of the inductor, we can get i kx (x=0,1,2,3) and τ x The matrix relationship between (x=0,1,2,3) is shown in formula (3);

[0073]

[0074] Among them, f s is the driving signal frequency, L r is the sum of the leakage inductance of the high-frequency transformer T1 and the external inductance, i k0 、i k1 、i k2 、i k3 is the current value of each inflection point of the primary current of the high-frequency transformer, recorded as matrix i k ;

[0075] Definition coefficient K = V1 / 2f s L r , since the DAB converter is a phase-shift operation, and the duty cycle of the switch signal of each power switch tube is 50%, the turn-on current and turn-off current of each switch tube are equal. Combining equations (1) and (3), the turn-on current and turn-off current can be obtained, as shown in equation (4);

[0076]

[0077] Among them, I Q1(on) and I Q3(on) is the turn-on current of power switch tubes Q1 and Q3, I′ Q5(on) and I′ Q7(on) I is the turn-on current of the power switch tubes Q5 and Q7 converted to the primary side; Q1(off) and I Q3(off) is the turn-off current of power switch tubes Q1 and Q3, I' Q5(off) and I' Q7(off)It is the turn-off current of power switch tubes Q5 and Q7 converted to the primary side.

[0078] Reference Figure 2 , the average value of the converter output current in a switching cycle can be obtained as shown in formula (5);

[0079]

[0080] Substituting equation (3) into equation (5), the matrix form of the average output current in region 1 is shown in equation (6);

[0081]

[0082] Among them, the phase shift duty cycle matrix D is defined as [D0 D1 D2] T Represents the triple phase-shift duty cycle. is the average current output by the converter in one switching cycle;

[0083] The prior art discloses a voltage and current diagram of a DAB converter operating in region 4. In region 4, the phase shift duty ratio satisfies the following relationship: <D0<D1<1;0<D2<1;1<D0+D2。

[0084] Similarly, according to Figure 3 , we can get the shift ratio compared with τ x The matrix relationship between (x=0,1,2,3) is shown in formula (7);

[0085]

[0086] Combining the voltage-current relationship of the inductor, we can get i kx (x=0,1,2,3) and τ x The matrix relationship between (x=0,1,2,3) is shown in formula (8);

[0087]

[0088] Combining equations (7) and (8), we can get the turn-on current and turn-off current as shown in equation (9);

[0089]

[0090] Reference Figure 3 , the average value of the converter output current in one switching cycle is shown in formula (10);

[0091]

[0092] Substituting equation (8) into equation (10), the matrix form of the average output current in region 4 is obtained as shown in equation (11).

[0093]

[0094] like Figure 4 As shown, the DAB converter modulation method with optimal off-current of the present invention is specifically as follows:

[0095] Step 1: In order to minimize the switching loss of the DAB during operation, the Karush-Kuhn-Tucker (KKT) condition is used to solve the TPS duty cycle with the optimal off current;

[0096] That is, the turn-off current of the switch tube is minimized under the premise of ensuring that all power switch tubes achieve full-range soft switching, so as to further reduce the switching loss and improve the efficiency of the DAB converter. The KKT condition method has the following standard form as shown in formula (12);

[0097]

[0098] Where L is the Lagrangian function; x is the independent variable; h is the equality constraint that needs to be satisfied; g i (i=1,2,…) is the inequality condition that needs to be satisfied. In order to find the optimal solution of the displacement ratio, the KKT condition in the construction area 1 is shown in formula (13), where λ i , μ is the KKT operator.

[0099]

[0100] In the formula, f(D) is the optimization target, with the minimum off-current as the target, which is equal to the sum of the off-currents of the switch tubes Q1, Q3, Q5, and Q7; h(D) is the equality constraint, which is used to constrain the output current to be equal to the given value; g i(i＝1,2,3…) (D) is an inequality constraint, which is used to limit the range of duty cycle. Solution D is defined as * represents the optimal duty cycle that meets the conditions. L(D,λ i ,μ) is the Lagrangian function in the KKT condition. Calculate λ i , μ, the partial derivative of L with respect to D should be equal to 0, h(D * )=0,g i (D * )≤0,λ i >0,λ i g i (D * )=0.

[0101] Solving equation (13) yields a situation where there is only one solution, namely when λ 1,3 ≥0;λ 2,4,5 =0; μ≤0; g 1,3 (D *)=0, the coefficients are shown in formula (14);

[0102]

[0103] Substituting the constraint condition h(D) into equation (14), the optimal solution is shown in equation (15);

[0104]

[0105] At this time, the given output current value range is shown in formula (16);

[0106]

[0107] in, is the average current expected to be output by the converter in one switching cycle;

[0108] Step 2, construct the KKT conditions in the working area 4 as shown in formula (17);

[0109]

[0110] Where f(D) is the optimization target, with the minimum shutdown current as the target; h(D) is the equality constraint, which is used to constrain the output current to be equal to the given value; g i(i＝1,2,3…) (D) is an inequality constraint, which is used to limit the range of duty cycle. Solution D is defined as * represents the optimal duty cycle that meets the conditions. L(D,λ i ,μ) is the Lagrangian function in the KKT condition. Calculate λ i , μ, the partial derivative of L with respect to D should be equal to 0, h(D * )=0,g i (D * )≤0,λ i >0,λ i g i (D * )=0.

[0111] Solving equation (17) shows that there are four possible solutions, namely, case 1 to case 4;

[0112] Case 1: When λ i =0; μ≤0; g2(D * )≤0, the coefficients are shown in formula (18);

[0113]

[0114] Substituting the constraint condition h(D) into equation (18), the optimal solution is shown in equation (19);

[0115]

[0116] Combining equation (18) with equation (19), the range of values ​​of the given output current can be obtained as shown in equation (20).

[0117]

[0118] Case 2: When λ 1,3,4 =0; λ2≥0; μ≤0; g2(D*)=0, the coefficients are shown in formula (21);

[0119]

[0120] Substituting the constraint condition h(D) into equation (21), the optimal solution is shown in equation (22);

[0121]

[0122] Satisfy the constraints: g1(D * )≤0, that is, m 3 +m 2 When +m-1≥0, combining equation (21) and equation (22), the range of the given output current value can be obtained as shown in equation (23);

[0123]

[0124] Satisfy the constraints: g3(D * )≤0, that is, m 3 +m 2 When +m-1<0, combining equation (21) and equation (22), the given output current value range can be obtained as shown in equation (24);

[0125]

[0126] Case 3: When λ 3,4,5… =0;λ 1,2 ≥0; μ≤0; g 1,2 (D * )=0;g3(D * )≤0, the coefficients are shown in formula (25);

[0127]

[0128] Substituting the constraint condition h(D) into equation (25), the optimal solution is shown in equation (26);

[0129]

[0130] In m 3 +m 2Under the premise of +m-1≥0, combining equation (25) and equation (26), the range of the given output current value can be obtained as shown in equation (27);

[0131]

[0132] Case 4: When λ 1,4,5… =0;λ 2,3 ≥0; μ≤0; g 2,3 (D * )=0;D * When ∈D4, the coefficients are shown in formula (28);

[0133]

[0134] Substituting the constraint condition h(D) into equation (28), the optimal solution is shown in equation (29);

[0135]

[0136] In m 3 +m 2 Under the premise of +m-1<0, combining equation (28) and equation (29), the range of the given output current value can be obtained as shown in equation (30);

[0137]

[0138] From all the situations analyzed above, it can be seen that when the given current is within different ranges, the optimal phase shift duty cycle value can be obtained after KKT condition calculation.

[0139] Therefore, for the calculation of the TPS duty cycle with the optimal off-current, it is first necessary to sample the input voltage V1 and the output voltage V2, combined with the switching frequency f in the main circuit s , external inductor L r The coefficient K and the voltage gain m of the DAB converter are calculated by the ratio of the primary and secondary coils of the high-frequency transformer to n. The expected average output current of the PI control module is determined by the calculated coefficient K and the voltage gain m of the DAB converter. If the expected output current average value is In the region shown in equation (16), the phase shift duty ratio is calculated by equation (14) and equation (15); if the expected output current average value is In the region shown in equation (20), the phase shift duty ratio is calculated by equation (18) and equation (19); when m 3 +m 2 +m-1≥0, if the expected output current average value In the region shown in equation (27), the phase shift duty ratio is calculated by equation (25) and equation (26). If the expected output current average value is In the region shown in equation (24), the phase shift duty ratio is calculated by equation (21) and equation (22); when m 3 +m 2 When +m-1<0, if the expected output current average value In the region shown in equation (30), the phase shift duty ratio is calculated by equation (28) and equation (29). If the expected output current average value is In the region shown in equation (23), the phase shift duty ratio is calculated by equation (21) and equation (22). The output of the optimal phase shift duty ratio calculation module is the optimal phase shift duty ratio D0, D1, D2 obtained by the above calculation.

[0140] Example 2

[0141] like Figure 5 As shown in FIG. 1 , the implementation block diagram of the DAB converter modulation method with optimal off-current of this embodiment includes a PI control link, an optimal shift phase calculation link, a voltage sampling link, a pulse generator link, and a DAB converter. Given a voltage Subtract the output voltage V2 sampled by the voltage sampling link, and its error is input into the PI control link. The output of the PI control link is the average current value expected to be output by the DAB converter. As an input of the optimal phase shift calculation link, the other inputs of the optimal phase shift calculation link are the sampled input voltage V1 and output voltage V2. The optimal phase shift calculation link outputs the phase shift duty ratios D0, D1, and D2, which are used as inputs of the pulse generator link. The pulse generator link outputs the switching signals g1 to g8 for controlling the switch tubes Q1 to Q8 of the DAB converter. The specific control process of each link is as follows:

[0142] PI control link:

[0143] The PI control link consists of a proportional controller (P) and an integral controller (I). The proportional controller is used for fast response, and the integral controller is used to eliminate system static errors and improve stability. By adjusting the proportional coefficient and the integral time constant, the performance of the system can be optimized to achieve the best control effect. The output of the PI control link is the average current value of the DAB converter's desired output.

[0144] Voltage sampling link:

[0145] Through the hardware sampling and conditioning circuit, the input voltage V1 and output voltage V2 of the DAB converter are sampled and obtained, and used as the input of the optimal shift ratio calculation link and the feedback of the closed-loop control system.

[0146] Optimal shift ratio calculation process:

[0147] The sampled input voltage V1 and output voltage V2 are combined with the switching frequency f in the main circuit. s , external inductor L r And the turns ratio n of the primary and secondary coils of the high-frequency transformer, through Figure 4 By performing calculations according to the calculation block diagram shown in the figure, the optimal shift ratios D0, D1, and D2 under the current working state can be obtained.

[0148] Pulse generator link:

[0149] The optimal shift phases D0, D1, and D2 calculated in the optimal shift phase calculation link are converted into switch signals g1 to g8 of power switch tubes Q1 to Q8. The switch signal g1 of power switch tube Q1 is used as a reference. The switch signal g1 is a square wave signal with a duty cycle of 50%. The switch signal g3 of power switch tube Q3 lags behind the switch signal g1 by D1 switching cycles. The switch signal g5 of power switch tube Q5 lags behind the switch signal g1 by D0 switching cycles. The switch signal g7 of power switch tube Q7 lags behind the switch signal g1 by D0+D2 switching cycles. The switch signals of the upper and lower switch tubes on the same bridge are complementary, that is, the switch signal g2 of Q2 is complementary to the switch signal g1 of Q1, the switch signal g4 of Q4 is complementary to the switch signal g3 of Q3, the switch signal g6 of Q6 is complementary to the switch signal g5 of Q5, and the switch signal g8 of Q8 is complementary to the switch signal g7 of Q7. At this point, the power switch tube Q 1～8 The switching signals of the DAB converter have been determined and can work normally.

[0150] In the above modulation method of the DAB converter based on the optimal shutdown current, the optimal phase shift ratio calculation link is designed. During the dynamic operation of the system, each phase shift duty ratio is calculated in real time according to the feedback state and the controller output. The shutdown current of the power device is minimized on the basis of ensuring that the power device achieves ZVS, and the on-state loss is maintained low, thereby achieving the operating goal of the DAB converter to maintain low switching loss and high transmission efficiency in the full operating range.

[0151] Example 3

[0152] like Figure 6 As shown, the simulation waveform diagram of the DAB converter modulation method with the optimal off current of this embodiment; the switch tube of the DAB converter in the simulation is an ideal switch; the capacitance values ​​of the filter capacitors C1 and C2 on the DC side are both 470μF; the inductor L rThe inductance value is 15μH; the turns ratio n of the primary and secondary coils of the high-frequency transformer is 1:1; the switching frequency f s The frequency is 150kHz. Set the input voltage V1 to 500V, the output voltage V2 to 300V, and the output current to 0 to 25A.

[0153] According to the DAB converter parameters in the simulation, the voltage transformation ratio m=0.6 is calculated, which satisfies m 3 +m 2 +m-1≥0. According to Figure 4 It can be obtained that as the output current of the DAB converter increases, the optimal duty cycle passes through four situations: area 1 case 1, area 4 case 3, area 4 case 2, and area 4 case 1. The corresponding high-frequency transformer primary side and the external inductor L r The total voltage V b1 , high frequency transformer secondary voltage V b2 and the inductor current i Lr like Figure 6-Figure 9 As shown; among them, Figure 6 The output current is 5A, and the duty cycle is in area 1, case 1; Figure 7 The output current is 14.5A, and the duty cycle is in area 4, case 3; Figure 8 The output current is 19A, and the duty cycle is in area 4, case 2; Fig. 9 The output current is 25A, and the duty cycle is in area 4, case 1.

[0154] Figure 6 In the figure, the turn-on current and turn-off current of the bridge arm where the Q1, Q5, and Q7 switch tubes are located are all zero, realizing ZVS and ZCS soft switching; the turn-on current of the bridge arm where the Q3 switch tube is located is negative, and the turn-off current is minimized by the proposed strategy, realizing ZVS soft switching.

[0155] Figure 7 In the figure, the turn-on current and turn-off current of the bridge arm where the Q1 and Q7 switch tubes are located are both zero, realizing ZVS and ZCS soft switching; the turn-on current of the bridge arm where the Q3 and Q5 switch tubes are located is negative, and the turn-off current is minimized by the proposed strategy, realizing ZVS soft switching.

[0156] Figure 8 In the figure, the turn-on current and turn-off current of the bridge arm where the Q7 switch tube is located are both zero, realizing ZVS and ZCS soft switching; the turn-on current of the bridge arm where the Q1, Q3, and Q5 switch tubes are located is negative, and the turn-off current is minimized by the proposed strategy, realizing ZVS soft switching.

[0157] Fig. 9 In the figure, the turn-on current of the bridge arm where the switch tubes Q1, Q3, Q5, and Q7 are located is negative, and the turn-off current is minimized by the proposed strategy, realizing ZVS soft switching.

[0158] In the above modulation method of the DAB converter based on the optimal off-current, the off-current of all the switch tubes is minimized on the basis of ensuring that the ZVS is achieved, thereby achieving the operation goal of the DAB converter to maintain low switching loss and high transmission efficiency in the full operating range.

Claims

1. A DAB converter modulation method with an optimal off-current, characterized in that: Specifically, to minimize the switching loss of DAB during operation, solutions are sought for Region 1 and Region 4. In Region 1, the phase-shifted duty ratios satisfy the following relationships: -1 < D0 < 0; 0 < D1 < 1; 0 < D2 < 1; D1 < D0 + D2. In Region 4, the phase-shifted duty ratios satisfy the following relationships: 0 < D0 < D1 < 1; 0 < D2 < 1; 1 < D0 + D2. D0, D1, and D2 are phase-shift ratios. When the given current is within different ranges, the TPS duty ratio with the optimal turn-off current is solved using the Karush-Kuhn-Tucker (KKT) conditions, and the optimal phase-shifted duty ratio values can be obtained.

2. The DAB converter modulation method with optimal off-current according to claim 1, characterized in that: The KKT condition method has the following standard form, as shown in Equation (12); Where L is the Lagrangian function; x is the independent variable; h is the equality constraint that needs to be satisfied; g i (i=1,2,…) is the inequality condition that needs to be satisfied.

3. The DAB converter modulation method with optimal off-current according to claim 2, characterized in that: To find the optimal solution of the phase-shift ratio, the KKT conditions in Region 1 are constructed as shown in Equation (13) below; In the formula, λ i , μ is the KKT operator; f(D) is the optimization target, with the minimum off-current as the target; h(D) is the equality constraint; g i(i＝1,2,3…) (D) is the inequality constraint; define the solution D * represents the optimal duty cycle that meets the conditions; L(D,λ i ,μ) is the Lagrangian function in the KKT condition; calculate λ i , μ, the partial derivative of L with respect to D should be equal to 0, h(D * )=0,g ii (D * )≤0,λ i >0,λ i g i (D * )=0; K is the coefficient; D is the phase shift duty cycle matrix; m is the converter voltage gain; n is the high frequency transformer coil turns ratio; V1 is the DC voltage on the primary side of the converter; Solving the above equation yields only one solution, that is, when λ 1,3 ≥0;λ 2,4,5 =0; μ≤0; g 1,3 (D * )=0, the coefficients are shown in the following formula (14); Substitute the constraint condition h(D) into Equation (14), and the optimal solution is obtained as shown in Equation (15); At this time, the given output current value range is as shown in Equation (16); in, is the average current expected to be output by the converter during one switching cycle.

4. The DAB converter modulation method with optimal off-current according to claim 3, characterized in that: Construct the KKT conditions in Working Region 4 as shown in Equation (17); 5. The DAB converter modulation method with optimal off-current according to claim 4, characterized in that: Solving Equation (17) gives four cases where there are solutions: Case 1: When λ i =0; μ≤0; g2(D * )≤0, the coefficients are shown in formula (18); Substitute the constraint condition h(D) into Equation (18), and the optimal solution is obtained as shown in Equation (19); Combining Equation (18) and Equation (19), the given output current value range is obtained as shown in Equation (20); Case 2: When λ 1,3,4 =0; λ2≥0; μ≤0; g2(D*)=0, the coefficients are shown in formula (21); Substitute the constraint condition h(D) into Equation (21), and the optimal solution is obtained as shown in Equation (22); Satisfy the constraints: g1(D * )≤0, that is, m 3 +m 2 When +m-1≥0, combining equation (21) and equation (22), the given output current value range can be obtained as shown in equation (23); Satisfy the constraints: g3(D * )≤0, that is, m 3 +m 2 When +m-1<0, combining equation (21) and equation (22), the given output current value range can be obtained as shown in equation (24); Case 3: When λ 3,4,5… =0;λ 1,2 ≥0; μ≤0; g 1,2 (D * )=0;g3(D * )≤0, the coefficients are shown in formula (25); Substitute the constraint condition h(D) into Equation (25), and the optimal solution is obtained as shown in Equation (26); In m 3 +m 2 Under the premise of +m-1≥0, combining equation (25) and equation (26), the range of the given output current value can be obtained as shown in equation (27); Case 4: When λ 1,4,5… =0;λ 2,3 ≥0; μ≤0; g 2,3 (D * )=0;D * When ∈D4, the coefficients are shown in formula (28); Substitute the constraint condition h(D) into Equation (28), and the optimal solution is obtained as shown in Equation (29); In m 3 +m 2 Under the premise of +m-1<0, combining equation (28) and equation (29), the range of the given output current value can be obtained as shown in equation (30); From all the cases analyzed above, it can be seen that when the given current is within different ranges, the optimal phase-shifted duty ratio values can be obtained after calculation using the KKT conditions.

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