Control method of resonant converter, computer equipment and storage medium

Abstract

The invention discloses a control method of a resonant converter, computer equipment and a storage medium, which are suitable for wide voltage application occasions. An active switch tube on the primary side of the resonant converter adopts inner phase shift control, an active switch tube on the secondary side of the resonant converter adopts outer phase shift control, and the control system comprises two control variables, namely a primary inner phase shift angle theta and a secondary outer phase shift angle phi. An internal phase shift angle theta of a primary side switch tube and an external phase shift angle phi of a secondary side active switch tube are synchronously adjusted through external circuit detection voltage and current signals, so that the gain adjustment capability of the resonant converter is improved, the backflow power of external phase shift (single phase shift and extended phase shift) is reduced, the conduction loss of the converter is reduced, and the conversion efficiency of the converter is improved. The efficiency of the whole system is the highest, and the gain adjustment capability meets the application requirements.

Description

Technical Field

[0001] The present invention relates to the field of power electronics technology, and in particular to a control method of a resonant converter, a computer device and a storage medium. Background Art

[0002] In the field of phase shift control technology for DC-DC converters, many experts and scholars have proposed a variety of phase shift control methods: single-phase-shift (SPS), dual-phase-shift (DPS), extended-phase-shift (EPS) and triple-phase-shift (TPS), which enable the converter to have the advantages of bidirectional energy flow, high power density, easy soft switching and high efficiency, and are widely used in energy storage systems, electric vehicles, data centers, and medical power supplies. At the same time, LLC resonant converter, as a typical isolated soft-switching high-efficiency DC-DC converter that achieves electrical isolation through a transformer, is quite popular in industrial electronic applications. Under any load condition, the primary side switch tube of the resonant converter achieves zero voltage turn-on (ZVS), and the secondary side rectifier diode achieves zero current turn-off (ZCS), which has the advantages of simple structure, magnetic integration, high conversion efficiency, and high power density.

[0003] However, under the control method where the primary full bridge and the secondary full bridge have internal and external phase shift angles, when the input voltage and output voltage do not match, and the switching frequency and internal and external phase shift angles do not match, the converter resonant cavity current stress is large, the secondary reflux power increases, and it is impossible to achieve zero voltage turn-on of all switches, resulting in reduced converter efficiency. At the same time, when the midpoint voltage of the two bridge arms of the primary full bridge is zero, the midpoint voltage of the two bridge arms of the secondary full bridge is also zero, and there is a resonant circulating current on the primary side of the transformer, which increases the circulating current loss of the converter.

[0004] In order to improve the efficiency of the resonant converter and reduce the backflow power and circulating loss, relevant researchers and inventors have proposed different improvement methods. Among them, Huiqing Wen, Lin Zheng and others published the paper "Minimum Backflow Power Control of Bidirectional Isolated DC-DC Converters" online in 2018. This paper is based on the circuit topology of the dual active bridge DAB converter. In order to reduce the backflow power under the external phase shift (EPS) control and improve the converter efficiency, a control strategy aimed at reducing the backflow power is adopted. In terms of control, the active switch tube on the secondary side uses phase shift control to adjust the backflow power, and the phase is shifted relative to the driving pulse of the primary side switch tube. However, due to the characteristics of the circuit structure, the primary side switch tube is turned off at the highest current point, which increases the turn-off loss and the switch tube current stress, making this control suitable for high voltage and low current applications. In addition, Tianhua Zhu, Fang Zhuo and others published "Optimization of Extended Phase-Shift Control for Full-Bridge CLLC Resonant Converter with Improved Light-Load Efficiency" in 2020. The paper proposed an optimized EPS control method for CLLC resonant converters, with the goal of reducing the optimization of the effective value of the resonant current, and adopted a control method of phase shifting the secondary side relative to the primary side. However, the control algorithm is complex and difficult to implement. In addition, Yin Shuai, Deng Yaru, Song Wensheng and others from Southwest Jiaotong University published a resonant converter with an extended phase-shift control working mode in their paper "Global Optimization Method for Extended Phase-Shift Control of Bidirectional Full-Bridge Series Resonant DC / DC Converter". The converter realizes zero voltage turn-on of the primary and secondary side switches, widens the gain adjustment range, and operates at a constant frequency. However, the resonant converter adopts a hybrid phase-shift control method that combines primary side phase shift with secondary side phase shift. When the excitation current cannot be ignored, there is still reflux power on the secondary side. The reflux power depends on the difference between the excitation current and the resonant current. The application of this control is limited due to its efficiency. Summary of the invention

[0005] The purpose of the present invention is to address the deficiencies of the prior art and to provide a control method, computer device and storage medium for a resonant converter. The circulating current loss on the primary side of the transformer and the reflux power on the secondary side under phase-shift control can be reduced, thereby improving the overall efficiency.

[0006] The objective of the present invention is achieved through the following technical solutions:

[0007] In a first aspect, a control method for a resonant converter is provided, wherein the resonant converter comprises a DC input source V in (Ⅰ), a primary inverter switch circuit (Ⅱ), an LLC resonant cavity (Ⅲ), a transformer (Ⅳ), a secondary active bridge rectifier circuit (Ⅴ) and a filter output circuit (Ⅵ); the primary inverter switch circuit (Ⅱ) includes a first switch tube (S1), a second switch tube (S2), a third switch tube (S3) and a fourth switch tube (S4); the secondary active bridge rectifier circuit (V) includes a fifth switch tube (S5) and a sixth switch tube (S6); the driving signals of the first switch tube (S1) and the second switch tube (S2) are complementary to each other; the driving signals of the third switch tube (S3) and the fourth switch tube (S4) are complementary to each other; the driving signals of the fifth switch tube (S5) and the sixth switch tube (S6) are complementary to each other; the duty cycle of the driving signals of all the switch tubes is 50% (ignoring the dead time);

[0008] The control method comprises:

[0009] During steady-state operation, the output voltage Vo and the load current Io of the resonant converter are detected, and the inner phase shift angle θ and the outer phase shift angle θ are calculated according to the output voltage Vo and the load current Io.

[0010] adjusting the drive signals of the first switch tube (S1) and the fourth switch tube (S4) according to the internal phase shift angle θ, so that there is an internal phase shift angle θ between the drive signal of the first switch tube (S1) and the drive signal of the fourth switch tube (S4), and the two drive signals partially overlap;

[0011] According to the external phase angle The driving signals of the first switch tube (S1) and the fifth switch tube (S5) are adjusted so that the driving signals of the first switch tube (S1) and the fifth switch tube (S5) have an external phase shift angle. The two driving signals partially overlap;

[0012] Inner phase shift angle θ and outer phase shift angle The same, and within the inner phase shift angle θ, the resonant current of the primary side of the transformer (Ⅳ) drops to the peak value of the excitation current of the transformer (Ⅳ).

[0013] Specifically, by adjusting the primary side inner phase shift angle θ and the primary side outer phase shift angle Within the inner phase shift angle θ, the primary resonant current of the transformer is in the decreasing stage of the secondary voltage clamping excitation inductance, and there is no primary circulating current loss; moreover, the primary resonant current of the transformer (Ⅳ) decreases to the peak value of the excitation current of the transformer (Ⅳ) within the inner phase shift angle θ. In the positive and negative half switching cycles, the primary side of the transformer (Ⅳ) transmits power to the secondary side, and there is no secondary side reflux power, that is, the current flowing from the output capacitor to the secondary winding through the active switch tube in the reverse direction is zero, thereby reducing the circulating current loss on the primary side of the transformer and the reflux power on the secondary side under phase shift control, improving the overall efficiency, and the gain adjustment capability meets the application requirements.

[0014] Preferably, the inner phase shift angle θ and the normalized voltage gain M of the resonant converter are in a monotonically decreasing relationship.

[0015] Preferably, the input waveform of the LLC resonant cavity (III) is a three-level square wave, wherein the three-level square wave includes a positive level +Vin, a negative level -Vin and a zero level; in the overlapping region of the driving signals of the first switch tube (S1) and the fourth switch tube (S4), the input voltage V AB It is a positive level +Vin. Within the inner phase shift angle θ, the input voltage of the LLC resonant cavity (III) is zero level.

[0016] Preferably, the method further comprises: adjusting the operating frequencies of all switching tubes according to the output voltage Vo and the load current Io.

[0017] Preferably, adjusting the driving signals of the first switch tube (S1) and the fourth switch tube (S4) according to the internal phase shift angle θ so that an internal phase shift angle θ exists between the driving signal of the first switch tube (S1) and the driving signal of the fourth switch tube (S4), specifically includes:

[0018] A phase shift control mode is adopted, and a driving signal of a fourth switch tube (S4) is controlled to perform internal phase shift relative to a driving signal of a first switch tube (S1) according to an internal phase shift angle θ, so that an internal phase shift angle θ exists between the driving signal of the first switch tube (S1) and the driving signal of the fourth switch tube (S4);

[0019] According to the external phase angle The driving signals of the first switch tube (S1) and the fifth switch tube (S5) are adjusted so that the driving signals of the first switch tube (S1) and the fifth switch tube (S5) have an external phase shift angle. Specifically include:

[0020] Adopt phase shift control mode, according to the external phase shift angle The driving signal of the fifth switch tube (S5) is controlled to perform an external phase shift relative to the driving signal of the first switch tube (S1), so that an external phase shift angle exists between the driving signal of the first switch tube (S1) and the driving signal of the fifth switch tube (S5).

[0021] Preferably, when the resonant converter operates under light load, an intermittent pulse mode is adopted for control.

[0022] Preferably, the inner phase shift angle θ and the outer phase shift angle θ are calculated according to the output voltage Vo and the load current Io. Specifically include:

[0023] The resonant converter is mathematically modeled using the segmented analysis method and defined as follows: θ0 = ω r t0,θ1=ω r t1,θ2=ω r t2,θ3=ω r t3,

[0024] The inner phase shift angle θ and the outer phase shift angle Similarly, the mathematical model of the phase shift angle θ is:

[0025]

[0026] 0=sin(θ1-θ2)+sin(θ2)-M·sin(θ2)+ilr0n·cos(θ2)-vcr0n·sin(θ2)+ilr0n

[0027] 0=cos(θ1-θ2)-M-cos(θ2)+M·cos(θ2)+ilr0n·sin(θ2)+vcr0n·cos(θ2)+vcr0n

[0028]

[0029] Internal phase shift angle θ = θ1-θ0

[0030] Where θ0 = ω r t0,θ1=ω r t1,θ2=ω r t2,θ3=ω r ·π is the switching frequency per unit value, ω r is the resonant angular frequency, t0 is the working time 0, t1 is the working time 1, t2 is the working time 2, t3 is the working time 3, θ0 = ω r t0, θ1 = ω r t1, θ2 = ω r t2 and θ3 = ω r·t3 is the per-unit value of the operating frequency of the resonant converter at the corresponding time; fr is the resonant frequency of the resonant cavity, fs is the operating frequency of the resonant converter; n is the transformer ratio, Vo is the output voltage of the resonant converter, Vin is the nominal input voltage of the resonant converter, M is the normalized gain of the resonant converter; RL is the impedance of the load, Io is the load current; Lm is the excitation inductance, Lr is the resonant inductance, λ is the inductance ratio; Zr is the characteristic impedance of the converter, Rac is the equivalent resistance of the transformer secondary side after conversion; Cr is the capacitance of the resonant capacitor; V B is the voltage reference; I B is the current reference; P B is the transmission power reference; iLr is the resonant inductor current value; vCr is the resonant capacitor voltage value; iLrn is the per-unit value of the resonant inductor current; vCrn is the per-unit value of the resonant capacitor voltage; iLr0n is the initial value of the resonant inductor current at t0, and vCr0n is the initial value of the resonant capacitor voltage at t0.

[0031] In a second aspect, a computer device is provided, comprising a memory and a processor, wherein the memory stores a computer program, and the processor implements the steps of the above method when executing the computer program.

[0032] According to a third aspect, a computer-readable storage medium is provided, on which a computer program is stored, and when the computer program is executed by a processor, the steps of the method described above are implemented.

[0033] Compared with the prior art, the invention has the following beneficial effects: the invention adjusts the primary side inner phase shift angle θ and the primary side outer phase shift angle Within the inner phase shift angle θ, the primary resonant current of the transformer is in the decreasing stage of the secondary voltage clamping excitation inductance, and there is no primary circulating current loss; moreover, the primary resonant current of the transformer (IV) decreases to the peak value of the excitation current of the transformer (IV) within the inner phase shift angle θ, and in the positive and negative half switching cycles, the primary side of the transformer (IV) transmits power to the secondary side, and there is no secondary side reflux power, that is, the current flowing from the output capacitor to the secondary winding through the active switch tube in the reverse direction is zero, thereby reducing the circulating current loss on the primary side of the transformer and the reflux power on the secondary side under phase shift control, while reducing the frequency adjustment range of the resonant converter, and improving the overall efficiency and gain adjustment capability; moreover, the parasitic capacitance of the switch tube on the secondary side of the present invention does not participate in the primary side resonance process, so that the gain non-monotonicity phenomenon will not occur. BRIEF DESCRIPTION OF THE DRAWINGS

[0034] Figure 1 It is the topological structure diagram of LLC resonant converter;

[0035] Figure 2 It is a schematic diagram of the external phase shift control method;

[0036] FIG3 is a schematic diagram of an extended phase shift control method;

[0037] Figure 4 It is a schematic diagram of the reflux power optimization control method;

[0038] Figure 5 The switch tube driving waveform and the primary side key working waveform of the control method described in the embodiment;

[0039] Figure 6 The secondary side key working waveform of the control method described in the embodiment;

[0040] Figure 7 is a curve of switching frequency f and voltage gain M of the resonant converter of the embodiment;

[0041] Figure 8 Graph showing the phase shift angle θ and the voltage gain M of the resonant converter of the embodiment. DETAILED DESCRIPTION

[0042] The technical solution of the present invention is described below in conjunction with the accompanying drawings and embodiments so that those skilled in the art can better understand the invention. However, the specific implementation of the technical solution of the present invention is not limited thereto.

[0043] like Figure 1 FIG. 1 is a circuit diagram of a resonant converter according to an embodiment of the present invention. In this embodiment, a control method of a resonant converter is provided, wherein the resonant converter includes a DC input source V in (Ⅰ), primary side inverter switch circuit (Ⅱ), LLC resonant cavity (Ⅲ), transformer (Ⅳ), secondary side active bridge rectifier circuit (Ⅴ) and filter output circuit (Ⅵ);

[0044] The primary inverter switch circuit (II) includes a first switch tube (S1), a second switch tube (S2), a third switch tube (S3), and a fourth switch tube (S4);

[0045] LLC resonant cavity (III) includes resonant capacitor (C r )、Resonant inductor (L r )、Magnetic inductance(L m );

[0046] Transformer (IV) is a high-frequency isolation transformer with a center tap on the secondary side, which has the functions of voltage regulation and electrical isolation;

[0047] The secondary side active bridge rectifier circuit (V) comprises a secondary side fifth active switch tube (S5) and a secondary side sixth active switch tube (S6);

[0048] The secondary side filter output circuit (VI) includes an output filter capacitor (C o ) and load (R o );

[0049] The first switch tube (S1) and the fourth switch tube (S4) share the same driving pulse, and the duty cycle is 50% (ignoring the dead time); the second switch tube (S2) and the third switch tube (S3) share the same driving pulse, and the duty cycle is 50% (ignoring the dead time); the driving pulse of the first switch tube (S1) and the driving pulse of the second switch tube (S2) are complementary to each other, and the driving pulse of the third switch tube (S3) and the driving pulse of the fourth switch tube (S4) are complementary to each other; the driving pulse duty cycle of the fifth switch tube (S5) is 50% (ignoring the dead time), the driving pulse duty cycle of the sixth switch tube (S6) is 50% (ignoring the dead time), and the driving pulse of the fifth switch tube (S5) is complementary to the driving pulse of the sixth switch tube (S6) on the secondary side.

[0050] The control method comprises:

[0051] During steady-state operation, the output voltage Vo and the load current Io of the resonant converter are detected, and the inner phase shift angle θ and the outer phase shift angle θ are calculated according to the output voltage Vo and the load current Io.

[0052] adjusting the drive signals of the first switch tube (S1) and the fourth switch tube (S4) according to the internal phase shift angle θ, so that there is an internal phase shift angle θ between the drive signal of the first switch tube (S1) and the drive signal of the fourth switch tube (S4), and the two drive signals partially overlap;

[0053] According to the external phase angle The driving signals of the first switch tube (S1) and the fifth switch tube (S5) are adjusted so that the driving signals of the first switch tube (S1) and the fifth switch tube (S5) have an external phase shift angle. The two driving signals partially overlap;

[0054] Inner phase shift angle θ and outer phase shift angle The same, and within the inner phase shift angle θ, the resonant current of the primary side of the transformer (Ⅳ) drops to the peak value of the excitation current of the transformer (Ⅳ).

[0055] Specifically, the inner phase shift angle θ refers to the phase shift control of the fourth switch tube (S4) relative to the first switch tube (S1) in the primary full bridge of the transformer, and the phase difference between the first switch tube (S1) and the fourth switch tube (S4) is θ; the outer phase shift angle It means that the fifth switch tube (S5) in the secondary side circuit of the transformer is phase-shifted relative to the first switch tube (S1), and the phase difference of the fifth switch tube (S5) relative to the first switch tube (S1) is

[0056] In this embodiment, it also includes: adjusting the operating frequency of all switch tubes according to the output voltage Vo and the load current Io.

[0057] In the specific implementation process, during the control process, the switching frequencies f of the first switch tube (S1), the second switch tube (S2), the third switch tube (S3), the fourth switch tube (S4), the fifth switch tube (S5), and the sixth switch tube (S6) are synchronously adjusted, and the switching frequencies f are changed within a frequency range higher than the series resonant frequency, and the switching frequencies f of the first switch tube (S1), the second switch tube (S2), the third switch tube (S3), the fourth switch tube (S4), the fifth switch tube (S5), and the sixth switch tube (S6) are kept consistent; and the fourth switch tube (S4) and the third switch tube (S3) are respectively subjected to internal phase shift control relative to the first switch tube (S1) and the second switch tube (S2), with an internal phase shift angle of θ. At the same time, the fifth switch tube (S5) and the sixth switch tube (S6) are respectively subjected to external phase shift control relative to the driving pulses of the first switch tube (S1) and the second switch tube (S2), with an external phase shift angle of Inner phase shift θ and outer phase shift angle In the control process, the switching frequency f and the internal (external) phase shift angle θ are changed to keep the switching frequency f and the internal (external) phase shift angle θ equal to each other, and the reflux power is optimized during the operation of the converter, thereby improving the efficiency of the converter. To realize the control of the converter transmission power, adjust the control variables inner phase shift angle θ and outer phase shift angle and switching frequency f, optimize the resonant converter reflux power, so that the efficiency of the whole system is the highest and the gain adjustment capability meets the application requirements. Compared with the traditional primary side internal phase shift control method, the method described in this embodiment significantly reduces the primary side circulation loss; compared with the single phase shift control method, the control method described in this embodiment effectively reduces the reflux power of the secondary side active switch tube and improves the transformer efficiency; at the same time, by controlling the primary side switch tube through phase shift, the frequency adjustment range is reduced, the transformer design is optimized, and the conversion efficiency and power density of the converter are improved in wide voltage applications.

[0058] Specifically, the inner phase shift angle θ and the normalized voltage gain M of the resonant converter are in a monotonically decreasing relationship.

[0059] In this embodiment, the input waveform of the LLC resonant cavity (III) is a three-level square wave, and the three-level square wave includes a positive level +Vin, a negative level -Vin and a zero level; in the overlapping area of ​​the driving signals of the first switch tube (S1) and the fourth switch tube (S4), the input voltage V AB It is a positive level +Vin. Within the inner phase shift angle θ, the input voltage of the LLC resonant cavity (III) is zero level.

[0060] Specifically, the overlapping part of the driving pulse of the first switch tube (S1) and the driving pulse of the fourth switch tube (S4) is a positive level +Vin, the overlapping part of the driving pulse of the second switch tube (S2) and the driving pulse of the fourth switch tube (S4) is a negative level -Vin, and the overlapping part of the driving pulse of the second switch tube (S2) and the driving pulse of the fourth switch tube (S4) is a zero level.

[0061] It should be noted that the control method is applied to an LLC resonant converter with a secondary active switching rectifier circuit and a center-tapped transformer, and the external phase angle The driving pulse (also called driving signal) of the secondary side active switch tube lags behind the driving pulse of the primary side active switch tube. In addition, the secondary side rectifier circuit can also be an active full-bridge rectifier circuit, and its circuit structure is the same as the primary side active full-bridge circuit, wherein the upper and lower active switch tube driving pulses of the rectifier circuit bridge arm are complementary and have a duty cycle of 50%, the diagonal switch tube driving pulses are the same, and the driving pulse of the secondary side active switch tube lags behind the driving pulse of the primary side active switch tube. The control method described in this embodiment is applicable to both of the above two circuits. This embodiment is applied in Figure 1 The topology shown, Figure 1 Middle: V in is the input voltage; S1-S4 are the primary switch tubes; C1-C4 are the parasitic capacitance of the primary switch tubes; D1-D4 are the primary switch tube body diodes; Lr is the auxiliary inductance and leakage inductance equivalent inductance; Cr is the resonant capacitor; fr and fs are the resonant frequency and switching frequency respectively; n is the transformer ratio; VAB and VCD are the primary H midpoint voltage and the transformer secondary output voltage respectively; i Lr Resonant cavity current; I o is the output load, and M is defined as the voltage gain, M = n*V o / V in .

[0062] During the power transmission process, due to the phase shift between VAB and VCD, the resonant current has a time period with a phase opposite to that of VAB and VCD, such as Figure 2 In [ω s *t0,ω s *t1] and [π,ω s *t3]. During this period, the transmission power flows in the reverse direction. s * During the time period of [α,ω] and [π,π+α], the primary transmission power flows in the reverse direction, which is defined as the primary return power. s *t1] and [π+α,ω s*t3] During the time period, the secondary transmission power flows in the reverse direction, which is defined as the secondary return power. When the transmission power is constant, the existence of the return power will increase the forward transmission power, increase the effective value of the resonant cavity current, increase the path loss of the system, and reduce the efficiency. Both the primary return power and the secondary return power will reduce the efficiency.

[0063] The schematic diagram of the extended phase shift (EPS) control method is shown in FIG3 , where θ is the internal phase shift angle between the driving pulses of the first switch tube S1 and the fourth switch tube S4 , is the outward shift phase angle of the driving signal of the fifth switch tube S5 and the first switch tube S1, and α is the phase angle of the resonant cavity current at the zero point. Satisfies 0<θ<π, The extended phase shift is achieved by controlling the inner phase shift θ and the outer phase shift The size of makes the H-bridge output voltage and the secondary output voltage zero when the power is refluxed, thereby reducing the reflux power and improving efficiency. The two working modes of EPS control are as follows Figure 3(a) and 3(b) As shown, corresponding to modes A and B, mode A: S5 control signal lags S1, φ<θ; mode B: S5 control signal lags S1, φ>θ. Under EPS control, by properly controlling the inner phase shift angle θ and the outer phase shift angle Can effectively reduce reflux power and switching loss.

[0064] In the control method proposed in this embodiment, the internal phase shift θ between the primary active switch tube driving pulses and the external phase shift angle between the primary and secondary switch tube driving pulses are are equal and are a specific phase shift angle calculated, which is related to the input voltage, output voltage, transformation ratio and resonant cavity parameters.

[0065] The calculation and proof process of the control method is as follows:

[0066] To simplify the analysis, the following assumptions are made:

[0067] 1. All switches are ideal switches, the switching process can be ignored, and there is no switching loss.

[0068] 2. The dead time of the switch can be ignored.

[0069] 3. The transformer is an ideal transformer.

[0070] 4. Ignore the stray resistance of the converter.

[0071] The FBSR converter (full-bridge-series-resonant converter) is mathematically modeled using the segmented analysis method and defined as follows: Let: θ0 = ωr t0,θ1=ω r t1,θ2=ω r t2,θ3=ω r t3, Voltage reference V B , current reference I B , transmission power reference P B , respectively V B =V in , Where θ0 = ω r t0,θ1=ω r t1,θ2=ω r t2,θ3=ω r ·π is the switching frequency per unit value.

[0072] Among them, ω r is the resonant angular frequency, t0 is the working time 0, t1 is the working time 1, t2 is the working time 2, t3 is the working time 3, θ0 = ω r t0, θ1 = ω r t1, θ2 = ω r t2 and θ3 = ω r ·t3 is the per-unit value of the operating frequency of the resonant converter at the corresponding time; fr is the resonant frequency of the resonant cavity, fs is the operating frequency of the resonant converter; n is the transformer ratio, Vo is the nominal output voltage of the resonant converter, Vin is the nominal input voltage of the resonant converter, M is the normalized gain of the resonant converter; RL is the impedance of the load, Io is the output current; Lm is the excitation inductance, Lr is the resonant inductance, λ is the inductance ratio; Zr is the characteristic impedance of the converter, Rac is the equivalent resistance of the transformer secondary side after conversion; Cr is the capacitance of the resonant capacitor; V B is the voltage reference; I B is the current reference; P B is the transmission power reference; iLr is the resonant inductor current value; vCr is the resonant capacitor voltage value; iLrn is the per-unit value of the resonant inductor current; vCrn is the per-unit value of the resonant capacitor voltage; iLr0n is the initial value of the resonant inductor current at t0, and vCr0n is the initial value of the resonant capacitor voltage at t0.

[0073] According to the basic assumptions, the FBSR converter works at the critical operating point between mode A and mode B, and the phase shift angle is a specific phase shift angle θ. The schematic diagram of the reflux power optimization control method is shown in Figure 4 shown.

[0074] The LLC external phase shift power optimization control mathematical model is:

[0075]

[0076] 0=sin(θ1-θ2)+sin(θ2)-M·sin(θ2)+ilr0n·cos(θ2)-vcr0n·sin(θ2)+ilr0n

[0077] 0=cos(θ1-θ2)-M-cos(θ2)+M·cos(θ2)+ilr0n·sin(θ2)+vcr0n·cos(θ2)+vcr0n

[0078]

[0079] The common parameters in the mathematical model are θ1, θ2, iLr0n, Vcr0n, M, Q, and λ. Among them, the inductance ratio λ is related to the parameters of the converter and is a known quantity, and Q can be set according to the load conditions. The mathematical model is a set of equations of dependent variables θ1, θ2, iLr0n, and Vcr0n with respect to the independent variable M. With the help of mathematical analysis tools, the optimal phase shift angle can be solved.

[0080] Figure 5 : The switch tube driving waveform and key working waveform of this embodiment; Figure 6 Medium V gs1 、V gs2 、V gs3 、V gs4 Respectively represent the driving pulses of the first switch tube (S1), the second switch tube (S2), the third switch tube (S3), and the fourth switch tube (S4); V gs5 、V gs6 Respectively represent the driving pulses of the fifth switch tube (S5) and the sixth switch tube (S6); V AB It represents the midpoint voltage of the two bridge arms of the primary side and is a three-level square wave, V CD is the transformer primary voltage, V TP Represents the transformer primary voltage and is a two-level square wave, i Lr Represents the current flowing through the resonant inductor (L r ) of the resonant current, i Lm Indicates the current flowing through the magnetizing inductance (L m ) of the excitation current.

[0081] Depend on Figure 5It can be seen that the driving pulse duty ratios of the first switch tube (S1) and the fourth switch tube (S4) of the primary inverter switch circuit (II) are both 50% (ignoring the dead time); the driving pulse duty ratios of the second switch tube (S2) and the third switch tube (S3) are both 50% (ignoring the dead time); the driving pulse of the first switch tube (S1) is complementary to the driving pulse of the second switch tube (S2), and the driving pulse of the third switch tube (S3) is complementary to the driving pulse of the fourth switch tube (S4). The driving pulse of the fourth switch tube (S4) is controlled to be internally phase-shifted relative to the driving pulse of the first switch tube (S1), and the internal phase-shift angle is θ, and the driving pulse of the third switch tube (S3) is controlled to be internally phase-shifted relative to the driving pulse of the second switch tube (S2), and the internal phase-shift angle is θ; the driving pulse duty ratio of the fifth switch tube (S5) of the secondary side active bridge rectifier circuit (V) is 50% (ignoring the dead time), and the driving pulse of the fifth switch tube (S5) is controlled to be externally phase-shifted relative to the first switch tube (S1), and the external phase-shift angle is The driving pulse duty cycle of the sixth switch tube (S6) of the secondary side active bridge rectifier circuit (V) is 50% (ignoring the dead time), and an external phase shift control is performed relative to the first switch tube (S2), and the phase shift angle is Under the condition that the converter load is 125A and the switching frequency is 120KHz (resonant frequency is 100KHz), after calculation by mathematical model, the phase shift angle of the embodiment is 21.039 degrees.

[0082] Figure 6 It is the key working waveform of the secondary side of the control method of the present invention, Figure 6 Here, IS1 represents the current flowing through the fifth switch tube (S5), IS2 represents the current flowing through the sixth switch tube (S6), and ICC represents the output current of the secondary side rectifier circuit.

[0083] Resonant current i Lr In V AB The non-zero level phase is in the rising phase and V AB The zero level stage (within the phase shift angle) is in the falling stage. When the fourth switch tube (S4) or the third switch tube (S3) is turned off, the resonant current i Lr Just drop to the excitation current i Lm Maximum value, in the positive and negative half cycles, the resonant current is always greater than the excitation current i Lm , the secondary circuit current does not flow to the primary side, that is, the primary side return power is suppressed, which significantly improves the conversion efficiency of the resonant converter.

[0084] Figure 6It can be seen that the current IS1 flowing through the fifth switch tube (S5) just drops to zero and the fifth switch tube (S5) is turned off, and the current IS2 flowing through the sixth switch tube (S6) just drops to zero and the sixth switch tube (S6) is turned off. It can be seen from the output current ICC that the secondary circuit operates in the current critical mode. At this time, the secondary active switch tube has no additional return power loss.

[0085] In the phase shift reflux power optimization control technology of this embodiment, the control variables are multiple variables: inner phase shift angle θ, outer phase shift angle And the switching frequency f. Among them, the inner phase angle θ and the outer phase angle are derived theoretically. Equality is conducive to simplifying control and facilitating application implementation.

[0086] Figure 7 The gain curve of the resonant converter in steady-state working mode under full load conditions in the embodiment of the present invention is shown in Figure 1. The horizontal axis is the converter switching frequency f, and the vertical axis is the normalized voltage gain M. The relationship curve between the normalized voltage gain M and the switching frequency f can be obtained using mathematical analysis tools. Figure 7 As shown in the figure, it can be seen that the voltage gain M and the switching frequency f present a monotonically decreasing relationship, that is, the voltage gain M decreases as the switching frequency f increases, which is beneficial to controlling the output of the converter by the switching frequency f.

[0087] Figure 8 : is the gain curve of the resonant converter in steady-state working mode under full load conditions in the embodiment of the present invention. The horizontal axis is the internal phase shift angle θ of the converter, and the vertical axis is the normalized voltage gain M. According to the mathematical analysis tool, the relationship curve between the normalized voltage gain M and the phase shift angle θ can be obtained as shown in Figure 8 As shown in the figure, it can be seen that the voltage gain M and the internal phase shift angle θ present a monotonically decreasing relationship, that is, the voltage gain M decreases as the internal phase shift angle θ increases, which is beneficial to controlling the output of the converter through the internal phase shift angle θ.

[0088] In this embodiment, when the resonant converter is operating under light load, an intermittent pulse mode is used for control. Indirect pulse mode means that the system periodically shuts down the drive pulse to reduce the drive loss of the converter and reduce the average operating frequency. In the intermittent pulse mode, the controller sets an operating frequency threshold. When the actual operating frequency attempts to exceed the threshold frequency, the converter enters the intermittent pulse mode. In the intermittent pulse mode, when the error value between the feedback voltage and the expected voltage is greater than a certain set threshold, the controller shuts down the drive signal, and when the output voltage is lower than another threshold, the drive signal is started again.

[0089] In the description of the above embodiments, due to the use of the phase-shifted reflux power optimization method described in this application, the resonant converter reduces the reflux power, significantly reduces the loss of the reflux power of the traditional external phase-shift control, and is beneficial to improving the efficiency of the converter. In addition, the converter operates in a frequency range higher than the series resonant frequency, which reduces the frequency adjustment range of the traditional frequency conversion control, is beneficial to the optimization design of the magnetic components of the resonant converter, reduces the volume of the transformer, and improves the power density. Compared with the disclosed external phase-shift (SPS, EPS) control technology, the control method described in this application significantly reduces the reflux power and reduces the circulation loss on the primary side, which is beneficial to improving the conversion efficiency. It can be seen from the phase-shifted normalized gain curve that the control method described in this application enables the resonant converter to have a wider gain adjustment capability, so that the resonant converter has more application advantages in wide voltage applications, and improves the efficiency and power density of the resonant converter.

[0090] In another embodiment, a computer device is provided, including a memory and a processor, wherein the memory stores a computer program, and the processor implements the steps of the above method when executing the computer program.

[0091] In another embodiment, a computer-readable storage medium is provided, on which a computer program is stored. When the computer program is executed by a processor, the steps of the method described above are implemented.

[0092] The embodiments described above are merely examples of the technical solutions and contents of the present invention. It should be noted that the above embodiments should not be regarded as limiting the present invention. For those skilled in the art, several improvements and modifications can be made without departing from the spirit and scope of the present invention, but they will not deviate from the spirit of the present invention or exceed the scope defined by the attached claims, and should also be regarded as the protection scope of the present invention.

Claims

1. A control method for a resonant converter, the resonant converter comprising a DC input source V in (Ⅰ), a primary inverter switch circuit (Ⅱ), an LLC resonant cavity (Ⅲ), a transformer (Ⅳ), a secondary active bridge rectifier circuit (Ⅴ) and a filter output circuit (Ⅵ); the primary inverter switch circuit (Ⅱ) includes a first switch tube (S1), a second switch tube (S2), a third switch tube (S3) and a fourth switch tube (S4); the secondary active bridge rectifier circuit (V) includes a fifth switch tube (S5) and a sixth switch tube (S6); the driving signals of the first switch tube (S1) and the second switch tube (S2) are complementary to each other; the driving signals of the third switch tube (S3) and the fourth switch tube (S4) are complementary to each other; the driving signals of the fifth switch tube (S5) and the sixth switch tube (S6) are complementary to each other; The duty cycle of the drive signal of all switches is 50% (ignoring the dead time); It is characterized in that The control method comprises: During steady-state operation, the output voltage Vo and the load current Io of the resonant converter are detected, and the inner phase shift angle θ and the outer phase shift angle φ are calculated according to the output voltage Vo and the load current Io; adjusting the drive signals of the first switch tube (S1) and the fourth switch tube (S4) according to the internal phase shift angle θ, so that there is an internal phase shift angle θ between the drive signal of the first switch tube (S1) and the drive signal of the fourth switch tube (S4), and the two drive signals partially overlap; adjusting the driving signals of the first switch tube (S1) and the fifth switch tube (S5) according to the external phase shift angle φ, so that the driving signal of the first switch tube (S1) and the driving signal of the fifth switch tube (S5) have an external phase shift angle φ, and the two driving signals partially overlap; The inner phase shift angle θ is the same as the outer phase shift angle φ, and within the inner phase shift angle θ, the resonant current of the primary side of the transformer (Ⅳ) drops to the peak value of the excitation current of the transformer (Ⅳ).

2. The control method of the resonant converter according to claim 1, characterized in that: The inner phase shift angle θ is in a monotonically decreasing relationship with the normalized voltage gain M of the resonant converter.

3. The control method of the resonant converter according to claim 2, characterized in that: The input waveform of the LLC resonant cavity (III) is a three-level square wave, and the three-level square wave includes a positive level +Vin, a negative level -Vin and a zero level; in the overlapping area of ​​the driving signals of the first switch tube (S1) and the fourth switch tube (S4), the input voltage V AB It is a positive level +Vin. Within the inner phase shift angle θ, the input voltage of the LLC resonant cavity (III) is zero level.

4. The control method of the resonant converter according to any one of claims 1 to 3, characterized in that: Also includes: The operating frequencies of all switching tubes are adjusted according to the output voltage Vo and the load current Io.

5. The control method of the resonant converter according to claim 1, characterized in that: The drive signals of the first switch tube (S1) and the fourth switch tube (S4) are adjusted according to the internal phase shift angle θ so that an internal phase shift angle θ exists between the drive signal of the first switch tube (S1) and the drive signal of the fourth switch tube (S4), specifically comprising: A phase shift control mode is adopted, and a driving signal of a fourth switch tube (S4) is controlled to perform internal phase shift relative to a driving signal of a first switch tube (S1) according to an internal phase shift angle θ, so that an internal phase shift angle θ exists between the driving signal of the first switch tube (S1) and the driving signal of the fourth switch tube (S4); The driving signals of the first switch tube (S1) and the fifth switch tube (S5) are adjusted according to the external phase shift angle φ, so that the driving signals of the first switch tube (S1) and the fifth switch tube (S5) have the external phase shift angle φ, specifically comprising: A phase shift control mode is adopted, and a driving signal of a fifth switch tube (S5) is controlled to perform an external phase shift relative to a driving signal of a first switch tube (S1) according to an external phase shift angle φ, so that an external phase shift angle φ exists between the driving signal of the first switch tube (S1) and the driving signal of the fifth switch tube (S5).

6. The control method of the resonant converter according to claim 1, characterized in that: When the resonant converter operates under light load, an intermittent pulse mode is adopted for control.

7. A computer device comprising a memory and a processor, wherein the memory stores a computer program, characterized in that: When the processor executes the computer program, the steps of the method according to any one of claims 1 to 7 are implemented.

8. A computer-readable storage medium having a computer program stored thereon, characterized in that: When the computer program is executed by a processor, the steps of the method according to any one of claims 1 to 6 are implemented.

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