Motor, motor control method and control system thereof

Abstract

The present application provides a motor control method, which includes obtaining a first electrical signal from a motor stator; obtaining a stator back electromotive force based on the first electrical signal; performing amplitude and phase compensation on the stator back electromotive force to obtain a compensated back electromotive force; processing the compensated back electromotive force using a low-pass filter to obtain a filtered signal; outputting motor rotor position information based on the filtered signal, and implementing motor control based on the motor rotor position information; wherein the motor rotor position information includes motor angular velocity and rotor position. The motor control method provided in the embodiment of the present application can greatly reduce the impact of sudden changes in the input signal on the accuracy of rotor position observation by preferentially compensating the stator back electromotive force and then performing low-pass filtering to obtain a filtered signal, thereby improving the system's ability to resist input signal disturbances.

Description

[0001] This application claims priority to Chinese patent application No. 202011022168.4, the contents of which are incorporated by reference into this application. Technical Field

[0002] The present application relates to the technical field of motor control, and in particular to a motor, a motor control method and a control system thereof. Background Art

[0003] With the development of power electronics technology and microprocessor technology, permanent magnet synchronous motors and their vector control have been widely used in the industrial field, such as artillery servo control, CNC machine tool spindle and feed control, etc., and have replaced DC motors and their servo systems.

[0004] Usually, high-performance control of permanent magnet synchronous motors requires feedback of rotor position or speed information. Generally, rotor position or speed sensors are used to obtain rotor position or speed information, which greatly increases hardware costs and is not conducive to the development of miniaturization. Summary of the invention

[0005] On the one hand, an embodiment of the present application provides a motor control method, which includes acquiring a first electrical signal from a motor stator; obtaining a stator back electromotive force based on the first electrical signal; performing amplitude and phase compensation on the stator back electromotive force to obtain a compensated back electromotive force; processing the compensated back electromotive force using a low-pass filter to obtain a filtered signal; outputting motor rotor position information based on the filtered signal, and realizing motor control based on the motor rotor position information; wherein the motor rotor position information includes motor angular velocity and rotor position.

[0006] On the other hand, an embodiment of the present application further provides a motor, including a stator, a rotor and a controller, wherein the stator and the rotor are respectively used to be electrically connected to the controller so that the controller can execute the motor control method described in the above embodiment.

[0007] On the other hand, an embodiment of the present application further provides a motor control system, including an acquisition unit, an operation unit, a compensation unit and a processing unit; the acquisition unit is used to acquire a first electrical signal of a motor stator; the operation unit is used to obtain a stator back electromotive force based on the first electrical signal; the compensation unit is used to compensate the stator back electromotive force to obtain a compensated back electromotive force; the processing unit is used to process the compensated back electromotive force to obtain a filtered signal, and output motor rotor position information based on the filtered signal; the control unit is used to control the motor based on the motor rotor position information.

[0008] The motor, motor control method, system and readable storage medium provided in the embodiments of the present application can greatly reduce the impact of sudden changes in the input signal on the accuracy of rotor position observation by preferentially compensating the stator back electromotive force and then performing low-pass filtering to obtain a filtered signal, thereby improving the system's ability to resist input signal disturbances. BRIEF DESCRIPTION OF THE DRAWINGS

[0009] In order to more clearly illustrate the technical solutions in the embodiments of the present application, the drawings required for use in the description of the embodiments will be briefly introduced below. Obviously, the drawings described below are only some embodiments of the present application. For ordinary technicians in this field, other drawings can be obtained based on these drawings without paying any creative work.

[0010] Figure 1 is a block diagram of a motor control system in some embodiments of the present application;

[0011] Figure 2 is a flow chart of a motor control method in some embodiments of the present application;

[0012] Figure 3 is a principle structural block diagram of a flux observer in some embodiments of the present application;

[0013] Figure 4 is a flowchart of a motor control method in some other embodiments of the present application;

[0014] Figure 5 is a principle structure block diagram of a flux observer in other embodiments of the present application;

[0015] Figure 6 It is a schematic diagram of a process for calibrating the quadrature-axis inductance of a motor based on a towing experimental platform in some embodiments of the present application;

[0016] Figure 7 is a graph showing the change of quadrature-axis inductance with quadrature-axis current in some embodiments of the present application;

[0017] Figure 8 is a principle block diagram of a flux normalization phase-locked loop in some embodiments of the present application;

[0018] Fig. 9 is a simplified principle block diagram of a normalized phase-locked loop in some embodiments of the present application;

[0019] Fig.10 is a flowchart of a motor control method in some other embodiments of the present application;

[0020] Fig.11 is a principle structural block diagram of a sliding mode observer in some other embodiments of the present application;

[0021] Fig.12 is a schematic diagram of a saturation function sat() in some other embodiments of the present application;

[0022] Fig.13 is a principle block diagram of a back electromotive force normalization phase-locked loop in other embodiments of the present application;

[0023] Fig.14 is a schematic block diagram of the structure of a motor in some embodiments of the present application;

[0024] Fig.15 is a schematic block diagram of the structure of a motor control system in some embodiments of the present application;

[0025] Fig.16 It is a schematic diagram of the structure of a computer-readable storage medium in some embodiments of the present application. DETAILED DESCRIPTION

[0026] The present application is further described in detail below in conjunction with the accompanying drawings and examples. It is particularly noted that the following examples are only used to illustrate the present application, but are not intended to limit the scope of the present application. Similarly, the following examples are only some embodiments of the present application rather than all embodiments, and all other embodiments obtained by ordinary technicians in the field without making creative work are within the scope of protection of the present application.

[0027] Reference to "embodiments" herein means that a particular feature, structure, or characteristic described in conjunction with the embodiments may be included in at least one embodiment of the present application. The appearance of the phrase in various locations in the specification does not necessarily refer to the same embodiment, nor is it an independent or alternative embodiment that is mutually exclusive with other embodiments. It is explicitly and implicitly understood by those skilled in the art that the embodiments described herein may be combined with other embodiments.

[0028] The term "motor" as used herein includes, but is not limited to, an electromagnetic device configured to achieve electrical energy conversion or transmission according to the law of electromagnetic induction, and its main function is to generate a driving torque as a power source for electrical appliances or various machines. Generally, the basic structure of a motor includes a stator and a rotor, and the direction in which the current-carrying conductor moves in a magnetic field is related to the direction of the current and the direction of the magnetic field. The basic working principle of a motor is to use an energized coil (i.e., a stator winding) to generate a rotating magnetic field and act on the rotor to form a magneto-electrodynamic rotational torque, which causes the motor to rotate.

[0029] Among them, the permanent magnet synchronous motor (PMSM) has significant advantages such as high power density, good speed regulation performance, and flexible and diverse structure. Therefore, it has been widely used in drive traction, position servo, efficient power output and other fields.

[0030] Specifically, the permanent magnet synchronous motor is mainly composed of components such as the stator, rotor and end cover. The stator is made of laminated sheets to reduce the iron loss generated when the motor is running, and is equipped with a three-phase AC winding, called an armature. The rotor can be made into a solid form or pressed from laminated sheets, on which permanent magnet materials are installed. Depending on the position of the permanent magnet material on the motor rotor, the permanent magnet synchronous motor can be divided into two structural forms: surface-mounted and built-in. The surface-mounted rotor has a simple magnetic circuit structure and low manufacturing cost.

[0031] The magnetic circuit structure of the built-in rotor mainly includes radial, tangential and hybrid types. The difference between them lies in the different relationship between the magnetization direction of the permanent magnet and the rotation direction of the rotor. Since the permanent magnet is placed inside the rotor, the rotor surface can be made into pole shoes. Copper bars or cast aluminum are placed inside the pole shoes to play a starting and damping role, and the steady-state and dynamic performance are good. Since the magnetic circuit of the built-in rotor is asymmetric, it will generate reluctance torque during operation, which helps to improve the power density and overload capacity of the motor itself, and such a structure is easier to achieve weak magnetic speed expansion.

[0032] It is understandable that high-performance control of permanent magnet synchronous motors requires feedback of rotor position or speed information. The current key research issue at home and abroad is how to achieve high-performance control of permanent magnet motors without using rotor position or speed sensors, thereby reducing hardware costs, reducing system size, and improving reliability.

[0033] The applicant found in the research that sensorless control of permanent magnet synchronous motor can be achieved by introducing motor position observation technology, that is, sensorless algorithms such as flux observer and sliding mode observer can be introduced. The flux observer has the advantages of simple structure, low implementation difficulty and fast calculation speed. Therefore, the introduction of flux observer can achieve sensorless control of motor and improve observation performance. Sliding mode observer is widely used in medium and high speed occasions of motors due to its insensitivity to parameter changes and external interference, simple algorithm and low engineering implementation difficulty.

[0034] Based on the above invention ideas, the embodiment of the present application provides a motor control system, which obtains the motor angular velocity and rotor position through a flux observer or a sliding mode observer to realize motor control operation. Figure 1 , Figure 1 is a block diagram of the motor control system in some embodiments of the present application. The control system adopts double closed-loop vector control, the outer loop is the speed loop, and the inner loop is the current loop. In the embodiments of the present application, the motor angular velocity w is obtained by a flux observer or a sliding mode observer. e and the motor rotor position θ e , the three-phase current i of the motor a 、i b 、ic The quadrature axis current i is obtained by Clark and Park transformation. q and the direct axis current i d . The angular velocity is given by w ref With angular velocity w e The difference is output to the speed loop proportional-integral (PI) regulator, and the speed loop PI regulator outputs the quadrature axis current i qref .

[0035] Among them, the output quadrature axis current i of the speed loop is qref With the quadrature axis current i q The difference is output to the current loop PI regulator, and the current loop PI regulator outputs the quadrature axis voltage u q ; Given direct axis current i dref With the direct axis current i d The difference is output to the current loop PI regulator, and the current loop PI regulator outputs the direct axis voltage u d .

[0036] Furthermore, the quadrature axis voltage u output by the current loop is q , direct axis voltage u d The α-β axis voltage u is obtained by inverse Park transformation α 、u β , and then input into the space vector pulse width modulation (SVPWM) to control the action of the three-phase inverter bridge switching devices, thereby controlling the operation of the PMSM, that is, realizing PMSM sensorless control.

[0037] The applicant further discovered that the motor angular velocity w can be obtained by using the flux observer e and the motor rotor position θ e The stator flux of the motor needs to be introduced, and the stator flux calculation needs to be implemented using an integrator, which will cause integral drift problems and changes in the saturation degree of the motor's cross-axis magnetic circuit, thereby affecting the accuracy of the motor rotor position flux observation.

[0038] To solve the above technical problems, the applicant discovered through experiments that introducing a low-pass filter instead of an integrator in the flux observer can solve the integral drift problem and compensate for the amplitude and phase errors that may be generated by the low-pass filter to ensure the accuracy of the motor rotor position flux observation.

[0039] Based on the technical ideas for solving the above technical problems, the present application further provides a motor control method, wherein the motor control method is implemented based on a flux observer. Figure 2 and Figure 3 , Figure 2 is a flow chart of a motor control method in some embodiments of the present application, Figure 3 : is a principle structural block diagram of a flux observer in some embodiments of the present application. The motor control method may generally include the following steps:

[0040] S201, obtaining a first electrical signal from a motor stator;

[0041] The first electrical signal may include electrical signals such as stator voltage and stator current. In the embodiment of the present application, the first electrical signal may be the stator voltage u in the stationary αβ coordinate system. α 、u β and / or stator current i α 、i β .

[0042] Furthermore, the first electrical signal is input to the flux observer so that the flux observer can output the motor rotor position information, and then the motor control can be realized according to the motor electronic rotor position information. In other words, the first electrical signal can be obtained by the flux observer, and then the motor rotor position information can be output after being processed by the flux observer.

[0043] S202, obtaining a stator back electromotive force according to the first electrical signal;

[0044] Specifically, the stator back electromotive force can be obtained by obtaining the stator voltage and / or stator current of the motor. Of course, in other embodiments, the stator back electromotive force can also be obtained by obtaining other electrical signals of the motor, which are not listed one by one in the embodiments of the present application.

[0045] In some embodiments of the present application, the stator voltage u in the stationary αβ coordinate system can be obtained by using the flux observer α 、u β and the stator current i α 、i β , and the stator back electromotive force e of the motor is calculated by the formula sα 、e sβ , and its calculation formula is as follows:

[0046] e sα =u α -i α R s ; (1)

[0047] e sβ =u β -i β R s ; (2)

[0048] In formula (1) and (2), R s is the stator resistance.

[0049] Among them, the stator voltage uα 、u β The AC and DC voltages output by the above current loop can be obtained by inverse Park transformation, and the stator current i α 、i β The three-phase current i of the above motor can be used a 、i b 、i c Of course, in other embodiments, the stator voltage u can also be obtained by directly obtaining α 、u β and the stator current i α 、i β .

[0050] S203, performing amplitude and phase compensation on the stator back electromotive force to obtain compensated back electromotive force;

[0051] Specifically, the stator back electromotive force e can be measured by using the flux observer. sα 、e sβ Compensation is performed and the compensated back electromotive force e' is obtained sα 、e' sβ , the compensation formula is as follows:

[0052] e' sα =e sα +e sαcomp ; (3)

[0053] e' sβ =e sβ +e sβcomp ; (4)

[0054] In formula (3) and (4), e sαcomp 、e sβcomp are the amplitude and phase compensation generated by the low-pass filter in the α and β coordinate systems, respectively, and can be obtained according to the amplitude and phase compensation functions.

[0055] Furthermore, according to the transfer function of the low-pass filter and the transfer function of the integrator, it can be concluded that when a low-pass filter is used instead of an integrator, the amplitude of the stator flux is equal to the amplitude of the integrator. times, the phase of the stator flux leads the phase of the integrator by π / 2-arctan(|w e | / w c ), where w e is the motor angular velocity, w cis the cut-off frequency of the low-pass filter. Therefore, the amplitude attenuation and the leading phase are correspondingly compensated for the amplitude and phase of the back electromotive force in the α and β coordinate systems to eliminate the amplitude and phase errors that will be generated by the introduction of the low-pass filter. It can be understood that the amplitude and phase compensation of the stator back electromotive force includes amplitude attenuation compensation and phase leading compensation.

[0056] Among them, the transfer function of the low-pass filter introduced in the flux observer is: In the formula, w c is the cutoff frequency of the low-pass filter. The transfer function of the integrator introduced in the flux observer is: Based on this, it can be concluded that the amplitude and phase compensation functions in the embodiment of the present application are:

[0057] In other words, compensating the stator back electromotive force is to compensate for the amplitude and phase errors of the stator flux obtained by the low-pass filter compared to the integrator. Compensating the amplitude and phase errors that will be generated by introducing the low-pass filter in advance can reduce the dynamic tracking error in the flux observation, improve the system's ability to resist input signal disturbances and the system's dynamic performance.

[0058] S204, using a low-pass filter to process the compensated back electromotive force to obtain a filtered signal;

[0059] Specifically, the compensated back electromotive force e' sα 、e' sβ The low-pass filter module is used as an input signal, and the low-pass filter module can output a corresponding filtered signal. In the embodiment of the present application, the filtered signal includes the stator flux, that is, the stator flux ψ corresponding to the α and β coordinate systems can be obtained through the low-pass filter. sα , sβ .

[0060] In the embodiment of the present application, the stator back electromotive force is preferentially compensated before being input into the low-pass filter, which can reduce the dynamic tracking error in the flux observation, improve the system's ability to resist input signal disturbances and the system's dynamic performance. In other words, the compensated back electromotive force is processed by a low-pass filter to obtain the stator flux, and the stator back electromotive force is compensated before being input into the low-pass filter.

[0061] Specifically, the method of first compensating the stator back electromotive force and then inputting it into a low-pass filter to obtain the stator flux is compared with the method of inputting the stator back electromotive force into a low-pass filter to obtain the stator flux and then compensating the stator flux. After comparison, it is found that the dynamic performance of the system is poor when the low-pass filter is first used and then the amplitude and phase errors caused by the introduction of the low-pass filter are compensated. For example, when the input voltage and current signals suddenly change, the stator flux Ψ in the α and β coordinate systems is obtained. sα , sβ There will be a large error, which will lead to large fluctuations in the final rotor position. Especially when the system is running at low speed, the stator flux error and rotor position fluctuation are greater.

[0062] The specific form of the above comparison is further explained below. Before that, it should be understood that the discrete expression of the low-pass filter is:

[0063] y k =ax k +(1-a)y k-1 , (5)

[0064] In formula (5), y k is the current output of the low-pass filter, y k-1 is the output of the previous cycle of the low-pass filter, x k is the input signal of the low-pass filter, and a is the filter coefficient of the low-pass filter. From the above discrete expression, it can be seen that the low-pass filter is similar to an integrator to a certain extent, that is, the output y of the low-pass filter k Not only with the current input x k It is also related to the historical process output, that is, the output y of the previous cycle k-1 related.

[0065] The motor angular velocity is w at time t1. e1 Mutation to w e2 For example, when low-pass filtering is first performed to obtain the stator flux, the stator flux is then compensated for amplitude and phase. That is, when the motor angular velocity is w e1 The stator flux Ψ after compensation is obtained sk The expression is:

[0066] Ψ sk =(ae sk +(1-a)Ψ s(k-1) )G(W e1 ), (6)

[0067] When the motor angular velocity is w e2 The stator flux Ψ after compensation is obtained s(k+1) The expression is:

[0068] Ψ s(k+1)=(ae s(k+1) +(1-a)Ψ sk )G(w e2 ), (7)

[0069] In formula (6) and (7), e sk 、e s(k+1) They are respectively before and after the motor angular velocity changes (i.e., they correspond to the motor angular velocity w e1 、w e2 The stator back electromotive force input at the time of sk , s(k+1) are the stator flux output before and after the motor angular velocity changes, Ψ s(k-1) is the stator flux output one cycle before the motor angular velocity changes (i.e. before time t1), G(W e1 )、G(w e2 ) are the amplitude and phase compensation functions before and after the motor angular velocity changes.

[0070] It can be seen from equations (6) and (7) that the compensation function acts not only on the stator back electromotive force input at the current moment, but also on the stator flux output at the previous moment. In other words, in order to obtain the stator flux at the current moment, not only the stator back electromotive force input at the current moment should be compensated according to the motor angular velocity at the current moment, but also the stator flux output at the previous moment (i.e., the historical state of the stator flux) should be compensated according to the motor angular velocity at the current moment. Obviously, this compensation method is contrary to the physical logic of error compensation and will cause the stator flux to mutate with the sudden change of the motor angular velocity, especially when the motor is running at low speed, the stator flux is greatly affected by interference.

[0071] The embodiment of the present application is based on the physical logic of error compensation, that is, the stator flux at the current moment depends only on the stator back electromotive force input at the current moment and the stator flux output at the previous moment (that is, the historical state of the stator flux). It is proposed to first compensate the stator back electromotive force for amplitude and phase and then input it into a low-pass filter to obtain the stator flux.

[0072] Similarly, the motor angular velocity at time t1 is w e1 Mutation to w e2 For example, the stator back electromotive force is first compensated for amplitude and phase, and then input into the low-pass filter to obtain the stator flux Ψ' sk , that is, when the motor angular velocity is w e1 The stator flux Ψ' obtained when sk The expression is:

[0073] Ψ' sk =ae sk G(W e1 )+(1-a)Ψ' s(k-1) , (8)

[0074] When the motor angular velocity is w e2 The stator flux Ψ' obtained when s(k+1) The expression is:

[0075] Ψ' s(k+1) =ae s(k+1) G(w e2 )+(1-a)Ψ' sk , (9)

[0076] In formula (8) and (9), e sk 、e s(k+1) They are respectively before and after the motor angular velocity changes (i.e., they correspond to the motor angular velocity w e1 、w e2 The stator back electromotive force input at the time of sk ,Ψ' s(k+1) are the stator flux output before and after the motor angular velocity changes, Ψ' s(k-1) is the stator flux output one cycle before the motor angular velocity changes (i.e. before time t1), G(W e1 )、G(w e2 ) are the amplitude and phase compensation functions before and after the motor angular velocity changes.

[0077] It can be seen from equations (8) and (9) that when the motor angular velocity changes suddenly, the stator back electromotive force is first compensated for amplitude and phase, and then input into the low-pass filter to obtain the stator flux. Specifically, when the motor angular velocity is w e1 When the stator back electromotive force e is input at the current moment sk Compensation is performed, that is, the compensation function only acts on the stator back electromotive force e input at the current moment sk When the motor angular velocity is w e2 When the stator back electromotive force e is input at the current moment s(k+1) Compensation is performed, that is, the compensation function only acts on the stator back electromotive force e input at the current moment s(k+1) The embodiment of the present application only tracks and compensates the stator back electromotive force input after the motor angular velocity suddenly changes according to the motor angular velocity after the sudden change, and maintains the stator flux historically obtained before the motor angular velocity suddenly changes unchanged, which conforms to the physical logic of error compensation.

[0078] S205, output the motor rotor position information according to the filtered signal, and implement motor control according to the motor rotor position information. Specifically, the stator flux is converted and the corresponding algorithm is calculated to obtain the motor rotor position information (motor angular velocity w e and the rotor position θ e Etc., which will be described in detail below), thereby realizing sensorless control of the motor.

[0079] The embodiment of the present application can greatly reduce the impact of input signal mutations on the rotor position observation accuracy by preferentially compensating the stator back electromotive force and then performing filtering processing to obtain the stator flux, thereby improving the system's ability to resist input signal disturbances.

[0080] Furthermore, the embodiment of the present application obtains the stator flux by introducing a low-pass filter, which can not only suppress the integral offset, but also improve the dynamic performance of the system. In addition, by first compensating the amplitude and phase of the stator back electromotive force and then inputting the low-pass filter to obtain the stator flux, the dynamic tracking error in the flux observation caused by the sudden change of the motor angular velocity can be avoided, thereby improving the system's ability to resist input signal disturbances.

[0081] It can be understood that the embodiment of the present application obtains the stator flux link and the compensation link by exchanging the low-pass filter, which can not only be applied to the motor control method based on the low-pass filter, but also in the filtering algorithm that conforms to the summation characteristics of the ideal integrator, the embodiment of the present application can be used to first perform amplitude and phase compensation on the stator back electromotive force, and then input the low-pass filter to obtain the stator flux, so as to avoid the dynamic tracking error in the flux observation caused by the sudden change of the motor angular velocity, thereby improving the system's ability to resist input signal disturbances.

[0082] Please refer to Figure 4 and Figure 5 , Figure 4 is a flow chart of a motor control method in other embodiments of the present application, Figure 5 is a principle structural block diagram of a flux observer in some other embodiments of the present application. The motor control method may generally include the following steps:

[0083] S401, obtaining a first electrical signal from a motor stator; this step may refer to the aforementioned step S201.

[0084] S402, obtaining stator back electromotive force according to the first electrical signal; this step may refer to the aforementioned step S202.

[0085] S403, performing amplitude and phase compensation on the stator back electromotive force to obtain compensated back electromotive force; this step may refer to the aforementioned step S203.

[0086] S404, using a low-pass filter to process the compensated back electromotive force to obtain stator flux; this step may refer to the aforementioned step S204.

[0087] S405, convert the stator flux into the rotor flux of the motor; specifically, convert the stator flux ψ in the α and β coordinate systems sα , sβAccording to the corresponding relationship between the stator flux and the rotor flux, the rotor flux ψ in the α and β coordinate systems is converted rα , rβ . Then proceed to step S406.

[0088] In some embodiments of the present application, in the α and β coordinate systems, the stator flux ψ sα , sβ With the rotor flux ψ rα , rβ The corresponding relationship can be realized by the following expression:

[0089] ψ rα =ψ sα -i α Lq; (10)

[0090] ψ rβ =ψ sβ -i β Lq; (11)

[0091] In equations (10) and (11), ψ sα , sβ are the stator flux in the α and β coordinate systems, ψ rα , rβ are the rotor flux in α and β coordinate systems respectively, and Lq is the quadrature-axis inductance of the motor.

[0092] It can be seen from equations (10) and (11) that the motor quadrature-axis inductance L q The stator flux ψ sα , sβ Converted into rotor flux ψ rα , rβ , rotor flux ψ rα , rβ The motor angular velocity w can be obtained by flux normalization phase-locked loop position observer e and the rotor position θ e Among them, the rotor flux ψ rα , rβ It is further affected by the quadrature-axis inductance of the motor. Generally, when the motor is running under load, the increase in quadrature-axis current will cause the saturation degree of the quadrature-axis magnetic circuit of the motor to change. This change will affect the quadrature-axis inductance of the motor, and then affect the rotor flux, thus affecting the rotor position observation accuracy.

[0093] Based on this, the applicant further discovered in the research that in order to reduce the influence of the motor cross-axis inductance on the accuracy of the flux observer, the motor cross-axis inductance can be calibrated by a segmented calibration method under different loads using the experimental platform, and the calibration is performed in multiple intervals from 0A to the motor peak current. The intermediate current can be calibrated by a fitting method to achieve the calibration of the motor cross-axis inductance, which can greatly reduce the influence of the motor inductance on the observation performance of the flux observer.

[0094] Specifically, the quadrature-axis inductance of the motor is calibrated in advance so that the calibrated quadrature-axis inductance of the motor can be directly called when the stator flux is converted into the rotor flux. In some embodiments of the present application, the quadrature-axis inductance of the motor can be calibrated at multiple points in sections through a test platform, and then the quadrature-axis inductance of the motor can be calibrated by fitting to reduce the influence of the change in the saturation degree of the quadrature-axis magnetic circuit of the motor on the accuracy of flux observation.

[0095] Furthermore, the present application calibrates the quadrature axis inductance of the motor by using a support test platform. Prior to this, it should be understood that the voltage equation of the permanent magnet synchronous motor in the dq rotating coordinate system is as follows:

[0096]

[0097]

[0098] It can be seen from equation (12) that when the direct axis current i d When it is 0, the quadrature axis inductance L q The direct axis voltage u d With the quadrature axis current i q Based on this, the embodiment of the present application tests the given quadrature axis current i q And record the corresponding direct axis voltage u d To calibrate the quadrature axis inductance L q .

[0099] Combined with reference Figure 6 , Figure 6 1 is a flow chart of calibrating the quadrature-axis inductance Lq of a motor based on a towing test platform in some embodiments of the present application, wherein the test calibration of the quadrature-axis inductance Lq of the motor may generally include the following test steps:

[0100] S601, the support motor runs at a fixed electrical frequency fe, and drives the motor under test to run. Specifically, the towing motor rotates the simulated load at a fixed speed, and the quadrature axis current can be considered as the load current. When the quadrature axis current changes, the load current will also change accordingly, but the speed remains unchanged.

[0101] S602: Divide the cross-axis current from 0 to the maximum current value into multiple intervals. Generally speaking, the cross-axis current can be evenly divided into multiple intervals from 0 to the maximum current value. For example, multiple cross-axis current interval points 0, i q1 、i q2 、i q3 …i qn , so as to divide the cross-axis current from 0 to the maximum current value into multiple intervals (0-i q1 )、(i q1 -i q2 )、(i q2 -i q3 )…(i q(n-1) -i qn ).

[0102] S603, test platform controls direct axis current i d =0, and given multiple quadrature axis current interval points 0, i q1 、i q2 、i q3 …i qn , and record the multiple direct axis voltages u corresponding to the quadrature axis current interval points d0 、u d1 、u d2 、u d3 …u dn .

[0103] S604, through the quadrature axis inductance L q With direct axis voltage u d And the quadrature axis current i q The corresponding relationship is calculated to obtain the corresponding quadrature-axis inductance L q Specifically, the quadrature-axis inductance L can be derived from equation (12): q The expression is:

[0104] L qn =(u d0 -u dn ) / (2*π*fe*i qn ) (14)

[0105] L can be calculated in turn through formula (14): q0 , L q1 , L q2 , L q3 …L qn .

[0106] Based on this, the step of using the quadrature-axis inductance of the motor to convert the stator flux into the rotor flux in the embodiment of the present application may further include: obtaining the quadrature-axis inductance of the motor.

[0107] Furthermore, the applicant has found through research that there are roughly two ways to obtain the quadrature-axis inductance of a motor, namely, through a table lookup method and a fitting method.

[0108] Specifically, the table lookup method is to establish a correspondence between the motor cross-axis inductance and the parameters related to the motor cross-axis inductance in advance. When the motor cross-axis inductance needs to be obtained, the motor cross-axis inductance can be obtained by looking up the parameters related to the motor cross-axis inductance. This application takes the correspondence between the cross-axis current and the cross-axis inductance as an example, calibrates multiple motor cross-axis inductances through the above test method, and establishes a correspondence between the cross-axis current and the cross-axis inductance. When the cross-axis inductance needs to be called, the calibrated motor cross-axis inductance corresponding to the current cross-axis current can be directly obtained through the table lookup method according to the current cross-axis current.

[0109] In the above test results, the following corresponding relationship can be established: 0-L q0 、i q1 -L q1 、i q2 -L q2 、i q3 -L q3 …i qn -L qn , in a certain state, the motor quadrature axis current is i q3 When the motor quadrature axis inductance is L, the above corresponding relationship can be directly obtained. q3 .

[0110] The fitting method is to establish the correspondence between the motor cross-axis inductance and the parameters related to the motor cross-axis inductance in advance. When the motor cross-axis inductance needs to be obtained, the motor cross-axis inductance can be obtained by calculating through the fitting function. This application takes the correspondence between the cross-axis current and the cross-axis inductance as an example, calibrates multiple motor cross-axis inductances through the above-mentioned test method, and establishes the correspondence between the cross-axis current and the cross-axis inductance through the fitting function, that is, the motor cross-axis inductance between adjacent motor cross-axis inductance intervals can be calibrated through the fitting method. When the cross-axis inductance needs to be called, the calibrated motor cross-axis inductance corresponding to the current cross-axis current can be obtained through the fitting function adopted by the fitting method according to the current cross-axis current. In other words, the motor cross-axis inductance between adjacent motor cross-axis inductance intervals obtained by the above-mentioned test can be obtained through the fitting function adopted by the fitting method according to the current cross-axis current.

[0111] Furthermore, the quadrature-axis inductance corresponding to the quadrature-axis current between the above-mentioned multiple quadrature-axis current intervals can be obtained by fitting. In some embodiments of the present application, the quadrature-axis inductance corresponding to the quadrature-axis current between the above-mentioned multiple quadrature-axis current intervals can be calibrated by fitting. For example, the quadrature-axis inductance corresponding to the quadrature-axis current between the above-mentioned multiple quadrature-axis current intervals can be calibrated by linear fitting method or curve fitting method, and the quadrature-axis inductance corresponding to the current quadrature-axis current can be obtained by real-time calculation.

[0112] like Figure 7 As shown, Figure 7 It is a graph showing the change of the cross-axis inductance with the cross-axis current in some embodiments of the present application. The embodiments of the present application are exemplified by the linear fitting method, that is, the cross-axis inductance corresponding to the cross-axis current between the above-mentioned multiple cross-axis current intervals is calibrated by the linear function fitting method. Of course, in other embodiments, the cross-axis inductance corresponding to the cross-axis current between the above-mentioned multiple cross-axis current intervals can also be calibrated by the curve function fitting method. The embodiments of the present application are only illustrated by the linear function fitting method.

[0113] Specifically, when the quadrature axis current i q Located in the interval (i q1 -i q2 ), the corresponding quadrature-axis inductance L q Located in L q1 -L q2 The straight line between them can then be used to obtain the quadrature-axis inductance corresponding to different quadrature-axis currents through a linear difference lookup table or a linear fitting function.

[0114] It is understandable that when the number of quadrature-axis current interval divisions increases, the quadrature-axis inductance obtained by segmented calibration will be more accurate, and of course, the workload will increase accordingly. Those skilled in the art can divide the intervals according to the actual quadrature-axis inductance or the variation range of the quadrature-axis current, and the embodiments of the present application do not specifically limit this.

[0115] Furthermore, the quadrature-axis inductance is calibrated by the above test method, and the calibrated quadrature-axis inductance can be changed accordingly with the change of the load current to cope with the change of the saturation degree of the quadrature-axis magnetic circuit. In the embodiment of the present application, the calibrated quadrature-axis inductance is directly applied to the flux observer algorithm, that is, the calibrated quadrature-axis inductance is input into the stator and rotor flux equations (Formulas (10) and (11)), so as to convert the stator flux into the rotor flux through the stator and rotor flux equations, thereby improving the position observation accuracy of the flux observer.

[0116] It is understandable that since the flux observer's non-sensing algorithm requires the use of the motor's quadrature-axis inductance, the accuracy of the quadrature-axis inductance has a greater impact on the position observation accuracy. When the quadrature-axis current increases, it will cause the motor's quadrature-axis magnetic circuit saturation to change, thereby affecting the change in the motor's quadrature-axis inductance. In the actual system, the motor's quadrature-axis inductance and quadrature-axis current are calibrated accordingly, which can reduce the impact of the change in quadrature-axis inductance due to the saturation of the quadrature-axis magnetic circuit on the position observation accuracy of the flux observer.

[0117] Combining equations (10) and (11), it can be seen that the quadrature-axis inductance obtained by segmented calibration has a higher accuracy, which makes the rotor flux ψ rα , rβ It has higher accuracy, so that the flux observer has higher rotor position observation accuracy.

[0118] The motor control method provided in the embodiment of the present application may further include a step of acquiring rotor position information according to the rotor flux, that is, after step S405 is completed, the process proceeds to step S406:

[0119] S406, processing the rotor flux through a normalized phase-locked loop to obtain rotor position information;

[0120] It can be understood that the rotor flux ψ rα , rβ is the rotor flux ψ in the stationary αβ coordinate system r Component, rotor flux ψ rα , rβ The motor angular velocity w can be obtained by flux normalization phase-locked loop position observer e and the rotor position θ e , that is, the rotor flux is processed through a flux normalization phase-locked loop to obtain the rotor position information, thereby realizing sensorless vector control of the PMSM.

[0121] Furthermore, the rotor flux ψ r The expression is:

[0122] In formula (15), ψ f is the rotor permanent magnet flux amplitude of the motor. The rotor flux amplitude can be obtained by calculation.

[0123] In some embodiments of the present application, the rotor flux is used to construct a position error signal Δθ, and the position error signal Δθ is input into a proportional integral (PI) regulator to obtain rotor position information. Specifically, the position error signal Δθ can be obtained by heterodyning, and its formula is as follows:

[0124] -ψ rα sinθ e+ψ rβ cosθ e =ψ f sin(θ-θ e )≈ψ f (θ-θ e )=ψ f Δθ (16)

[0125] It can be seen from formula (16) that Δθ = θ - θ e , where θ is the theoretical position of the motor rotor, θ e The actual position of the motor rotor is obtained by constructing a position error signal Δθ and inputting the signal into a phase-locked loop to realize rotor position observation when only the position error signal Δθ is input.

[0126] Specifically, refer to Figure 8 , Figure 8 This is a principle block diagram of a flux normalization phase-locked loop in some embodiments of the present application. When the rotor flux is input into the phase-locked loop, the flux amplitude is removed. To normalize the flux linkage, the phase-locked loop input is only a scalar Δθ. Realize vectorless input and output the motor angular velocity w through the proportional integral (PI) regulator e , motor angular velocity w e Integrate to get the rotor position θ e .

[0127] Further, see Fig. 9 , Fig. 9 : is a simplified principle block diagram of a normalized phase-locked loop in some embodiments of the present application. The closed-loop transfer function of the phase-locked loop before flux normalization is:

[0128]

[0129] And, after the flux is normalized, the closed-loop transfer function of the phase-locked loop is:

[0130]

[0131] Combining equations (17) and (18), it can be seen that since the amplitude of the rotor permanent magnet flux is affected by factors such as the motor temperature, the permanent magnet flux will change during operation, which will cause the closed-loop characteristics to change with the motor temperature. The system bandwidth is a variable value. For a certain PI parameter, the system exhibits a time-varying characteristic. In actual control systems, in order to improve the observation performance, it is often hoped that the rotor position observer has a fixed bandwidth under different conditions. The flux normalization link can effectively solve the problem that the bandwidth of the above observer changes with the motor temperature and improve the dynamic performance of the system.

[0132] It can be understood that the processing method of the flux normalization phase-locked loop can transform the input position error signal from ψ f Δθ becomes Δθ, eliminating the influence of the rotor permanent magnet flux on the phase-locked loop position observer, so that the bandwidth of the phase-locked loop does not change with the change of the permanent magnet flux.

[0133] Further, by Fig. 9 It can be seen that the phase-locked loop PI parameters need to be adjusted to at least K p , K i Based on this, the embodiment of the present application further simplifies the phase-locked loop PI parameter to achieve the setting of the phase-locked loop K by setting one parameter. p and K i Two parameters.

[0134] In some embodiments of the present application, the pole configuration method is used to set the parameters of the flux normalized phase-locked loop, so that the parameter setting of the normalized phase-locked loop can be achieved by setting one parameter. Generally speaking, the two poles of the system are placed on the real axis of the left half plane of s, that is, K p =2ρ, K i =ρ 2 , so that the system has two overlapping poles (-ρ, 0). The embodiment of the present application designs the flux linkage normalized phase-locked loop PI parameters through the pole configuration method, so that in the actual control of the system, only one parameter ρ needs to be changed to achieve the K of the phase-locked loop. p and K i The adjustment of these two parameters can greatly reduce the difficulty of debugging PI parameters.

[0135] It can be understood that the PI parameter setting of the phase-locked loop is simplified so that the closed-loop transfer function of the phase-locked loop is consistent with ψ f Therefore, the setting formula of the phase-locked loop Kp and Ki is simplified, thereby reducing the difficulty of debugging the PI parameters.

[0136] The motor control method provided in the embodiment of the present application adopts a low-pass filter to replace the integrator to suppress the integral drift problem, and swaps the order of the low-pass filter link with the amplitude and phase compensation link, which can solve the integral drift problem caused by the integrator and increase the anti-disturbance capability of the sensorless system, especially the anti-low-speed disturbance capability. The setting of the phase-locked loop PI parameters is simplified, and only one parameter needs to be set to set the two parameters of the phase-locked loop Kp and Ki; the phase-locked loop position observer has a fixed bandwidth to improve the dynamic performance of the system. In addition, the motor cross-axis inductance is calibrated at multiple points using the experimental platform, and the cross-axis inductance is obtained by segmented calibration in the middle interval, and the calibrated cross-axis inductance is applied to the implementation of the flux observer algorithm, which can reduce the influence of the cross-axis inductance on the flux observer non-sensing system and increase the position observation accuracy of the flux observer non-sensing system.

[0137] It can be understood that the embodiment of the present application swaps the order of the low-pass filtering link with the amplitude and phase compensation link, which is not only applicable to the filtering link using a low-pass filter, but also to the algorithm with an ideal integral summation characteristic in the filtering link. In addition, the embodiment of the present application can effectively improve the dynamic performance of the system by swapping the order of the low-pass filtering link with the amplitude and phase compensation link.

[0138] The applicant continued to study and found that when obtaining the motor angular velocity and motor rotor position through a sliding mode observer, due to the inherent jitter phenomenon of the sliding mode observer, the estimated back electromotive force contains a large number of high-order harmonics, which affects the observation performance of the sliding mode observer.

[0139] To solve the above technical problems, the applicant found through experiments that the introduction of a low-pass filter in the sliding mode observer can filter out the high-order harmonics in the back electromotive force to reduce the system's jitter. At the same time, the amplitude and phase errors that may be generated by the low-pass filter are compensated to ensure the observation accuracy of the motor rotor position.

[0140] Based on the technical ideas for solving the above technical problems, the present application further provides a motor control method, wherein the motor control method is implemented based on a sliding mode observer. Fig.10 and Fig.11 , Fig.10 is a flow chart of a motor control method in other embodiments of the present application, Fig.11 : is a principle structural block diagram of a sliding mode observer in some other embodiments of the present application. The motor control method may generally include the following steps:

[0141] S1001, obtaining a first electrical signal from a motor stator;

[0142] The first electrical signal may include electrical signals such as stator voltage and stator current. In the embodiment of the present application, the electrical signal of the motor stator is the stator voltage u in the stationary αβ coordinate system. α 、u β and / or stator current i α 、i β .

[0143] Furthermore, the first electrical signal is input to a sliding mode observer so that the sliding mode observer can output the motor rotor position information, and then the motor control can be realized according to the motor electronic rotor position information. In other words, the first electrical signal can be obtained by the sliding mode observer, and then the motor rotor position information can be output after being processed by the sliding mode observer.

[0144] S1002, obtaining a stator back electromotive force according to the first electrical signal;

[0145] Specifically, the stator back electromotive force can be obtained by acquiring the stator voltage and / or stator current of the motor. Of course, in other embodiments, the stator back electromotive force can also be obtained by acquiring other electrical signals of the motor, which are not listed one by one in the embodiments of the present application.

[0146] In some embodiments of the present application, the second electrical signal can be calculated based on the first electrical signal, and the difference between the first electrical signal and the second electrical signal can be converted into the stator back electromotive force using a saturation function. For example, the stator voltage u in the stationary αβ coordinate system can be first obtained using a sliding mode observer. α 、u β and / or stator current i α 、i β (i.e., the first electrical signal), and recalculate the estimated stator current in the stationary αβ coordinate system according to the stator voltage equation of the PMSM (i.e., the second electrical signal). Among them, the stator voltage equation of PMSM is as follows:

[0147]

[0148] In the formula, are the estimated α and β axis stator currents, u α 、u β are the α and β axis stator voltages, e α 、e β are the compensated α and β axis back electromotive force, respectively, α 、z β are the α and β axis back electromotive force before filtering, L d , L q They are the orthogonal axis inductance of the motor, R s is the stator resistance of the motor, w e is the estimated motor angular velocity.

[0149] The estimated α and β axis stator currents are converted into and the actual α and β axis stator current i α 、i β The difference is converted into the α and β axis back electromotive force components before filtering, that is, the stator back electromotive force z of the motor α 、z β Among them, the expression of sat saturation function is:

[0150] Where k is the sliding mode gain, which is a positive constant.

[0151] Among them, the stator voltage u α 、u β The AC and DC voltages output by the above current loop can be obtained by inverse Park transformation, and the stator current iα 、i β The three-phase current i of the above motor can be used a 、i b 、i c Of course, in other embodiments, the stator voltage u can also be obtained by directly obtaining α 、u β and the stator current i α 、i β .

[0152] Combined with reference Fig.12 , Fig.12 Schematic diagram of the saturation function sat() in some other embodiments of the present application, where ε in the figure is the sliding mode boundary and is a positive constant. The saturation function sat() can avoid sudden changes in output around zero like the sign function, which aggravates the system jitter of the sliding mode observer. The output value of the saturation function sat() changes linearly between -ε and ε, avoiding sudden changes in output and reducing the jitter phenomenon of the sliding mode observer.

[0153] S1003, performing amplitude and phase compensation on the stator back electromotive force to obtain compensated back electromotive force;

[0154] Specifically, the sliding mode observer can be used to calculate the stator back electromotive force z α 、z β Compensation is performed and the compensated back electromotive force e′ is obtained α , e′ β In the embodiment of the present application, amplitude and phase compensation are performed before low-pass filtering of the back electromotive force, and the expression of the compensation function is:

[0155] In the formula, w e is the motor angular velocity, w c is the cutoff frequency of the low-pass filter. The transfer function of the low-pass filter introduced in the sliding mode observer is: In the formula, w c is the cut-off frequency of the low-pass filter. That is, the expression of the α and β axis back electromotive force after the amplitude and phase compensation link is:

[0156]

[0157] Among them, the low-pass filter can filter out the high-order harmonics in the back electromotive force and reduce the chattering of the sliding mode control system. From the transfer function of the low-pass filter, it can be seen that the introduction of the low-pass filter will attenuate the amplitude of the input signal to the value when it is not filtered. Phase lag unfiltered Based on this, the amplitude attenuation and phase lag caused by the low-pass filter can be compensated. After the amplitude and phase compensation, the position estimation of the motor rotor becomes more accurate, which can make up for the phase delay caused by the low-pass filter and improve the control accuracy and operation efficiency of the sliding mode observer. In order to further improve the dynamic performance of the sliding mode observer, the present application swaps the order of the low-pass filter link and the amplitude and phase compensation link, that is, the amplitude and phase compensation are performed first, and then the low-pass filter is performed, which greatly reduces the impact of the sudden change of the input signal on the accuracy of the rotor position observation and improves the dynamic performance of the system. It can be understood that the amplitude and phase compensation of the stator back electromotive force are performed, and the amplitude and phase compensation include amplitude attenuation compensation and phase advance compensation.

[0158] S1004, using a low-pass filter to process the compensated back electromotive force to obtain a filtered back electromotive force;

[0159] Specifically, the compensated back electromotive force is input as an input signal to the low-pass filter module, and the low-pass filter module can output a corresponding filtered signal. In the embodiment of the present application, the filtered signal may include the filtered back electromotive force, that is, the filtered back electromotive force can be obtained by the low-pass filter. α 、e β The expression is:

[0160] In the embodiment of the present application, the stator back electromotive force is preferentially compensated before being input into the low-pass filter, which can reduce the dynamic tracking error in the sliding mode observation, improve the system's ability to resist input signal disturbances and the system's dynamic performance. In other words, the compensated back electromotive force is processed by a low-pass filter to obtain a filtered back electromotive force, and the stator back electromotive force is compensated before being input into the low-pass filter.

[0161] Specifically, the method of first compensating the stator back EMF and then inputting it into a low-pass filter to obtain the filtered back EMF is compared with the method of inputting the stator back EMF into a low-pass filter and then compensating. After comparison, it is found that when a low-pass filter is first used and then the amplitude and phase errors caused by the introduction of the low-pass filter are compensated, the system's dynamic performance and anti-disturbance ability are poor, which is particularly evident in the low-speed dynamic operation performance.

[0162] In the embodiment of the present application, the compensation function acts not only on the stator back electromotive force input at the current moment, but also on the stator back electromotive force output at the previous moment. In other words, in order to obtain the stator back electromotive force at the current moment, not only the stator back electromotive force input at the current moment is compensated according to the motor angular velocity at the current moment, but also the stator back electromotive force output at the previous moment (i.e., the historical state of the stator back electromotive force) is compensated according to the motor angular velocity at the current moment. Obviously, this compensation method is contrary to the physical logic of error compensation, and will cause the stator back electromotive force to mutate with the sudden change of the motor angular velocity, especially when the motor is running at low speed, the stator back electromotive force is greatly affected by interference.

[0163] The embodiment of the present application is based on the physical logic of error compensation, that is, the stator back electromotive force at the current moment only depends on the stator back electromotive force input at the current moment and the stator back electromotive force output at the previous moment (that is, the historical state of the stator magnetic flux). It is proposed to first perform amplitude and phase compensation on the stator back electromotive force and then input it into a low-pass filter to obtain the filtered stator back electromotive force.

[0164] It is understandable that when the motor angular velocity suddenly changes, the stator back electromotive force is first compensated for amplitude and phase, and then input into the low-pass filter to obtain the filtered stator back electromotive force. The embodiment of the present application only tracks and compensates the stator back electromotive force input after the motor angular velocity suddenly changes according to the motor angular velocity after the sudden change, and maintains the stator back electromotive force historically obtained before the motor angular velocity suddenly changes, which conforms to the physical logic of error compensation.

[0165] S1005, outputting motor rotor position information according to the filtered back electromotive force, and realizing motor control according to the motor rotor position information. Specifically, the motor rotor position information (motor angular velocity and rotor position, etc., which will be described in detail below) can be obtained by converting the filtered back electromotive force and performing corresponding algorithm calculation, so as to realize motor sensorless control.

[0166] The embodiment of the present application can greatly reduce the impact of sudden changes in the input signal on the accuracy of rotor position observation by preferentially compensating the stator back electromotive force and then filtering to obtain the filtered back electromotive force, thereby improving the system's ability to resist input signal disturbances.

[0167] Furthermore, the embodiments of the present application can avoid dynamic tracking errors in flux observation caused by sudden changes in motor angular velocity by preferentially performing amplitude and phase compensation on the stator back electromotive force and then inputting the compensated back electromotive force into a low-pass filter to obtain the filtered back electromotive force, thereby improving the system's ability to resist input signal disturbances.

[0168] It can be understood that the embodiment of the present application swaps the order of the low-pass filtering link with the amplitude and phase compensation link, which is not only applicable to the filtering link using a low-pass filter, but also to the algorithm with an ideal integral summation characteristic in the filtering link. In addition, the embodiment of the present application can effectively improve the dynamic performance of the system by swapping the order of the low-pass filtering link with the amplitude and phase compensation link. In addition, the scheme of compensating first and then filtering is not only applicable to the sliding mode observer in the embodiment of the present application, but other algorithms can also exchange the order of the filtering link and the compensation link when compensating for the amplitude and phase errors caused by filtering, thereby increasing the dynamic performance of the system.

[0169] In the embodiment of the present application, since the sliding mode observer's sensorless algorithm needs to use the motor's quadrature-axis inductance, the accuracy of the quadrature-axis inductance has a greater impact on the position observation accuracy. When the quadrature-axis current increases, it will cause a change in the degree of saturation of the motor's quadrature-axis magnetic circuit, thereby affecting the change in the motor's quadrature-axis inductance. In an actual system, the motor's quadrature-axis inductance and quadrature-axis current are calibrated accordingly, which can reduce the impact of the change in the quadrature-axis inductance due to the saturation of the quadrature-axis magnetic circuit on the position observation accuracy of the flux observer. For example, the quadrature-axis inductance that can be obtained by means of segmented calibration has a higher accuracy, so that the sliding mode observer has a higher rotor position observation accuracy. It can be understood that the calibration method of the motor's quadrature-axis inductance in the embodiment of the present application can refer to the specific description in the embodiment of the flux observer, so the embodiment of the present application will not be repeated.

[0170] See also Fig.13 , Fig.13 It is a principle block diagram of a back-electromotive force normalized phase-locked loop in other embodiments of the present application, wherein, in step S1005, outputting the motor rotor position information according to the filtered back-electromotive force further includes: processing the filtered back-electromotive force through a normalized phase-locked loop to obtain the rotor position information.

[0171] It can be understood that the filtered back EMF e α 、e β is the component of the stator back electromotive force in the stationary αβ coordinate system, and the filtered back electromotive force e α 、e β The motor angular velocity w can be obtained by normalizing the phase-locked loop position observer through the back electromotive force e and the rotor position θ e , that is, the filtered back-EMF is processed through a back-EMF normalization phase-locked loop to obtain the rotor position information, thereby realizing sensorless vector control of PMSM.

[0172] Furthermore, the expression of stator back electromotive force is: Where E is the vector magnitude of the back EMF and θ is the actual rotor angle of the motor.

[0173] In some embodiments of the present application, the position error signal Δθ is constructed using the stator back electromotive force, and the position error signal Δθ is input into a proportional integral (PI) regulator to obtain the rotor position information. Specifically, the position error signal Δθ can be obtained by heterodyning, and its formula is as follows: α cosθ e -e β sinθ e = Esin(θ-θ e )=EΔθ, where Δθ is the difference between the actual rotor angle and the estimated angle. The back electromotive force amplitude is calculated by When the stator back electromotive force is input into the phase-locked loop, the back electromotive force amplitude E is removed so that the phase-locked loop input is only Δθ, so as to perform back electromotive force normalization processing, thereby making the phase-locked loop input only a scalar. That is, construct Realize vectorless input and output the motor angular velocity w through the proportional integral (PI) regulator e , motor angular velocity w e Integrate to get the rotor position θ e .

[0174] Further, see Fig. 9 , Fig. 9 : is a simplified principle block diagram of a normalized phase-locked loop in some embodiments of the present application. Among them, the closed-loop transfer function of the phase-locked loop before the back electromotive force is normalized is: And, after the back EMF is normalized, the closed-loop transfer function of the phase-locked loop is:

[0175] From the normalized phase-locked loop transfer function, we can see that since the back-EMF amplitude changes with the speed, the closed-loop characteristics change with the motor speed, and the system bandwidth is a variable value. For a certain PI parameter, the system exhibits a time-varying characteristic. In actual control systems, in order to improve the observation performance, it is often hoped that the rotor position observer has a fixed bandwidth under different conditions. The back-EMF normalization link can effectively solve the problem that the bandwidth of the above observer changes with the motor temperature and improve the dynamic performance of the system.

[0176] It can be understood that the processing method of the back-EMF normalization phase-locked loop can change the input position error signal from EΔθ to Δθ, eliminate the influence of the back-EMF amplitude on the phase-locked loop position observer, and make the bandwidth of the phase-locked loop not change with the change of the back-EMF amplitude.

[0177] Further, by Fig. 9 It can be seen that the phase-locked loop PI parameters need to be adjusted to at least K p , K iBased on this, the embodiment of the present application further simplifies the phase-locked loop PI parameter to achieve the setting of the phase-locked loop K by setting one parameter. p and K i Two parameters.

[0178] In some embodiments of the present application, the parameters of the back electromotive force normalized phase-locked loop are set by using the pole configuration method, so that the parameter setting of the normalized phase-locked loop can be achieved by setting one parameter. Generally speaking, the two poles of the system are placed on the real axis of the left half plane of s, that is, K p =2ρ, K i =ρ 2 , so that the system has two overlapping poles (-ρ, 0). The embodiment of the present application designs the back electromotive force normalization phase-locked loop PI parameters through the pole configuration method, so that in the actual control of the system, only one parameter ρ needs to be changed to achieve the K of the phase-locked loop. p and K i The adjustment of these two parameters can greatly reduce the difficulty of debugging PI parameters.

[0179] It can be understood that simplifying the PI parameter setting of the phase-locked loop makes the closed-loop transfer function of the phase-locked loop independent of E, thereby simplifying the setting formulas of the phase-locked loop Kp and Ki, thereby reducing the difficulty of debugging the PI parameters.

[0180] The motor control method provided in the embodiment of the present application can increase the observation accuracy of the sliding mode observer angle, improve the operating efficiency and load capacity of the system, and is easy to implement in engineering. It can also increase the dynamic performance of the system and greatly reduce the impact of sudden changes in input signals on position observation. The setting of the phase-locked loop PI parameters is simplified. Only one parameter needs to be set to set the two parameters of the phase-locked loop Kp and Ki, so that the phase-locked loop position observer has a fixed bandwidth and improves the dynamic performance of the system.

[0181] In addition, the present application embodiment also provides a motor 14, see Fig.14 , Fig.14 1 is a schematic block diagram of the structure of the motor 10 in some embodiments of the present application, and the motor 10 may generally include a stator 11, a rotor 12 and a controller 13. The controller 13 is used to control the motor, for example, to control the start, acceleration, operation, deceleration and stop of the motor.

[0182] Specifically, the stator 11 and the rotor 12 are respectively used to be electrically connected to the controller 13 so that the controller 13 can execute the motor control method described in the above embodiment.

[0183] The motor provided in the embodiment of the present application obtains the motor rotor position information through a flux observer or a sliding mode observer to achieve sensorless control of the motor.

[0184] In addition, the present application also provides a motor control system, see Fig.15 , Fig.15 3 is a schematic block diagram of a motor control system in some embodiments of the present application. The system 30 may generally include:

[0185] An acquisition unit 31 is used to acquire a first electrical signal of a motor stator;

[0186] A computing unit 32, configured to obtain a stator back electromotive force according to the first electrical signal;

[0187] A compensation unit 33, used for compensating the stator back electromotive force to obtain a compensated back electromotive force;

[0188] The processing unit 34 is used to process the compensated back electromotive force to obtain a filtered signal, and output the motor rotor position information according to the filtered signal;

[0189] The control unit 35 is used to implement motor control according to the motor rotor position information.

[0190] It can be understood that the acquisition unit 31, the operation unit 32, the compensation unit 33, the processing unit 34 and the control unit 35 in the embodiment of the present application correspond to the relevant steps in the aforementioned method embodiment, and those skilled in the art can implement the corresponding functions in the aforementioned method embodiment by using the system in the embodiment of the present application. In the embodiment of the present application, the filtered signal can be the stator flux or the filtered back electromotive force in the aforementioned embodiment.

[0191] The present application embodiment also provides a computer-readable storage medium 300. Specifically, see Fig.16 , Fig.16 3 is a schematic diagram of the structure of a computer-readable storage medium 300 in some embodiments of the present application, wherein the computer-readable storage medium 300 stores a computer program 301 for electronic data exchange, and the computer program 301 enables a computer to execute some or all steps of any method recorded in the above method embodiments. It can be understood that the above computer includes an electronic device. In the above embodiments, the description of each embodiment has its own emphasis. For parts that are not described in detail in a certain embodiment, please refer to the relevant description of other embodiments.

[0192] Those skilled in the art will appreciate that all or part of the steps in the various methods of the above embodiments can be completed by instructing the relevant hardware through a program, and the program can be stored in a computer-readable storage medium. The computer-readable storage medium may include: a USB flash drive, a read-only memory (ROM), a random access memory (RAM), a magnetic disk or an optical disk, etc.

[0193] It is understandable that in the above embodiments, it can be implemented in whole or in part by software, hardware, firmware or any combination thereof. When the use is implemented in whole or in part in the form of a computer program product, the computer program product includes one or more computer instructions. The computer can be a general-purpose computer, a special-purpose computer, a computer network, or other programmable devices. The computer instructions can be stored in a computer-readable storage medium, or transmitted from one computer-readable storage medium to another computer-readable storage medium. The computer-readable storage medium can be any available medium that the computer can access or a data storage device such as a server or a data center that includes one or more available media integration. The available medium can be a magnetic medium, (e.g., a floppy disk, a hard disk, a tape), an optical medium (e.g., a DVD), or a semiconductor medium (e.g., a solid-state hard disk SolidState Disk (SSD)), etc.

[0194] It should be noted that the terms "include" and "have" and any variations thereof are intended to cover non-exclusive inclusions. For example, a process, method, system, product or device comprising a series of steps or units is not limited to the listed steps or units, but may optionally include steps or units not listed, or may optionally include other steps or units inherent to these processes, methods, products or devices.

[0195] The above descriptions are only some embodiments of the present application, and do not limit the protection scope of the present application. Any equivalent device or equivalent process transformation made using the contents of the present application specification and drawings, or directly or indirectly used in other related technical fields, are also included in the patent protection scope of the present application.

Claims

1. A motor control method, characterized in that: include: Acquiring a first electrical signal from a motor stator; Obtaining a stator back electromotive force according to the first electrical signal; Performing amplitude and phase compensation on the stator back electromotive force to obtain compensated back electromotive force, and inputting the compensated back electromotive force into a low-pass filter as an input signal; The compensated back electromotive force is processed by the low-pass filter to obtain a stator flux, and the stator flux is converted into a rotor flux of the motor by using the motor quadrature-axis inductance; Processing the rotor flux through a normalized phase-locked loop to obtain the rotor position information, and implementing motor control according to the motor rotor position information; wherein the motor rotor position information includes the motor angular velocity and the rotor position; Wherein, the transfer function of the low-pass filter is: The amplitude and phase compensation functions for performing amplitude and phase compensation are: In the formula, w c is the cut-off frequency of the low-pass filter, w e is the motor angular velocity; Calibrate the quadrature-axis inductance of the motor by using a segmented calibration method; The rotor flux is subjected to normalized phase-locked loop processing, including: constructing a position error signal using the rotor flux, removing the flux amplitude when inputting the rotor flux into the phase-locked loop, and inputting the position error signal into a regulator to obtain rotor position information.

2. The motor control method according to claim 1, characterized in that: The step of outputting the motor rotor position information according to the rotor flux comprises: The rotor flux is processed through a normalized phase-locked loop to obtain the rotor position information.

3. The motor control method according to claim 1, characterized in that: The step of converting the stator flux into the rotor flux using the motor quadrature-axis inductance comprises: calibrating a plurality of the motor quadrature-axis inductances, and obtaining the calibrated motor quadrature-axis inductances by a table lookup method.

4. The motor control method according to claim 1, characterized in that: The step of converting the stator flux into the rotor flux using the motor quadrature-axis inductance includes: calibrating a plurality of the motor quadrature-axis inductances, calibrating the motor quadrature-axis inductances between adjacent motor quadrature-axis inductance intervals by a fitting method, and further obtaining the motor quadrature-axis inductances between adjacent motor quadrature-axis inductance intervals by a fitting function.

5. The motor control method according to claim 1, characterized in that: The parameters of the normalized phase-locked loop processing are set using a pole configuration method, so that the parameter tuning of the normalized phase-locked loop processing can be achieved by setting one parameter.

6. A motor, characterized in that: It comprises a stator, a rotor and a controller, wherein the stator and the rotor are respectively used to be electrically connected to the controller so that the controller can execute the motor control method described in any one of claims 1-5.

7. A motor control system, characterized in that: include: An acquisition unit, used for acquiring a first electrical signal from a stator of a motor; A computing unit, used for obtaining a stator back electromotive force according to the first electrical signal; A compensation unit, used for compensating the stator back electromotive force to obtain a compensated back electromotive force; A processing unit, used for processing the compensated back electromotive force to obtain a filtered signal, and outputting motor rotor position information according to the filtered signal; A control unit, used to control the motor according to the motor rotor position information; Wherein, the motor control system is used to implement the motor control method described in any one of claims 1-5.

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