Power quality improvement method and system for eliminating modulation of multiple zero-crossing drive vacancy zones

Abstract

The present invention belongs to the field of power electronics technology, and relates to a method and system for improving power quality by eliminating modulation of multiple zero-crossing drive vacancy areas. The method comprises the following steps: two-phase static coordinate transformation of three-phase grid voltage; calculating the phase angle relationship of grid voltage; performing two-phase rotating coordinate transformation on three-phase distorted current and three-phase output current according to the phase angle relationship; obtaining two-phase command current according to two-phase distorted current and two-phase sinusoidal current obtained by filtering the two-phase distorted current; obtaining two-phase command voltage according to DC side reference voltage and actual voltage, two-phase command current and two-phase output current; inverse coordinate transformation of two-phase command current and two-phase command voltage according to the phase angle relationship; and obtaining two-phase command voltage according to the command voltage. 、 The carrier period and the actual voltage on the DC side are used to obtain the initial modulation wave; the driving signal is generated according to the size of the final modulation wave and the positive load wave obtained by adjusting the initial modulation wave. The present invention effectively eliminates the driving vacancy area in the modulation, reduces the harmonic distortion rate, and improves the power quality.

Description

Technical Field

[0001] The present invention belongs to the field of power electronics technology, and relates to inverter modulation technology, and in particular to a method and system for improving power quality by multi-zero-point drive vacancy elimination modulation. Background Art

[0002] Inverters are important power equipment in power systems and are widely used in the fields of renewable energy generation, harmonic compensation, and electric traction. If the modulation and control strategies used are inappropriate, a large amount of harmonic current will be generated. When the harmonic current content in the power system is large and under weak grid conditions, it will further cause grid voltage distortion and seriously affect the power supply quality of the power system. If the power supply quality of the power system decreases, it will further cause electrical equipment to generate more harmonic currents, forming a vicious circle. Therefore, it is necessary to effectively control the inverter output current harmonics to purify the grid "environment".

[0003] Traditional silicon (Si) material power switching devices have low withstand voltage, slow switching speed, and large switching loss, which seriously limits the further improvement of inverter performance. New silicon carbide (SiC) material power switching devices have higher withstand voltage, faster switching speed, and lower switching loss. New silicon carbide metal-oxide semiconductor field effect transistor (English: Silicon Carbide Metal-Oxide-Semiconductor Field-Effect Transistor, referred to as: SiC MOSFET) power switching devices can be used to improve inverter performance. However, when the SiC MOSFET power switching device is at a high switching frequency, the influence of the drive vacancy area in the modulation cannot be ignored, which will increase the current harmonics of the inverter output. Therefore, it is necessary to try to eliminate the influence of the drive vacancy area in the modulation to reduce the harmonic current.

[0004] In traditional renewable energy power generation scenarios, the current output from the AC side of the inverter has good sinusoidality and contains only two zero crossings in a fundamental cycle, so it is relatively easy to eliminate the influence of the drive vacancy zone in the inverter modulation. However, when the harmonic current content in the system is large, the current distortion is serious, and the current contains multiple zero crossings in a fundamental cycle, which greatly increases the difficulty of eliminating the influence of the drive vacancy zone in the modulation. It is urgent to take effective measures to propose effective solutions to this problem, so as to effectively eliminate the influence of the drive vacancy zone in the modulation on the grid current waveform, reduce the harmonic distortion rate, and improve the power quality. Summary of the invention

[0005] In view of the problems existing in the prior art, the present invention provides a method and system for improving the power quality of multi-zero-crossing driving vacancy area elimination modulation, which can effectively eliminate the influence of the driving vacancy area in the modulation on the grid current waveform when the inverter output current contains multiple zero-crossing points, reduce the harmonic distortion rate, and improve the power quality. At the same time, when an error occurs in the given current polarity, the dependence of the modulation strategy on the current polarity can be weakened, and the degree of distortion of the grid current when an error occurs in the given current polarity can be reduced.

[0006] In a first aspect, the present invention provides a method for improving power quality by eliminating modulation of multiple zero-crossing drive vacancy regions, the specific steps of which are as follows:

[0007] Grid voltage coordinate transformation steps: transform the three-phase grid voltage into a two-phase grid voltage in a two-phase stationary coordinate system;

[0008] Trigonometric relationship calculation steps: Calculate the phase angle relationship of the grid voltage based on the two-phase grid voltage;

[0009] Distortion current coordinate transformation steps: transform the three-phase distortion current into a two-phase distortion current in a two-phase rotating coordinate system according to the phase angle relationship;

[0010] Current filtering steps: eliminate the high-frequency current ripple in the two-phase distorted current to obtain the two-phase sinusoidal current, and obtain the two-phase command current according to the two-phase distorted current and the two-phase sinusoidal current;

[0011] Output current coordinate transformation steps: convert the three-phase output current into a two-phase output current in a two-phase rotating coordinate system according to the phase angle relationship;

[0012] Command voltage generation steps: DC side reference voltage V dc * Subtract the actual DC side voltage V dc The difference is multiplied by the voltage proportional integral coefficient K PIV An active command current is obtained, and a two-phase command voltage is obtained according to the active command current, the two-phase command current and the two-phase output current;

[0013] The command current reverse coordinate transformation steps are: converting the two-phase command current into the three-phase command current in the three-phase stationary coordinate system according to the phase angle relationship;

[0014] The command voltage reverse coordinate transformation step: convert the two-phase command voltage into a three-phase command voltage in a three-phase stationary coordinate system according to the phase angle relationship;

[0015] Modulation wave generation steps: i-phase command voltage u i * Multiply by the carrier period Ts Divide by the actual DC voltage V dc , minus the i-phase command voltage u i * Multiply by the sign function sign and then multiply by the carrier period T s Divide by 4 to get the first modulation wave u i1 * , the sum of the maximum value of the first modulation wave plus the minimum value of the first modulation wave multiplied by 0.5 to obtain the second modulation wave u i2 * , i-phase command voltage u i * Multiply by the carrier period T s Divide by the actual DC voltage V dc Subtract the second modulation wave u i2 * Get the initial modulation wave of phase i m i * , i=A,B,C;

[0016] Modulation wave adjustment steps: Set the first adjustment amount equal to the dead time, when the i-phase command current i hi >0, the second adjustment amount is equal to the first adjustment amount. When the i-phase command current i hi ≤0, the second adjustment amount is equal to -1 times of the first adjustment amount; the initial modulation wave of phase i m i * Add the first adjustment amount Δ u d of – 1 times to get the down-shifted modulation wave, i-phase initial modulation wave m i * Add the first adjustment amount to get the up-shifted modulation wave, and add the second adjustment amount to get the final modulation wave of phase i. m i12 * , the up-shifted modulation wave plus the second adjustment amount is obtained to obtain the final modulation wave of phase i m i34 * ;

[0017] Driving signal generation steps: According to the final modulation wave of phase i m i12* , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i1 , according to the final modulation wave of phase i m i12 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i2 , according to the final modulation wave of phase i m i34 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i3 , according to the final modulation wave of phase i m i34 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i4 .

[0018] In some embodiments, in the grid voltage coordinate transformation step, a method for transforming the three-phase grid voltage into a two-phase grid voltage in a two-phase stationary coordinate system is:

[0019] A phase grid voltage u gA 2 times minus the B phase grid voltage u gB and C phase grid voltage u gC Then, multiply by 0.333 to get the grid voltage in the two-phase stationary coordinate system. u gα ;

[0020] Phase B grid voltage u gB Subtract C phase grid voltage u gC Then, multiply by 0.577 to get the grid voltage in the two-phase stationary coordinate system. u gβ .

[0021] In some embodiments, in the trigonometric relationship calculation, the method for calculating the phase angle relationship of the grid voltage according to the two-phase grid voltage is: u gα Divide by the grid voltage ugα The square of the grid voltage u gβ The square root of the sum of the squares of the grid is the cosine function cosθ of the grid phase angle; the grid voltage u gβ Divide by the grid voltage u gα The square of the grid voltage u gβ The square root of the sum of the squares of the grid is the sinusoidal function sinθ of the grid phase angle.

[0022] In some embodiments, in the distortion current coordinate transformation step, a method for converting the three-phase distortion current into a two-phase distortion current in a two-phase rotating coordinate system according to the phase angle relationship is:

[0023] The difference between 0.707 times the negative sine function sinθ and 0.408 times the cosine function cosθ multiplied by the C phase distortion current i lC The product of the sine function sinθ (0.707 times) and the cosine function cosθ (0.408 times) multiplied by the B-phase distortion current i lB The product of the cosine function cosθ times 0.816 times the A phase distortion current i lA The product of the two phases is the distortion current in the rotating coordinate system. i ld ;

[0024] The sum of 0.408 times the sine function sinθ and 0.707 times the cosine function cosθ multiplied by the B-phase distortion current i lB The product of the sine function sinθ minus 0.816 times multiplied by the A phase distortion current i lA The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the C phase distortion current i lC The product of the two phases is the distortion current in the rotating coordinate system. i lq .

[0025] In some embodiments, in the output current coordinate transformation step, a method for converting the three-phase output current into a two-phase output current in a two-phase rotating coordinate system according to the phase angle relationship is:

[0026] The difference between 0.707 times the negative sine function sinθ and 0.408 times the cosine function cosθ multiplied by the C phase output current i fCThe product of the sine function sinθ times 0.707 and the cosine function cosθ times 0.408 times multiplied by the B phase output current i fB The product of the cosine function cosθ times 0.816 times the output current of phase A i fA The product of the distorted current and the output current in the two-phase rotating coordinate system is obtained. i fd ;

[0027] The sum of 0.408 times the sine function sinθ and 0.707 times the cosine function cosθ multiplied by the B phase output current i fB The product of the sine function sinθ minus 0.816 times multiplied by the A phase output current i fA The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the C phase output current i fC The product of the distorted current and the output current in the two-phase rotating coordinate system is obtained. i fq .

[0028] In some embodiments, in the command current inverse coordinate transformation step, the method of converting the two-phase command current into the three-phase command current in the three-phase stationary coordinate system according to the phase angle relationship is:

[0029] Cosine function cosθ multiplied by command current i hd The product of plus the sine function sinθ multiplied by the command current i hq The product of the two-phase command currents is the A-phase command current in the three-phase stationary coordinate system. i hA ;

[0030] The difference between 0.707 times the sine function sinθ and 0.408 times the cosine function cosθ multiplied by the command current i hd The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the command current i hq The product of the two-phase command currents is obtained as the B-phase command current in the three-phase stationary coordinate system. i hB ;

[0031] The difference between 0.707 times the negative sine function sinθ and 0.408 times the cosine function cosθ multiplied by the command current command current ihd The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the command current i hq The product of the two-phase command currents is the C-phase command current in the three-phase stationary coordinate system. i hC .

[0032] In some embodiments, in the command voltage inverse coordinate transformation step, the method of converting the two-phase command voltage into the three-phase command voltage in the three-phase stationary coordinate system according to the phase angle relationship is:

[0033] Cosine function cosθ multiplied by command voltage u d * The product of plus the sine function sinθ multiplied by the command voltage u q * The product of the two-phase command voltages is the A-phase command voltage in the three-phase stationary coordinate system. u A * ;

[0034] The difference between 0.707 times the sine function sinθ and 0.408 times the cosine function cosθ multiplied by the command voltage u d * The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the command voltage u q * The product of the two-phase command voltages is obtained as the B-phase command voltage in the three-phase stationary coordinate system. u B * ;

[0035] The difference between 0.707 times the negative sine function sinθ and 0.408 times the cosine function cosθ multiplied by the command voltage u d * The product of the sine function sinθ and the cosine function cosθ multiplied by the command voltage u q * The product of the two-phase command voltages is obtained as the C-phase command voltage in the three-phase stationary coordinate system. u C * .

[0036] In some embodiments, in the current filtering step, the method for obtaining the two-phase command current according to the two-phase distorted current and the two-phase sinusoidal current is: i ld Subtract the sinusoidal current i ld * Get the command current i hd , distortion current i lq Subtract the sinusoidal current i lq * Get the command current i hq .

[0037] In some embodiments, in the command voltage generating step, a method for obtaining the two-phase command voltage according to the active command current, the two-phase command current and the two-phase output current is:

[0038] Active command current i dc With command current i hd The output current is subtracted from the sum of i fd The difference is multiplied by the current proportional integral coefficient K PII Get the command voltage u d * ;

[0039] Command current i hq Subtract output current i fq The difference is multiplied by the current proportional integral coefficient K PII Get the command voltage u q * ;

[0040] Voltage proportional integral coefficient K PIV =0.3+100 / s , current proportional integral coefficient K PII =5+1500 / s , s is a complex variable in the frequency domain, representing the frequency.

[0041] In some embodiments, in the step of generating the driving signal, the final modulation wave of phase i is m i12 * , positive carrier uc1 * The magnitude of the i-phase drive signal is generated S i1 The method is: m i12 * > u c1 * hour, S i1 is high level, m i12 * ≤ u c1 * hour, S i1 is low level;

[0042] According to the final modulation wave of phase i m i12 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i2 The method is: m i12 * > u c2 * hour, S i2 is high level, m i12 * ≤ u c2 * hour, S i2 is low level;

[0043] According to the final modulation wave of phase i m i34 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i3 The method is: m i34 * > u c1 * hour, S i3 is high level, m i34 * ≤ uc1 * hour, S i3 is low level;

[0044] According to the final modulation wave of phase i m i34 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i4 The method is: m i34 * > u c2 * hour, S i4 When high level m i34 * ≤ u c2 * hour, S i4 is low level.

[0045] In a second aspect of the present invention, a system for improving power quality by eliminating modulation in a vacancy zone with multiple zero-crossing driving is provided, which is used to implement the method for improving power quality by eliminating modulation in a vacancy zone with multiple zero-crossing driving as described in the first aspect of the present invention, comprising:

[0046] A grid voltage coordinate transformation module transforms the three-phase grid voltage into a two-phase grid voltage in a two-phase stationary coordinate system;

[0047] A trigonometric relationship calculation module calculates the phase angle relationship of the grid voltage based on the two-phase grid voltage;

[0048] The distortion current coordinate transformation module converts the three-phase distortion current into a two-phase distortion current in a two-phase rotating coordinate system according to the phase angle relationship;

[0049] The current filter module eliminates the high-frequency current ripple in the two-phase distorted current to obtain the two-phase sinusoidal current, and obtains the two-phase command current according to the two-phase distorted current and the two-phase sinusoidal current;

[0050] An output current coordinate transformation module converts the three-phase output current into a two-phase output current in a two-phase rotating coordinate system according to the phase angle relationship;

[0051] The command voltage generation module generates the voltage according to the DC side reference voltage, the actual DC side voltage and the voltage proportional integral coefficient. K PIVAn active command current is obtained, and a two-phase command voltage is obtained according to the active command current, the two-phase command current and the two-phase output current;

[0052] The command current inverse coordinate transformation module converts the two-phase command current into the three-phase command current in the three-phase stationary coordinate system according to the phase angle relationship;

[0053] The command voltage inverse coordinate transformation module converts the two-phase command voltage into a three-phase command voltage in a three-phase stationary coordinate system according to the phase angle relationship;

[0054] The modulation wave generation module generates an initial modulation wave of the i-phase according to the i-phase command voltage, the carrier cycle, and the actual voltage on the DC side;

[0055] A modulation wave adjustment module, adjusting the initial modulation wave of phase i according to the first adjustment amount and the second adjustment amount to obtain a final modulation wave of phase i;

[0056] The driving signal generating module generates the i-phase driving signal according to the magnitude of the i-phase final modulation wave, the positive carrier wave and the load wave.

[0057] In some embodiments, the current filtering module includes:

[0058] The filter is used to eliminate the high-frequency current ripple in the two-phase distorted current to obtain a two-phase sinusoidal current. The transfer function of the filter is expressed as:

[0059]

[0060] In the formula, L ( ω ) is the transfer function of the filter, s is a complex variable in the frequency domain, representing the frequency;

[0061] The first instruction current generating module converts the distorted current i ld Subtract the sinusoidal current i ld * Get the command current i hd

[0062] The second instruction current generating module converts the distorted current i lq Subtract the sinusoidal current i lq * Get the command current i hq .

[0063] In some embodiments, the command voltage generation module includes:

[0064] Active command current generation module converts the DC side reference voltage V dc * Subtract the actual DC side voltage V dc The difference is multiplied by the voltage proportional integral coefficient K PIV Get active command current;

[0065] The first command voltage generating module converts the active command current i dc With command current i hd The output current is subtracted from the sum of i fd The difference is multiplied by the current proportional integral coefficient K PII Get the command voltage u d * ;

[0066] The second instruction voltage generating module converts the instruction current i hq Subtract output current i fq The difference is multiplied by the current proportional integral coefficient K PII Get the command voltage u q * .

[0067] In some embodiments, the modulation wave adjustment module includes:

[0068] A setting module, used for setting a first adjustment amount;

[0069] Comparison module, when the i-phase command current i hi >0, the second adjustment value is equal to the first adjustment value. i hi When ≤0, the second adjustment amount is outputted equal to -1 times of the first adjustment amount;

[0070] Adjust the module to make the initial modulation wave of phase i m i * Add the first adjustment amount Δ u d of – 1 times to get the down-shifted modulation wave, i-phase initial modulation wave m i * Add the first adjustment amount to get the up-shifted modulation wave, and add the second adjustment amount to get the final modulation wave of phase i. mi12 * , the up-shifted modulation wave plus the second adjustment amount is obtained to obtain the final modulation wave of phase i m i34 * .

[0071] In some embodiments, the driving signal generating module includes:

[0072] Logic module I, according to the final modulation wave of phase i m i12 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i1 ;

[0073] Logic module II, according to the final modulation wave of phase i m i12 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i2 ;

[0074] Logic module III, according to the final modulation wave of phase i m i34 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i3 ;

[0075] Logic module IV, according to the final modulation wave of phase i m i34 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i4 .

[0076] Compared with the prior art, the advantages and positive effects of the present invention are:

[0077] The power quality improvement method and system for eliminating modulation in the vacancy area of ​​multiple zero-crossing drive provided by the present invention obtains the phase angle relationship of the grid voltage through two-phase static coordinate transformation, simplifies the complexity of multi-phase current control through two-phase rotating coordinate transformation according to the phase angle relationship, and obtains an ideal sinusoidal current by performing two-phase processing on the distorted current and filtering the noise components therein, thereby generating an accurate two-phase command current to ensure that the output current is consistent with the ideal current. The two-phase command voltage is obtained by the DC side reference voltage and actual voltage, the two-phase command current and the two-phase output current, effectively alleviating voltage distortion. The initial modulation wave is calculated by the command voltage, the carrier period and the actual DC side voltage, and the modulation wave is further fine-tuned to generate an optimized final modulation wave, and a drive signal is generated based on the relative size of the final modulation wave and the positive load wave. The power quality improvement method and system of multi-zero-crossing drive vacancy elimination modulation provided by the present invention can effectively eliminate the influence of the drive vacancy area on the grid current waveform when the inverter output current contains multiple zero-crossing points, reduce the harmonic distortion rate and improve the power quality; at the same time, when an error occurs in the given current polarity, it can weaken the dependence of the modulation strategy on the current polarity and reduce the degree of distortion of the grid current when an error occurs in the given current polarity. BRIEF DESCRIPTION OF THE DRAWINGS

[0078] Figure 1 A circuit diagram of a power grid and a distorted current generating source according to an embodiment of the present invention;

[0079] Figure 2 A circuit diagram of a Si IGBT and SiC MOSFET three-level harmonic compensation converter containing isolation inductors according to an embodiment of the present invention;

[0080] Figure 3 It is a schematic diagram of a method for improving power quality by eliminating modulation of multiple zero-crossing driving vacancy zones according to an embodiment of the present invention;

[0081] Figure 4 It is a principle diagram of the grid voltage coordinate transformation link described in an embodiment of the present invention;

[0082] Figure 5 A schematic diagram of the triangular relationship calculation according to an embodiment of the present invention;

[0083] Figure 6 This is a schematic diagram of the coordinate transformation of the distorted current according to an embodiment of the present invention;

[0084] Figure 7 This is a schematic diagram of the current filtering described in an embodiment of the present invention;

[0085] Figure 8 It is a principle diagram of output current coordinate transformation according to an embodiment of the present invention;

[0086] Fig. 9A schematic diagram of the command voltage generation according to an embodiment of the present invention;

[0087] Fig.10 It is a principle diagram of the inverse coordinate transformation of the command current according to the embodiment of the present invention;

[0088] Fig.11 It is a principle diagram of the inverse coordinate transformation of the command voltage according to an embodiment of the present invention;

[0089] Fig.12 A schematic diagram of the modulation wave generation according to an embodiment of the present invention;

[0090] Fig.13 A schematic diagram of the modulation wave adjustment according to an embodiment of the present invention;

[0091] Fig.14 A schematic diagram of the pulse signal generation according to an embodiment of the present invention;

[0092] Fig.15 This is a schematic diagram of the final modulation wave waveform of phase A after the modulation wave is adjusted according to an embodiment of the present invention;

[0093] Fig.16 It is a waveform diagram of the output voltage when the polarity of the A phase current is wrong and the ideal voltage;

[0094] Fig.17 This is a structural block diagram of a single-carrier modulation system of a SiC device isolated output inverter according to an embodiment of the present invention;

[0095] Fig.18 The current waveform and THD of the three-phase grid using the traditional modulation method;

[0096] Fig.19 The current waveform and THD of a three-phase grid using the power quality improvement method and system of the multi-zero-crossing drive vacancy elimination modulation of the present invention;

[0097] Fig. 20 The current waveform and THD of the three-phase grid using the conventional vacancy elimination method when the current polarity is wrong;

[0098] Fig.21 The three-phase grid current waveform and THD of the power quality improvement method and system using the multi-zero-crossing drive vacancy zone elimination modulation of the present invention are disclosed when the current polarity is wrong.

[0099] In the figure, 1. grid voltage coordinate transformation module, 2. trigonometric relationship calculation module, 3. distortion current coordinate transformation module, 4. current filtering module, 5. output current coordinate transformation module, 6. command voltage generation module, 7. command current inverse coordinate transformation module, 8. command voltage inverse coordinate transformation module, 9. modulation wave generation module, 10. modulation wave adjustment module, 11. drive signal generation module. DETAILED DESCRIPTION

[0100] The present invention will be described in detail below by way of exemplary embodiments in conjunction with the accompanying drawings. However, it should be understood that, without further description, elements, structures and features in one embodiment may also be beneficially combined in other embodiments.

[0101] See also Figure 1 , in the power grid and the distortion current generating source circuit, u gA , u gB , u gC is the three-phase grid voltage, i gA , i gB , i gC is the three-phase grid current, i lA , i lB , i lC The three-phase distorted current is generated by the distorted current generating source, and the three-phase grid voltage is connected to the distorted current generating source through points X, Y, and Z respectively.

[0102] Figure 2 In the circuit of the Si IGBT and SiC MOSFET three-level harmonic compensation converter with isolation inductor shown in FIG. V dc is the DC side voltage, C 1. C 2 is the DC bus capacitance, Q A1 , Q A2 , Q A3 , Q A4 A is the SiC MOSFET power switch device, D A1 , D A2 , D A3 , D A4 Four SiC Schottky diodes for phase A, L A1 , L A2 is the isolation inductance of phase A; Q B1 , Q B2 , Q B3 , Q B4 The SiC MOSFET power switch device of phase B, B1 , D B2 , D B3 , D B4 is the Schottky diode of phase B,L B1 , L B2 is the isolation inductor of phase B; Q C1 , Q C2 , Q C3 , Q C4 SiC MOSFET power switch device for phase C, D C1 , D C2 , D C3 , D C4 is the Schottky diode of phase C, L C1 , L C2 Isolation inductor for phase C; L A , L B , L C is the three-phase filter inductor, i fA , i fB , i fC The three-phase output current generated for the main topology circuit.

[0103] Continue to see Figure 2 Since the ABC three-phase bridge arms are completely symmetrical, phase A is used as an example to explain the connection method of the main circuit. C 1 and DC bus capacitor C 2 connected to point O; Q A1 One end and the DC bus capacitor C 1 is connected through point P and the other end is connected to point D A1 By X A1 Connect the dots, D A1 The other end and D A4 Connect through point O, D A4 The other end and Q A4 By X A4 Connect the dots, Q A4 The other end is connected to the DC bus capacitor C2 through point N; Q A2 One end and X A1 point, and the other end is connected to D A3 By X A2 Connect the dots, D A3 The other end and X A4 Connect the dots; D A2 One end and X A1 point connection, and the other end with Q A3 By X A3 Connect the dots, Q A3 The other end and X A4Connect the dots; X A2 Click through L A1 With X A4 Connect, X A3 Click through L A2 With X A4 Connect, X A4 Click through L A Connect to output point X. The connection method of the B-phase and C-phase main circuits is the same as that of the A-phase, so it will not be repeated here.

[0104] For the above-mentioned converter, the present invention provides a method and system for improving power quality by eliminating modulation of multiple zero-crossing drive vacancy areas, performing two-phase static coordinate transformation on the grid voltage, calculating the phase angle relationship of the grid voltage according to the transformed two-phase grid, performing two-phase rotating coordinate transformation on the three-phase distorted current and the three-phase output current according to the phase angle relationship, simplifying the complexity of multi-phase current control, and filtering the noise components therein by two-phase processing of the distorted current to obtain an ideal sinusoidal current, thereby generating an accurate two-phase command current to ensure that the output current is consistent with the ideal current. The two-phase command voltage is obtained by the DC side reference voltage and actual voltage, the two-phase command current and the two-phase output current, effectively alleviating voltage distortion. The initial modulation wave is calculated by the command voltage, the carrier period and the actual DC side voltage, and the modulation wave is further fine-tuned to generate an optimized final modulation wave, and a drive signal is generated based on the relative size of the final modulation wave and the positive load wave. The present invention effectively eliminates the drive vacancy area caused by the carrier zero point position in the modulation, reduces the harmonic distortion rate, and thus significantly improves the power quality of the system.

[0105] The power quality improvement method and system of the multi-zero-crossing drive vacancy elimination modulation of the present invention are described in detail below in conjunction with the accompanying drawings and embodiments.

[0106] See also Figure 3 The first aspect of the present invention provides a method for improving power quality by eliminating modulation of multiple zero-crossing drive vacancy areas, and the specific steps are as follows:

[0107] S1. Grid voltage coordinate transformation step: transform the three-phase grid voltage into a two-phase grid voltage in a two-phase stationary coordinate system.

[0108] Specifically, see Figure 4 , the method of transforming the three-phase grid voltage into the two-phase grid voltage in the two-phase stationary coordinate system is:

[0109] A phase grid voltage u gA 2 times minus the B phase grid voltage u gBand C phase grid voltage u gC Then, multiply by 0.333 to get the grid voltage in the two-phase stationary coordinate system. u gα ;

[0110] Phase B grid voltage u gB Subtract C phase grid voltage u gC Then, multiply by 0.577 to get the grid voltage in the two-phase stationary coordinate system. u gβ .

[0111] S2. Trigonometric relationship calculation steps: Calculate the phase angle relationship of the grid voltage based on the two-phase grid voltage.

[0112] Specifically, see Figure 5 , the method for calculating the phase angle relationship of the grid voltage based on the two-phase grid voltage is: grid voltage u gα Divide by the grid voltage u gα The square of the grid voltage u gβ The square root of the sum of the squares of the grid is the cosine function cosθ of the grid phase angle; the grid voltage u gβ Divide by the grid voltage u gα The square of the grid voltage u gβ The square root of the sum of the squares of the grid is the sinusoidal function sinθ of the grid phase angle.

[0113] S3, distortion current coordinate transformation step: transform the three-phase distorted current into a two-phase distorted current in a two-phase rotating coordinate system according to the phase angle relationship.

[0114] Specifically, see Figure 6 , the method of converting the three-phase distorted current into the two-phase distorted current in the two-phase rotating coordinate system according to the phase angle relationship is:

[0115] The difference between 0.707 times the negative sine function sinθ and 0.408 times the cosine function cosθ multiplied by the C phase distortion current i lC The product of the sine function sinθ (0.707 times) and the cosine function cosθ (0.408 times) multiplied by the B-phase distortion current i lB The product of the cosine function cosθ times 0.816 times the A phase distortion current i lA The product of the two phases is the distortion current in the rotating coordinate system.i ld ;

[0116] The sum of 0.408 times the sine function sinθ and 0.707 times the cosine function cosθ multiplied by the B-phase distortion current i lB The product of the sine function sinθ minus 0.816 times multiplied by the A phase distortion current i lA The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the C phase distortion current i lC The product of the two phases is the distortion current in the rotating coordinate system. i lq .

[0117] S4, current filtering step: eliminate the high-frequency current ripple in the two-phase distorted current to obtain the two-phase sinusoidal current, and obtain the two-phase command current according to the two-phase distorted current and the two-phase sinusoidal current.

[0118] Specifically, see Figure 7 , the method for obtaining the two-phase sinusoidal current is: eliminating the high-frequency current ripple in the two-phase distorted current by a filter to obtain the two-phase sinusoidal current, and the transfer function of the filter is expressed as:

[0119]

[0120] In the formula, L ( ω ) is the transfer function of the filter, s is a complex variable in the frequency domain, representing the frequency.

[0121] The method of obtaining the two-phase command current based on the two-phase distorted current and the two-phase sinusoidal current is: i ld Subtract the sinusoidal current i ld * Get command current i hd , distortion current i lq Subtract the sinusoidal current i lq * Get command current i hq .

[0122] S5, output current coordinate transformation step: transform the three-phase output current into a two-phase output current in a two-phase rotating coordinate system according to the phase angle relationship.

[0123] Specifically, see Figure 8, the method of converting the three-phase output current into the two-phase output current in the two-phase rotating coordinate system according to the phase angle relationship is:

[0124] The difference between 0.707 times the negative sine function sinθ and 0.408 times the cosine function cosθ multiplied by the C phase output current i fC The product of the sine function sinθ times 0.707 and the cosine function cosθ times 0.408 times multiplied by the B phase output current i fB The product of the cosine function cosθ times 0.816 times the output current of phase A i fA The product of the distorted current and the output current in the two-phase rotating coordinate system is obtained. i fd ;

[0125] The sum of 0.408 times the sine function sinθ and 0.707 times the cosine function cosθ multiplied by the B phase output current i fB The product of the sine function sinθ minus 0.816 times multiplied by the A phase output current i fA The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the C phase output current i fC The product of the distorted current and the output current in the two-phase rotating coordinate system is obtained. i fq .

[0126] S6, command voltage generation steps: see Fig. 9 , DC side reference voltage V dc * Subtract the actual DC side voltage V dc The difference is multiplied by the voltage proportional integral coefficient K PIV An active command current is obtained, and a two-phase command voltage is obtained according to the active command current, the two-phase command current and the two-phase output current.

[0127] Specifically, see Fig. 9 , the method of obtaining the two-phase command voltage according to the active command current, the two-phase command current and the two-phase output current is:

[0128] Active command current i dc With command current i hd The output current is subtracted from the sum of i fdThe difference is multiplied by the current proportional integral coefficient K PII Get the command voltage u d * ;

[0129] Command current i hq Subtract output current i fq The difference is multiplied by the current proportional integral coefficient K PII Get the command voltage u q * ;

[0130] Voltage proportional integral coefficient K PIV =0.3+100 / s , current proportional integral coefficient K PII =5+1500 / s , s is a complex variable in the frequency domain, representing the frequency.

[0131] S7, command current reverse coordinate transformation step: convert the two-phase command current into a three-phase command current in a three-phase stationary coordinate system according to the phase angle relationship.

[0132] Specifically, see Fig.10 , the method of converting the two-phase command current into the three-phase command current in the three-phase stationary coordinate system according to the phase angle relationship is:

[0133] Cosine function cosθ multiplied by command current i hd The product of plus the sine function sinθ multiplied by the command current i hq The product of the two-phase command currents is the A-phase command current in the three-phase stationary coordinate system. i hA ;

[0134] The difference between 0.707 times the sine function sinθ and 0.408 times the cosine function cosθ multiplied by the command current i hd The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the command current i hq The product of the two-phase command currents is obtained as the B-phase command current in the three-phase stationary coordinate system. i hB ;

[0135] The difference between 0.707 times the negative sine function sinθ and 0.408 times the cosine function cosθ multiplied by the command current command current i hd The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the command current i hq The product of the two-phase command currents is the C-phase command current in the three-phase stationary coordinate system. i hC .

[0136] S8, command voltage reverse coordinate transformation step: convert the two-phase command voltage into a three-phase command voltage in a three-phase stationary coordinate system according to the phase angle relationship.

[0137] Specifically, see Fig.11 , the method of converting the two-phase command voltage into the three-phase command voltage in the three-phase stationary coordinate system according to the phase angle relationship is:

[0138] Cosine function cosθ multiplied by command voltage u d * The product of plus the sine function sinθ multiplied by the command voltage u q * The product of the two-phase command voltages is the A-phase command voltage in the three-phase stationary coordinate system. u A * ;

[0139] The difference between 0.707 times the sine function sinθ and 0.408 times the cosine function cosθ multiplied by the command voltage u d * The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the command voltage u q * The product of the two-phase command voltages is obtained as the B-phase command voltage in the three-phase stationary coordinate system. u B * ;

[0140] The difference between 0.707 times the negative sine function sinθ and 0.408 times the cosine function cosθ multiplied by the command voltage u d * The product of the sine function sinθ and the cosine function cosθ multiplied by the command voltage u q *The product of the two-phase command voltages is obtained as the C-phase command voltage in the three-phase stationary coordinate system. u C * .

[0141] S9, modulation wave generation step: i-phase command voltage u i * Multiply by the carrier period T s Divide by the actual DC voltage V dc , minus the i-phase command voltage u i * Multiply by the sign function sign and then multiply by the carrier period T s Divide by 4 to get the first modulation wave u i1 * , the sum of the maximum value of the first modulation wave plus the minimum value of the first modulation wave multiplied by 0.5 to obtain the second modulation wave u i2 * , i-phase command voltage u i * Multiply by the carrier period T s Divide by the actual DC voltage V dc Subtract the second modulation wave u i2 * Get the initial modulation wave of phase i m i * , i=A,B,C.

[0142] Specifically, see Fig.12 , to generate the initial modulation wave of phase A m A * For example, the A phase command voltage u A * Multiply by the carrier period T s Divide by the actual DC voltage V dc , minus the A phase command voltage u A * Multiply by the sign function sign and then multiply by the carrier period T s Divide by 4 to get the first modulation wave u A1 *, the maximum value of the first modulation wave plus the first modulation wave u A1 * The sum of the minimum values ​​of the two modulated waves is multiplied by 0.5 to obtain the second modulated wave. u A2 * , A phase command voltage u A * Multiply by the carrier period T s Divide by the actual DC voltage V dc Subtract the second modulation wave u A2 * Get the initial modulation wave of phase A m A * .

[0143] S10, modulation wave adjustment step: setting the first adjustment amount Δ u d Equal to the dead time, when the i-phase command current i hi >0, the second adjustment amount is equal to the first adjustment amount. When the i-phase command current i hi ≤0, the second adjustment amount is equal to -1 times of the first adjustment amount; the initial modulation wave of phase i m i * Add the first adjustment amount Δ u d of – 1 times to get the down-shifted modulation wave m i12 , initial modulation wave of phase i m i * Add the first adjustment amount Δ u d Get the up-shifted modulation wave m i34 , down-shift the modulation wave m i12 Add the second adjustment amount Δ u h Get the final modulation wave of phase i m i12 * , up-shift the modulation wave m i34 Add the second adjustment amount Δ u h Get the final modulation wave of phase i m i34 * .

[0144] Specifically, see Fig.13 , to adjust the initial modulation wave of phase A m A * As an example, set the first adjustment amount Δ u d Equal to the dead time, when the A phase command current i hA >0, the second adjustment amount is equal to the first adjustment amount. When the A phase command current i hA ≤0, the second adjustment amount is equal to -1 times of the first adjustment amount; the initial modulation wave of phase A m A * Add the first adjustment amount Δ u d of – 1 times to get the down-shifted modulation wave m A12 , initial modulation wave of phase A m A * Add the first adjustment amount Δ u d Get the up-shifted modulation wave m A34 , down-shift the modulation wave m A12 Add the second adjustment amount Δ u h Get the final modulation wave of phase A m A12 * , up-shift the modulation wave m A34 Add the second adjustment amount Δ u h Get the final modulation wave of phase A m A34 * .

[0145] S11, driving signal generation step: according to the final modulation wave of phase i m i12 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i1 , according to the final modulation wave of phase i m i12 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i2, according to the final modulation wave of phase i m i34 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i3 , according to the final modulation wave of phase i m i34 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i4 .

[0146] Specifically, see Fig.14 , according to the final modulation wave of phase i m i12 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i1 The method is: m i12 * > u c1 * hour, S i1 is high level, m i12 * ≤ u c1 * hour, S i1 is low level;

[0147] According to the final modulation wave of phase i m i12 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i2 The method is: m i12 * > u c2 * hour, S i2 is high level, m i12 * ≤ u c2* hour, S i2 is low level;

[0148] According to the final modulation wave of phase i m i34 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i3 The method is: m i34 * > u c1 * hour, S i3 is high level, m i34 * ≤ u c1 * hour, S i3 is low level;

[0149] According to the final modulation wave of phase i m i34 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i4 The method is: m i34 * > u c2 * hour, S i4 When high level m i34 * ≤ u c2 * hour, S i4 is low level.

[0150] Continue to see Fig.14 , generating A phase drive signal S A1 , S A2 , S A3 , S A4 The generation process is as follows:

[0151] S A1 The final modulation wave of phase A mA12 * With positive carrier u c1 * The value obtained by performing a logical operation with “>”, m A12 * > u c1 * hour, S A1 is high level, m A12 * ≤ u c1 * hour, S A1 is low level.

[0152] S A2 The final modulation wave of phase A m A12 * With load wave u c2 * The value obtained by performing a logical operation with “>”, m A12 * > u c2 * hour, S A2 is high level, m A12 * ≤ u c2 * hour, S A2 is low level;

[0153] S A3 The final modulation wave of phase A m A34 * With positive carrier u c1 * The value obtained by performing a logical operation with “>”, m A34 * > u c1 * hour, S A3 is high level, m A34 * ≤u c1 * hour, S A3 is low level;

[0154] S A4 The final modulation wave of phase A m A34 * With load wave u c2 * The value obtained by performing a logical operation with “>”, m A34 * > u c2 * hour, S A4 When high level m A34 * ≤ u c2 * hour, S A4 is low level.

[0155] In order to more intuitively demonstrate the basic principle of the power quality improvement method of multiple zero-crossing drive vacancy elimination modulation, Fig.15 The following figure shows the final modulation wave waveform generated by the modulation wave adjustment step. m A12 , Up-shift modulation wave m A34 and the adjusted modulation wave m A12 * , m A34 * exist i hA The changing waveform when the zero point changes. Fig.16 The waveform diagram of the output voltage when the polarity of phase A current is wrong is shown compared with the ideal voltage, and the difference between the two is given.

[0156] The above specifically introduces the A phase drive signal S A1 , S A2 , S A3 , S A4 The generation process of the driving signal of phase B and phase C is the same, which will not be repeated here.

[0157] See also Fig.17The second aspect of the present invention provides a single-carrier modulation system for an isolated output inverter of a SiC device, which is used to implement the method for improving the power quality of the multi-zero-crossing drive vacancy elimination modulation described in the first aspect of the present invention, including:

[0158] A grid voltage coordinate transformation module 1 transforms the three-phase grid voltage into a two-phase grid voltage in a two-phase stationary coordinate system;

[0159] A trigonometric relationship calculation module 2 calculates the phase angle relationship of the grid voltage according to the two-phase grid voltage;

[0160] The distortion current coordinate transformation module 3 transforms the three-phase distortion current into a two-phase distortion current in a two-phase rotating coordinate system according to the phase angle relationship;

[0161] The current filter module 4 eliminates the high-frequency current ripple in the two-phase distorted current to obtain the two-phase sinusoidal current, and obtains the two-phase command current according to the two-phase distorted current and the two-phase sinusoidal current;

[0162] The output current coordinate transformation module 5 converts the three-phase output current into a two-phase output current in a two-phase rotating coordinate system according to the phase angle relationship;

[0163] The command voltage generating module 6 generates a voltage according to the DC side reference voltage, the DC side actual voltage and the voltage proportional integral coefficient. K PIV An active command current is obtained, and a two-phase command voltage is obtained according to the active command current, the two-phase command current and the two-phase output current;

[0164] The command current inverse coordinate transformation module 7 converts the two-phase command current into a three-phase command current in a three-phase stationary coordinate system according to the phase angle relationship;

[0165] The command voltage inverse coordinate transformation module 8 converts the two-phase command voltage into a three-phase command voltage in a three-phase stationary coordinate system according to the phase angle relationship;

[0166] The modulation wave generating module 9 generates an initial modulation wave of the i-phase according to the i-phase command voltage, the carrier cycle, and the actual voltage on the DC side;

[0167] A modulation wave adjustment module 10, which adjusts the initial modulation wave of phase i according to the first adjustment amount and the second adjustment amount to obtain the final modulation wave of phase i;

[0168] The driving signal generating module 11 generates an i-phase driving signal according to the magnitudes of the i-phase final modulation wave, the positive carrier wave and the load wave.

[0169] In some embodiments, the current filtering module includes:

[0170] The filter is used to eliminate the high-frequency current ripple in the two-phase distorted current to obtain a two-phase sinusoidal current. The transfer function of the filter is expressed as:

[0171]

[0172] In the formula, L ( ω ) is the transfer function of the filter, s is a complex variable in the frequency domain, representing the frequency;

[0173] The first instruction current generating module converts the distorted current i ld Subtract the sinusoidal current i ld * Get the command current i hd

[0174] The second instruction current generating module converts the distorted current i lq Subtract the sinusoidal current i lq * Get the command current i hq .

[0175] In some embodiments, the command voltage generation module includes:

[0176] Active command current generation module converts the DC side reference voltage V dc * Subtract the actual DC side voltage V dc The difference is multiplied by the voltage proportional integral coefficient K PIV Get active command current;

[0177] The first command voltage generating module converts the active command current i dc With command current i hd The output current is subtracted from the sum of i fd The difference is multiplied by the current proportional integral coefficient K PII Get the command voltage u d * ;

[0178] The second instruction voltage generating module converts the instruction current i hq Subtract output current i fqThe difference is multiplied by the current proportional integral coefficient K PII Get the command voltage u q * .

[0179] In some embodiments, the modulation wave adjustment module includes:

[0180] A setting module, used for setting a first adjustment amount;

[0181] Comparison module, when the i-phase command current i hi >0, the second adjustment value is equal to the first adjustment value. i hi When ≤0, the second adjustment amount is outputted equal to -1 times of the first adjustment amount;

[0182] Adjust the module to make the initial modulation wave of phase i m i * Add the first adjustment amount Δ u d of – 1 times to get the down-shifted modulation wave, i-phase initial modulation wave m i * Add the first adjustment amount to get the up-shifted modulation wave, and add the second adjustment amount to get the final modulation wave of phase i. m i12 * , the up-shifted modulation wave plus the second adjustment amount is obtained to obtain the final modulation wave of phase i m i34 * .

[0183] Specifically, in some embodiments, the comparison module selects a selection switch Switch for comparison and judgment. The Switch is divided into an input side and an output side. The input side has three signal inputs, terminals "+" and "-" are signal input terminals, terminal I is a comparison signal input terminal, and terminal O is a signal output terminal. When the signal of the comparison signal terminal I is greater than 0, the O terminal outputs the signal received by the "+" terminal; when the comparison signal received by the I terminal is less than or equal to 0, the O terminal outputs the signal received by the "-" terminal.

[0184] In some embodiments, the driving signal generating module includes:

[0185] Logic module I, according to the final modulation wave of phase i m i12 * , positive carrier u c1 *The magnitude of the i-phase drive signal is generated S i1 ;

[0186] Logic module II, according to the final modulation wave of phase i m i12 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i2 ;

[0187] Logic module III, according to the final modulation wave of phase i m i34 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i3 ;

[0188] Logic module IV, according to the final modulation wave of phase i m i34 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i4 .

[0189] In order to verify the effectiveness of the above-mentioned method and system for improving power quality by eliminating modulation of multiple zero-crossing drive vacancy zones, simulation verification is carried out in MATLAB / Simulink. The DC bus capacitor is C 1= C 2 = 240μF, the effective value of the grid line voltage is 304V, the switching frequency f s =10kHz, filter inductor L A = L B = L C =1mH. In the simulation model, the conventional vacancy elimination method and the power quality improvement method of multi-zero-crossing driving vacancy elimination modulation and the system elimination modulation driving vacancy elimination method proposed in the present invention are respectively used for the Si IGBT and SiC MOSFET three-level harmonic compensation converters with isolation inductors. Fig.18 The output current using the traditional modulation method is given, and the THD of the three-phase currents A, B, and C are 5.63%, 5.77%, and 5.65% respectively; Fig.19In the present invention, after adopting the power quality improvement method and system for eliminating modulation by driving multiple zero-crossing vacancy areas, the THD of the three-phase currents A, B, and C are 3.74%, 3.72%, and 3.77% respectively, and the sinusoidality of the three-phase current waveform is significantly improved, which effectively improves the current waveform of the power grid when distorted current pollution exists. Fig. 20 The output current of the conventional vacancy elimination method when the current polarity is wrong is given. The THD of the three-phase currents A, B, and C are 14.73%, 14.67%, and 14.79% respectively, making the current distortion more obvious; Fig.21 In the present invention, after adopting the method and system for improving power quality by eliminating modulation with multiple zero-crossing driving vacancy areas, the THD of the three-phase currents A, B, and C are 3.84%, 3.80%, and 3.82% respectively when the current polarity is wrong, which is only slightly increased compared with when the current polarity is correct. The dependence of the modulation strategy on the current polarity is successfully reduced, and the influence of the driving vacancy area on the grid current waveform can be effectively eliminated, the harmonic distortion rate is reduced, and the power quality is improved; at the same time, when an error occurs in the given current polarity, the dependence of the modulation strategy on the current polarity is weakened, and the degree of distortion of the grid current when the given current polarity is wrong is reduced.

[0190] The above embodiments are used to explain the present invention rather than to limit the present invention. Any modification and change made to the present invention within the spirit of the present invention and the protection scope of the claims shall fall within the protection scope of the present invention.

Claims

1. A method for improving power quality by eliminating modulation of multiple zero-crossing drive vacancy areas, characterized in that: The following steps are involved: Grid voltage coordinate transformation steps: transform the three-phase grid voltage into a two-phase grid voltage in a two-phase stationary coordinate system; Trigonometric relationship calculation steps: Calculate the phase angle relationship of the grid voltage based on the two-phase grid voltage; Distortion current coordinate transformation steps: transform the three-phase distortion current into a two-phase distortion current in a two-phase rotating coordinate system according to the phase angle relationship; Current filtering steps: eliminate the high-frequency current ripple in the two-phase distorted current to obtain the two-phase sinusoidal current, and obtain the two-phase command current according to the two-phase distorted current and the two-phase sinusoidal current; Output current coordinate transformation steps: convert the three-phase output current into a two-phase output current in a two-phase rotating coordinate system according to the phase angle relationship; Command voltage generation steps: DC side reference voltage V dc * Subtract the actual DC side voltage V dc The difference is multiplied by the voltage proportional integral coefficient K PIV An active command current is obtained, and a two-phase command voltage is obtained according to the active command current, the two-phase command current and the two-phase output current; The command current reverse coordinate transformation steps are: converting the two-phase command current into the three-phase command current in the three-phase stationary coordinate system according to the phase angle relationship; The command voltage reverse coordinate transformation step: convert the two-phase command voltage into a three-phase command voltage in a three-phase stationary coordinate system according to the phase angle relationship; Modulation wave generation steps: i-phase command voltage u i * Multiply by the carrier period T s Divide by the actual DC voltage V dc , minus the i-phase command voltage u i * Multiply by the sign function sign and then multiply by the carrier period T s Divide by 4 to get the first modulation wave u i1 * , the sum of the maximum value of the first modulation wave plus the minimum value of the first modulation wave multiplied by 0.5 to obtain the second modulation wave u i2 * , i-phase command voltage u i * Multiply by the carrier period T s Divide by the actual DC voltage V dc Subtract the second modulation wave u i2 * Get the initial modulation wave of phase i m i * , i=A,B,C; Modulation wave adjustment steps: Set the first adjustment amount Δ u d Equal to the dead time, when the i-phase command current i hi >0, the second adjustment amount is equal to the first adjustment amount. When the i-phase command current i hi ≤0, the second adjustment amount is equal to -1 times the first adjustment amount; Phase i initial modulation wave m i * Add the first adjustment amount Δ u d of – 1 times to get the down-shifted modulation wave m i12 , initial modulation wave of phase i m i * Add the first adjustment amount Δ u d Get the up-shifted modulation wave m i34 , down-shift the modulation wave m i12 Add the second adjustment amount Δ u h Get the final modulation wave of phase i m i12 * , up-shift the modulation wave m i34 Add the second adjustment amount Δ u h Get the final modulation wave of phase i m i34 * ; Driving signal generation steps: According to the final modulation wave of phase i m i12 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i1 , according to the final modulation wave of phase i m i12 * , Load Wave u c2 * The magnitude of the i-phase drive signal is generated S i2 , according to the final modulation wave of phase i m i34 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i3 , according to the final modulation wave of phase i m i34 * , Load Wave u c2 * The magnitude of the i-phase drive signal is generated S i4 .

2. The method for improving power quality by eliminating modulation of multiple zero-crossing drive vacancy zones as claimed in claim 1, characterized in that: In the grid voltage coordinate transformation step, the method for transforming the three-phase grid voltage into the two-phase grid voltage in the two-phase stationary coordinate system is: A phase grid voltage u gA 2 times minus the B phase grid voltage u gB and C phase grid voltage u gC Then, multiply by 0.333 to get the grid voltage in the two-phase stationary coordinate system. u gα ; Phase B grid voltage u gB Subtract C phase grid voltage u gC Then, multiply by 0.577 to get the grid voltage in the two-phase stationary coordinate system. u gβ .

3. The method for improving power quality by eliminating modulation of multiple zero-crossing drive vacancy zones as claimed in claim 1, characterized in that: In the trigonometric relationship calculation, the method for calculating the phase angle relationship of the grid voltage based on the two-phase grid voltage is: grid voltage u gα Divide by the grid voltage u gα The square of the grid voltage u gβ The square root of the sum of the squares of the grid is the cosine function cosθ of the grid phase angle; the grid voltage u gβ Divide by the grid voltage u gα The square of the grid voltage u gβ The square root of the sum of the squares of the grid is the sinusoidal function sinθ of the grid phase angle.

4. The method for improving power quality by eliminating modulation of multiple zero-crossing drive vacancy zones as claimed in claim 3, characterized in that: In the distortion current coordinate transformation step, the method of converting the three-phase distortion current into the two-phase distortion current in the two-phase rotating coordinate system according to the phase angle relationship is: The difference between 0.707 times the negative sine function sinθ and 0.408 times the cosine function cosθ multiplied by the C phase distortion current i lC The product of the sine function sinθ (0.707 times) and the cosine function cosθ (0.408 times) multiplied by the B-phase distortion current i lB The product of the cosine function cosθ times 0.816 times the A phase distortion current i lA The product of the two phases is the distortion current in the rotating coordinate system. i ld ; The sum of 0.408 times the sine function sinθ and 0.707 times the cosine function cosθ multiplied by the B-phase distortion current i lB The product of the sine function sinθ minus 0.816 times multiplied by the A phase distortion current i lA The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the C phase distortion current i lC The product of the two phases is the distortion current in the rotating coordinate system. i lq ; In the output current coordinate transformation step, the method of converting the three-phase output current into the two-phase output current in the two-phase rotating coordinate system according to the phase angle relationship is: The difference between 0.707 times the negative sine function sinθ and 0.408 times the cosine function cosθ multiplied by the C phase output current i fC The product of the sine function sinθ times 0.707 and the cosine function cosθ times 0.408 times multiplied by the B phase output current i fB The product of the cosine function cosθ times 0.816 times the output current of phase A i fA The product of the distorted current and the output current in the two-phase rotating coordinate system is obtained. i fd ; The sum of 0.408 times the sine function sinθ and 0.707 times the cosine function cosθ multiplied by the B phase output current i fB The product of the sine function sinθ minus 0.816 times multiplied by the A phase output current i fA The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the C phase output current i fC The product of the distorted current and the output current in the two-phase rotating coordinate system is obtained. i fq .

5. The method for improving power quality by eliminating modulation of multiple zero-crossing drive vacancy zones as claimed in claim 3, characterized in that: In the command current inverse coordinate transformation step, the method of converting the two-phase command current into the three-phase command current in the three-phase stationary coordinate system according to the phase angle relationship is: Cosine function cosθ multiplied by command current i hd The product of plus the sine function sinθ multiplied by the command current i hq The product of the two-phase command currents is the A-phase command current in the three-phase stationary coordinate system. i hA ; The difference between 0.707 times the sine function sinθ and 0.408 times the cosine function cosθ multiplied by the command current i hd The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the command current i hq The product of the two-phase command currents is obtained as the B-phase command current in the three-phase stationary coordinate system. i hB ; The difference between 0.707 times the negative sine function sinθ and 0.408 times the cosine function cosθ multiplied by the command current command current i hd The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the command current i hq The product of the two-phase command currents is the C-phase command current in the three-phase stationary coordinate system. i hC ; In the command voltage inverse coordinate transformation step, the method of converting the two-phase command voltage into the three-phase command voltage in the three-phase stationary coordinate system according to the phase angle relationship is: Cosine function cosθ multiplied by command voltage u d * The product of plus the sine function sinθ multiplied by the command voltage u q * The product of the two-phase command voltages is the A-phase command voltage in the three-phase stationary coordinate system. u A * ; The difference between 0.707 times the sine function sinθ and 0.408 times the cosine function cosθ multiplied by the command voltage u d * The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the command voltage u q * The product of the two-phase command voltages is obtained as the B-phase command voltage in the three-phase stationary coordinate system. u B * ; The difference between 0.707 times the negative sine function sinθ and 0.408 times the cosine function cosθ multiplied by the command voltage u d * The product of the sine function sinθ (0.408 times) and the cosine function cosθ (0.707 times) multiplied by the command voltage u q * The product of the two-phase command voltages is obtained as the C-phase command voltage in the three-phase stationary coordinate system. u C * .

6. The method for improving power quality by eliminating modulation of multiple zero-crossing drive vacancy zones as claimed in claim 1, characterized in that: In the current filtering step, the method of obtaining the two-phase command current according to the two-phase distorted current and the two-phase sinusoidal current is: i ld Subtract the sinusoidal current i ld * Get command current i hd , distortion current i lq Subtract the sinusoidal current i lq * Get command current i hq .

7. The method for improving power quality by eliminating modulation of multiple zero-crossing driving vacancy zones as claimed in claim 6, characterized in that: In the command voltage generation step, the method of obtaining the two-phase command voltage according to the active command current, the two-phase command current and the two-phase output current is: Active command current i dc With command current i hd The output current is subtracted from the sum of i fd The difference is multiplied by the current proportional integral coefficient K PII Get the command voltage u d * ; Command current i hq Subtract output current i fq The difference is multiplied by the current proportional integral coefficient K PII Get the command voltage u q * ; Voltage proportional integral coefficient K PIV =0.3+100 / s , current proportional integral coefficient K PII =5+1500 / s , s is a complex variable in the frequency domain, representing the frequency.

8. The method for improving power quality by eliminating modulation of multiple zero-crossing drive vacancy zones as claimed in claim 1, characterized in that: In the driving signal generating step, according to the final modulation wave of phase i m i12 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i1 The method is: m i12 * > u c1 * hour, S i1 is high level, m i12 * ≤ u c1 * hour, S i1 is low level; According to the final modulation wave of phase i m i12 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i2 The method is: m i12 * > u c2 * hour, S i2 is high level, m i12 * ≤ u c2 * hour, S i2 is low level; According to the final modulation wave of phase i m i34 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i3 The method is: m i34 * > u c1 * hour, S i3 is high level, m i34 * ≤ u c1 * hour, S i3 is low level; According to the final modulation wave of phase i m i34 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i4 The method is: m i34 * > u c2 * hour, S i4 When high level m i34 * ≤ u c2 * hour, S i4 is low level.

9. A system for improving power quality by eliminating modulation of multiple zero-crossing driving vacancy zones, used to implement the method for improving power quality by eliminating modulation of multiple zero-crossing driving vacancy zones as claimed in any one of claims 1 to 8, characterized in that: include: A grid voltage coordinate transformation module transforms the three-phase grid voltage into a two-phase grid voltage in a two-phase stationary coordinate system; A trigonometric relationship calculation module calculates the phase angle relationship of the grid voltage based on the two-phase grid voltage; The distortion current coordinate transformation module converts the three-phase distortion current into a two-phase distortion current in a two-phase rotating coordinate system according to the phase angle relationship; The current filter module eliminates the high-frequency current ripple in the two-phase distorted current to obtain the two-phase sinusoidal current, and obtains the two-phase command current according to the two-phase distorted current and the two-phase sinusoidal current; An output current coordinate transformation module converts the three-phase output current into a two-phase output current in a two-phase rotating coordinate system according to the phase angle relationship; The command voltage generation module generates the voltage according to the DC side reference voltage, the actual DC side voltage and the voltage proportional integral coefficient. K PIV An active command current is obtained, and a two-phase command voltage is obtained according to the active command current, the two-phase command current and the two-phase output current; The command current inverse coordinate transformation module converts the two-phase command current into the three-phase command current in the three-phase stationary coordinate system according to the phase angle relationship; The command voltage inverse coordinate transformation module converts the two-phase command voltage into a three-phase command voltage in a three-phase stationary coordinate system according to the phase angle relationship; The modulation wave generation module generates an initial modulation wave of the i-phase according to the i-phase command voltage, the carrier cycle, and the actual voltage on the DC side; A modulation wave adjustment module, adjusting the initial modulation wave of phase i according to the first adjustment amount and the second adjustment amount to obtain a final modulation wave of phase i; The driving signal generating module generates the i-phase driving signal according to the magnitude of the i-phase final modulation wave, the positive carrier wave and the load wave.

10. The power quality improvement system of multiple zero-crossing drive vacancy elimination modulation as claimed in claim 9, characterized in that: The current filtering module comprises: The filter is used to eliminate the high-frequency current ripple in the two-phase distorted current to obtain a two-phase sinusoidal current. The transfer function of the filter is expressed as: In the formula, L ( ω ) is the transfer function of the filter, s is a complex variable in the frequency domain, representing the frequency; The first instruction current generating module converts the distorted current i ld Subtract the sinusoidal current i ld * Get command current i hd ; The second instruction current generating module converts the distorted current i lq Subtract the sinusoidal current i lq * Get command current i hq ; The command voltage generating module comprises: Active command current generation module converts the DC side reference voltage V dc * Subtract the actual DC side voltage V dc The difference is multiplied by the voltage proportional integral coefficient K PIV Get active command current; The first command voltage generating module converts the active command current i dc With command current i hd The output current is subtracted from the sum of i fd The difference is multiplied by the current proportional integral coefficient K PII Get the command voltage u d * ; The second instruction voltage generating module converts the instruction current i hq Subtract output current i fq The difference is multiplied by the current proportional integral coefficient K PII Get the command voltage u q * ; The modulation wave adjustment module comprises: A setting module, used for setting a first adjustment amount; Comparison module, when the i-phase command current i hi >0, the second adjustment value is equal to the first adjustment value. i hi When ≤0, the second adjustment amount is outputted equal to -1 times of the first adjustment amount; Adjust the module to make the initial modulation wave of phase i m i * Add the first adjustment amount Δ u d of – 1 times to get the down-shifted modulation wave, i-phase initial modulation wave m i * Add the first adjustment amount to get the up-shifted modulation wave, and add the second adjustment amount to get the final modulation wave of phase i. m i12 * , the up-shifted modulation wave plus the second adjustment amount is obtained to obtain the final modulation wave of phase i m i34 * ; The driving signal generating module comprises: Logic module I, according to the final modulation wave of phase i m i12 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i1 ; Logic module II, according to the final modulation wave of phase i m i12 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i2 ; Logic module III, according to the final modulation wave of phase i m i34 * , positive carrier u c1 * The magnitude of the i-phase drive signal is generated S i3 ; Logic module IV, according to the final modulation wave of phase i m i34 * , load wave u c2 * The magnitude of the i-phase drive signal is generated S i4 .

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