Output matching network applied to double-frequency power amplifier

Abstract

The invention discloses an output matching network applied to a dual-frequency power amplifier, comprising: a dual transmission network, the input end of which is used as the input end of the output matching network; the dual-transmission network comprises a microstrip line TL01 and a microstrip line TL02 which are connected in parallel; one end of the microstrip line TL1 is connected with the output end of the dual-transmission network, and the other end is grounded or open-circuited; one end of the microstrip line TL2 is connected with the output end of the dual-transmission network, and the other end of the microstrip line TL2 serves as the output end of the output matching network; the open-circuit microstrip line TL31 is connected with the other end of the microstrip line TL2; according to the dual-frequency power amplifier, dual-frequency matching and harmonic control are achieved, meanwhile, the output matching circuit design of the power amplifier is remarkably simplified, the design efficiency and performance are improved, and a novel and efficient method is provided for the design of the dual-frequency power amplifier.

Description

Technical Field

[0001] The present application relates to the field of amplifier circuits, and in particular to an output matching network applied to a dual-band power amplifier. Background Art

[0002] At present, with the continuous evolution of communication formats and mobile communication standards, microwave radio frequency systems need to adapt to the trend of multi-band, multi-format and coexistence. This means that a single set of circuits needs to be matched on different frequency bands to meet the needs of multi-band wireless communication systems, thereby reducing system volume and cost and improving overall performance. However, traditional power amplifiers (PAs) can no longer meet the requirements of large span and high efficiency in high frequency bands. Therefore, it is necessary to design a wireless communication system that can cover multiple frequency bands and has a certain bandwidth, which requires the support of multi-band power amplifiers. Multi-band power amplifiers can not only support multiple working modes and cover multiple communication frequency bands, but also be compatible with multiple communication standards, effectively alleviating the problem of tight spectrum resources for future wireless communication. There are many ways to achieve multi-band. Multiple power amplifier tubes can be used to realize single-band signal synthesis respectively, or reconfigurable components can be used to realize multi-band power amplifiers, but these methods will increase the size of the power amplifier and do not meet the requirements of circuit miniaturization. Designing a power amplifier circuit that can process signals of different frequencies, this method occupies a relatively small circuit area and is more suitable for today's wireless communication systems. Therefore, studying dual-band and multi-band power amplifiers has important practical significance for the development of wireless communication systems. At the same time, existing dual-band matching network designs usually require a separate harmonic control network, which increases the complexity and cost of the design. Therefore, developing a dual-band matching network that can simplify the design without the need for a separate harmonic control network is crucial for efficient output matching design.

[0003] Traditional dual-frequency matching network design methods include using simple dual-frequency matching networks (such as two strings of microstrip lines to match the real impedance at any two frequencies, or π-type structure matching) and complex dual-frequency matching networks (such as three strings of microstrip lines to match the complex impedance at any two frequencies, or T-type structure matching). In addition, existing dual-frequency filtering structures, double-sided parallel stripline structures, and stepped impedance transformers are often used to design harmonic control networks to manage second, third, and higher harmonics respectively. These methods cause the output matching network of the power amplifier to become complicated, including a separate harmonic control network, a fundamental frequency matching network, and additional components. Summary of the invention

[0004] In view of the above-mentioned deficiencies in the prior art, the output matching network for a dual-band power amplifier provided by the present invention solves the problem that the output matching network of the power amplifier becomes complicated due to the traditional dual-band matching network design method.

[0005] In order to achieve the above-mentioned purpose of the invention, the technical solution adopted by the present invention is: an output matching network applied to a dual-band power amplifier, comprising:

[0006] A dual transmission network, wherein the input end of the dual transmission network serves as the input end of the output matching network; the dual transmission network includes parallel microstrip lines TL 01 and microstrip line TL 02 ;

[0007] Microstrip Line TL 1 , one end is connected to the output end of the dual transmission network, and the other end is grounded or open;

[0008] Microstrip Line TL 2 , one end is connected to the output end of the dual transmission network, and the other end serves as the output end of the output matching network;

[0009] Open microstrip line TL 31 , with the microstrip line TL 2 The other end is connected;

[0010] Ground microstrip line TL 32 , with the microstrip line TL 2 The other end of the connection.

[0011] Furthermore: the microstrip line TL 01 and microstrip line TL 02 Used for fundamental wave matching;

[0012] The microstrip line TL 1 Used to eliminate the first frequency point f 1 The electrical impedance at

[0013] The microstrip line TL 2 Used to convert the second frequency point f 2 The conductivity at

[0014] The open microstrip line TL 31 and grounded microstrip line TL 32 For the second frequency f 2 Perform susceptance compensation.

[0015] Further: the characteristic impedance at the input end of the dual transmission network is:

[0016]

[0017] Among them, Z 01 Microstrip line TL 01 Characteristic impedance, Z 02 Microstrip line TL 02 The characteristic impedance, θ 01 Microstrip line TL01 The electrical length, θ 02 Microstrip line TL 02 The electrical length, f is the frequency variable, Z in,A is the output admittance Y of the dual transmission network in,A The reciprocal of .

[0018] Furthermore: the microstrip line TL 01 and microstrip line TL 02 Satisfy θ 01 +θ 02 =π.

[0019] Further when B 01 <0, the microstrip line TL 1 For an open microstrip line,

[0020] When B 01 >0, the microstrip line TL 1 For grounded microstrip lines,

[0021] Among them, Z 1 is the characteristic impedance of microstrip line TL1, θ 1 Microstrip line TL 1 The electrical length, B 01 Y in,A The imaginary part, B 11 It is a microstrip line TL 1 At the first frequency f 1 The susceptance value generated at .

[0022] Further: For the microstrip line TL 2 ,satisfy:

[0023]

[0024] Where k is the frequency f 2 and frequency f 1 The frequency ratio, Z 2 Microstrip line TL 1 Characteristic impedance, Z in,B Y in,B The reciprocal of Y in,B Microstrip line TL 2 The input admittance, Z in,C Y in,C The reciprocal of Y in,C Microstrip line TL 2 The output admittance, θ 2 Microstrip line TL 2 The electrical length.

[0025] Further: For the open microstrip line TL31 and grounded microstrip line TL 32 ,satisfy:

[0026]

[0027] Among them, θ 31 Microstrip line TL 31 The electrical length, θ 32 Microstrip line TL 32 The electrical length, B 32 It is a microstrip line TL 32 At the second frequency f 2 The susceptance value generated at .

[0028] The beneficial effects of the present invention are:

[0029] 1. Simple structure: By integrating harmonic control into the dual-frequency matching network, the need for a separate harmonic control network is reduced, simplifying the design complexity;

[0030] 2. High efficiency: The designed dual-band matching network exhibits excellent bandwidth and efficiency performance at both target frequencies of the dual-band power amplifier, verifying the effectiveness of the design method;

[0031] 3. Accurate calculation: Through the mathematical methods of transmission line equations and transmission matrices, matching network parameters can be accurately calculated to ensure the accuracy and repeatability of the design;

[0032] 4. Comprehensiveness: This method is not only applicable to the design of dual-band power amplifiers, but can also be applied to other RF and microwave circuit designs that require dual-band matching and harmonic control. BRIEF DESCRIPTION OF THE DRAWINGS

[0033] Figure 1 Output matching network structure diagram;

[0034] Figure 2 A diagram of an embodiment of applying an output matching network to a dual-band power amplifier with a dual transmission line structure. DETAILED DESCRIPTION

[0035] The specific implementation modes of the present invention are described below so that those skilled in the art can understand the present invention. However, it should be clear that the present invention is not limited to the scope of the specific implementation modes. For those of ordinary skill in the art, as long as various changes are within the spirit and scope of the present invention as defined and determined by the attached claims, these changes are obvious, and all inventions and creations utilizing the concept of the present invention are protected.

[0036] like Figure 1 As shown, in one embodiment of the present invention, an output matching network applied to a dual-band power amplifier is provided, comprising:

[0037] A dual transmission network, wherein the input end of the dual transmission network serves as the input end of the output matching network; the dual transmission network includes parallel microstrip lines TL 01 and microstrip line TL 02 ;

[0038] Microstrip Line TL 1 , one end is connected to the output end of the dual transmission network, and the other end is grounded or open;

[0039] Microstrip Line TL 2 , one end is connected to the output end of the dual transmission network, and the other end serves as the output end of the output matching network;

[0040] Open microstrip line TL 31 , with the microstrip line TL 2 The other end is connected;

[0041] Ground microstrip line TL 32 , with the microstrip line TL 2 The other end of the connection.

[0042] In this embodiment, the microstrip line TL 01 and microstrip line TL 02 Used for fundamental wave matching;

[0043] The microstrip line TL 1 Used to eliminate the first frequency point f 1 The electrical impedance at

[0044] The microstrip line TL 2 Used to convert the second frequency point f 2 The conductivity at

[0045] The open microstrip line TL 31 and grounded microstrip line TL 32 For the second frequency f 2 Perform susceptance compensation.

[0046] In this embodiment, the microstrip line TL 01 , microstrip line TL 02 , microstrip line TL 1 , microstrip line TL 2 , microstrip line TL 31 , microstrip line TL 32 The characteristic impedances are Z 01 , Z 02 , Z 1 , Z 2 , Z 31 , Z 32 , the electrical lengths are θ 01 ,θ02 ,θ 1 ,θ 2 ,θ 31 ,θ 32 .

[0047] It can be deduced that the characteristic impedance at the input end of the dual transmission network is:

[0048]

[0049] In order to make the calculation and derivation clearer and more convenient, the admittance form will be used for derivation:

[0050]

[0051] Among them, Y in is the admittance at the input of the dual transmission network, Z in,A is the output admittance Y of the dual transmission network in,A The reciprocal of .

[0052] The microstrip line TL 01 and microstrip line TL 02 The characteristic impedances are equal and equal to Z 0 When the electrical length of the two transmission lines is 180°, that is, θ 01 +θ 02 =π.

[0053] The above formula can be simplified to:

[0054]

[0055] Afterwards, as well as After simplifying, we get:

[0056] At the first frequency:

[0057]

[0058] Among them G 0 The standard conductivity is 0.02S, Z 0 is the characteristic impedance of the dual transmission network;

[0059] Since θ is satisfied 01 +θ 02 =π, in this embodiment, f=f 1 , Simplifying at the first frequency point gives:

[0060] Y in =G 1 +jB 1 @f1 Substitute it into:

[0061]

[0062] Right now

[0063] Among them, G 1 Yes in The conductivity value, B 1 Yes in The susceptance value.

[0064] According to the above results, it can be deduced that Due to G 1 , B 1 , G 0 is a certain value, we can easily find B 01 , and then put it into the above formula to get Z 0 :

[0065]

[0066] At the same time, the second frequency point f is obtained by using the transmission line formula 2 Y in,A :

[0067] Right now

[0068] The frequency ratio Simplified:

[0069]

[0070] in, Yes in,A The conjugate of It's Z in The conjugation of.

[0071] So we can get Y in,A @f 2 , so far we have obtained Y at two frequency points in,A From the properties of the dual transmission lines, we can see that we have controlled the harmonics at the first frequency point and achieved partial fundamental matching.

[0072] For the microstrip line TL 1 , its function is to eliminate the susceptance at the first frequency point, so the relationship to be satisfied is B 11 =B 01 ;

[0073] When B 01 <0, the microstrip line TL 1 It is an open microstrip line; at this time

[0074] When B 01 >0, the microstrip line TL 1 is the grounded microstrip line;

[0075] Among them, Z 1 is the characteristic impedance of microstrip line TL1, θ 1 Microstrip line TL 1 The electrical length, B 01 Y in,A The imaginary part, B 11 It is a microstrip line TL 1 At the first frequency f 1 The susceptance value generated at Z 1 ,θ 1 is a set of free variables, that is, an equation with two unknowns, so we can transform Z 1 ,θ 1 The value of is controlled within an appropriate range.

[0076] At the same time, we can also calculate the parallel TL 1 The susceptance generated at the second frequency point is:

[0077] (TL 1 for open microstrip lines) or (TL 1 is a grounded microstrip line);

[0078] Among them, B 12 It is a microstrip line TL 1 At the second frequency f 2 Susceptance generated;

[0079] Therefore, we can use this parallel microstrip line TL 1 Eliminate the susceptance at the first frequency point:

[0080]

[0081] And B 22 =B 02 -B 12 ;

[0082] At this point, the first part of the design is completed, and the first frequency point has been converted into a standard pure conductance G 0 , the admittance at the second frequency point is G 22 +jB 22 ;

[0083] The microstrip line TL 2 The function is to convert the conductance at the second frequency point to G 0, and in order not to affect the first frequency point that has been matched, TL 2 The characteristic impedance is 50Ω;

[0084] For the microstrip line TL 2 ,satisfy:

[0085]

[0086] Right now

[0087] Where k is the frequency f 2 Sum frequency f 1 The frequency ratio, Z 2 Microstrip line TL 1 Characteristic impedance, Z in,B Y in,B The reciprocal of Y in,B Microstrip line TL 2 The input admittance, Z in,C Y in,C The reciprocal of Y in,C Microstrip line TL 2 The output admittance of

[0088] It can be deduced that:

[0089]

[0090] Y in,B @f 2 =G 22 +jB 22 , Y in,C @f 2 =G 0 +jB 32 Substituting into the above formula, we get

[0091]

[0092] Simplified:

[0093]

[0094] Calculating the real part of the left and right sides of the above formula yields:

[0095]

[0096] Among them, G 22 Yes in,B At the second frequency f 2 Susceptance value at ;

[0097] Simplified:

[0098]

[0099] In order to find θ 3 , for tan(kθ 3 ) is processed, and the above formula is simplified to obtain

[0100]

[0101] This is a question about tan(kθ 3 ) quadratic equation

[0102] make It is easy to solve:

[0103]

[0104] n is an integer;

[0105] Find TL 2 After the relevant parameters are obtained, it can be easily solved by substituting them into the above formula or using the transmission line formula;

[0106] TL 31 ,TL 32 The function is to eliminate the susceptance at the second frequency point without increasing the susceptance at one frequency point:

[0107] Right now

[0108] So we can get:

[0109] Solving the equation gives

[0110]

[0111] like Figure 2 As shown, in one embodiment of the present invention, based on the proposed output matching network, a dual-band power amplifier operating at 2.6 GHz and 3.5 GHz is designed and manufactured. The amplifier uses a commercial 10W GaN HEMT CreeCG2H40010F device, and circuit simulation and optimization are performed through ADS software, ultimately achieving efficient dual-band power amplification.

[0112] The experimental results show that the designed power amplifier exhibits excellent performance indicators at both 2.6GHz and 3.5GHz frequencies, including:

[0113] In the frequency range of 2.43GHz to 2.79GHz, the saturated output power is 39.5-41.5dBm, the efficiency (DE) is between 60.2%-71.8%, and the bandwidth is 350MHz.

[0114] In the frequency range of 3.4GHz to 3.57GHz, the saturated output power is 39.5-41.4dBm, the efficiency (DE) is between 60.1%-70.1%, and the bandwidth is 170MHz.

[0115] The above results verify the effectiveness and superiority of the proposed design method.

[0116] Through the above technical scheme, the present invention significantly simplifies the output matching circuit design of the power amplifier while achieving dual-frequency matching and harmonic control, improves the efficiency and performance of the design, and provides a novel and efficient method for the design of dual-frequency power amplifiers.

[0117] The above embodiments are only used to illustrate the technical solutions of the present application, rather than to limit them. Although the present application has been described in detail with reference to the aforementioned embodiments, those skilled in the art should understand that they can still modify the technical solutions described in the aforementioned embodiments, or make equivalent replacements for some of the technical features therein. However, these modifications or replacements do not deviate the essence of the corresponding technical solutions from the spirit and scope of the technical solutions of the embodiments of the present application.

Claims

1. An output matching network for a dual-band power amplifier, It is characterized in that include: A dual transmission network, wherein the input end of the dual transmission network serves as the input end of the output matching network; the dual transmission network includes parallel microstrip lines TL 01 and microstrip line TL 02 ; Microstrip Line TL 1 , one end is connected to the output end of the dual transmission network, and the other end is grounded or open; Microstrip Line TL 2 , one end is connected to the output end of the dual transmission network, and the other end serves as the output end of the output matching network; Open microstrip line TL 31 , with the microstrip line TL 2 The other end is connected; Ground microstrip line TL 32 , with the microstrip line TL 2 The other end of the connection.

2. The output matching network for a dual-band power amplifier according to claim 1, It is characterized in that The microstrip line TL 01 and microstrip line TL 02 Used for fundamental wave matching; The microstrip line TL 1 Used to eliminate the first frequency point f 1 The electrical impedance at The microstrip line TL 2 Used to convert the second frequency point f 2 The conductivity at The open microstrip line TL 31 and grounded microstrip line TL 32 For the second frequency f 2 Perform susceptance compensation.

3. The output matching network for a dual-band power amplifier according to claim 2, It is characterized in that The characteristic impedance at the input of the dual transmission network is: Among them, Z 01 Microstrip line TL 01 Characteristic impedance, Z 02 Microstrip line TL 02 The characteristic impedance, θ 01 Microstrip line TL 01 The electrical length, θ 02 Microstrip line TL 02 The electrical length, f is the frequency variable, Z in,A is the output admittance Y of the dual transmission network in,A The reciprocal of .

4. The output matching network for a dual-band power amplifier according to claim 3, It is characterized in that The microstrip line TL 01 and microstrip line TL 02 Satisfy θ 01 +θ 02 =π.

5. The output matching network for a dual-band power amplifier according to claim 1, It is characterized in that When B 01 <0, the microstrip line TL 1 For an open microstrip line, When B 01 >0, the microstrip line TL 1 For grounded microstrip lines, Among them, Z 1 Microstrip line TL 1 The characteristic impedance, θ 1 Microstrip line TL 1 The electrical length, B 01 Y in,A The imaginary part, B 11 It is a microstrip line TL 1 At the first frequency f 1 The susceptance value generated at .

6. The output matching network for a dual-band power amplifier according to claim 1, It is characterized in that For the microstrip line TL 2 ,satisfy: Where k is the frequency f 2 Sum frequency f 1 The frequency ratio, Z 2 Microstrip line TL 1 Characteristic impedance, Z in,B Y in,B The reciprocal of Y in,B Microstrip line TL 2 The input admittance, Z in,C Y in,C The reciprocal of Y in,C Microstrip line TL 2 The output admittance, θ 2 Microstrip line TL 2 The electrical length.

7. The output matching network for a dual-band power amplifier according to claim 6, It is characterized in that For the open microstrip line TL 31 and grounded microstrip line TL 32 ,satisfy: Among them, θ 31 Microstrip line TL 31 The electrical length, θ 32 Microstrip line TL 32 The electrical length, B 32 It is a microstrip line TL 32 At the second frequency f 2 The susceptance value generated at .

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