Drive control device, drive control system, and state estimation method

Abstract

When a control period for executing a process related to sensorless vector control is set as Ts (= 2 * Tc = Th), an estimator applies a high-frequency voltage of a period Th corresponding to a frequency outside an audible domain on a gamma axis of a gamma-delta quasi-synchronous coordinate system in synchronization with Ts and Tc, and detects a stator current including a drive fundamental frequency component and a high-frequency component for each period Tc. At least one of a [gamma]-axis element signal c [gamma] and a [delta]-axis element signal s [delta] is generated by time-differentiating the most recent two stator current detection values for each period Ts in a [gamma] [delta] quasi-synchronous coordinate system, and a [gamma]-axis element signal c [gamma] and / or a [delta]-axis element signal s [delta] is generated using the generated axis element signal. A positive correlation signal pc having a positive correlation with respect to the deviation between the rotor phase and the gamma-axis phase is generated for each cycle Ts, and the positive correlation signal pc is signal-processed so that the positive correlation signal pc becomes zero, thereby generating an estimated value of the rotor phase or speed.

Description

Technical Field

[0001] The present invention relates to a drive control device, a drive control system and a state estimation method. Background Art

[0002] Conventionally, as a so-called sensorless vector control method that does not use a position or speed sensor, there is known a technique for estimating the phase or speed of a rotor by applying a high-frequency voltage having a frequency higher than a driving fundamental frequency (see, for example, Patent Document 1).

[0003] <Prior Art Literature>

[0004] <Patent Documents>

[0005] Patent Document 1: Japanese Patent No. 6150211

[0006] Patent Document 2: Japanese Patent Application Publication No. 2015-208071

[0007] Patent Document 3: Japanese Patent Application Publication No. 2021-164281 Summary of the invention

[0008] <Problems to be Solved by the Invention>

[0009] However, when the frequency of the applied high-frequency voltage is within the human audible range, there is a concern that noise may be generated.

[0010] The present invention provides a drive control device, a drive control system and a state estimation method which can estimate the phase or speed of a rotor by suppressing noise within the audible range of a human being.

[0011] <Methods used to solve the problem>

[0012] One embodiment of the present invention provides a drive control device for controlling the driving of an AC motor whose rotor exhibits salient pole characteristics in response to application of a high-frequency voltage having a frequency higher than a driving fundamental frequency.

[0013] The drive control device includes an estimator for estimating the phase or speed of the rotor.

[0014] The estimator,

[0015] When the control period as the period for executing the processing related to the sensorless vector control is Ts, the current detection period is Tc, and the period of the applied high-frequency voltage is Th, the relationship Ts=2×Tc=Th is established.

[0016] A high-frequency voltage of a period Th corresponding to a frequency outside the audible range is applied in synchronization with a control period Ts and a current detection period Tc on a γ-axis of a γδ quasi-synchronous coordinate system consisting of a γ-axis whose phase changes in such a way that the deviation from the phase of the rotor becomes zero and a δ-axis orthogonal to the γ-axis,

[0017] The stator current including the driving fundamental frequency component and the high frequency component is detected at the current detection period Tc, and the time difference of the two most recent stator current detection values ​​is performed at each period Ts on the γδ quasi-synchronous coordinate system to generate the γ-axis element signal c γ And the delta axis element signal s δ At least one of the above is used, and a positive correlation signal p having a positive correlation with respect to the deviation between the rotor phase and the γ-axis phase is generated at each period Ts using the generated axis element signal. c ,

[0018] So that the positive correlation signal p c The positive correlation signal p c Signal processing is performed to generate an estimated value of the phase or speed of the rotor.

[0019] As another aspect of the present invention, there is provided a drive control system including the drive control device and the AC motor.

[0020] As another aspect of the present invention, there is provided a state estimation method, wherein an estimator provided in a drive control device for controlling the drive of an AC motor whose rotor exhibits salient pole characteristics by applying a high-frequency voltage having a frequency higher than a driving base frequency estimates the phase or speed of the rotor, wherein:

[0021] When the control period as the period for executing the processing related to the sensorless vector control is Ts, the current detection period is Tc, and the period of the applied high-frequency voltage is Th, the relationship Ts=2×Tc=Th is established.

[0022] A high-frequency voltage of a period Th corresponding to a frequency outside the audible range is applied in synchronization with a control period Ts and a current detection period Tc on a γ-axis of a γδ quasi-synchronous coordinate system consisting of a γ-axis whose phase changes in such a way that the deviation from the phase of the rotor becomes zero and a δ-axis orthogonal to the γ-axis,

[0023] The stator current including the driving fundamental frequency component and the high frequency component is detected at the current detection period Tc, and the time difference of the two most recent stator current detection values ​​is performed at each period Ts on the γδ quasi-synchronous coordinate system to generate the γ-axis element signal c γ and the delta axis element signal s δ At least one of the above, and using the generated axis element signal, each cycle T sGenerate a positive correlation signal p that has a positive correlation with the deviation between the rotor phase and the γ-axis phase c ,

[0024] So that the positive correlation signal p c The positive correlation signal p c Signal processing is performed to generate an estimated value of the phase or speed of the rotor.

[0025] <Effects of the Invention>

[0026] According to one embodiment of the present invention, it is possible to estimate the phase or speed of the rotor while suppressing noise within the human audible range. BRIEF DESCRIPTION OF THE DRAWINGS

[0027] Figure 1 This is a diagram showing an example of the relationship between three coordinate systems and the rotor phase.

[0028] Figure 2 This is a block diagram showing a configuration example of a drive control system according to an embodiment.

[0029] Figure 3 This is a timing chart for explaining a state estimation method according to an embodiment.

[0030] Figure 4 is a positive correlation signal p c A diagram showing an example of the positive correlation characteristics of .

[0031] Figure 5 It is shown that the positive correlation signal p c A block diagram of a configuration example of a phase synchronizer that generates a rotor phase estimate and a rotor speed estimate by converging to zero.

[0032] Figure 6 This is a block diagram showing a configuration example of a fundamental wave current converter.

[0033] Figure 7 This is a timing chart for explaining a state estimation method according to an embodiment. DETAILED DESCRIPTION

[0034] Hereinafter, embodiments will be described.

[0035] The technology of the present invention is applicable to an AC motor whose rotor shows salient pole characteristics when a high-frequency voltage having a frequency higher than the driving base frequency is applied. Examples of such AC motors include a permanent magnet synchronous motor having a permanent magnet in the rotor, a wound-rotor synchronous motor, a synchronous reluctance motor, a hybrid excitation synchronous motor having a permanent magnet and an excitation coil in the rotor, and an induction motor.

[0036] The embedded magnet type permanent magnet synchronous motor, synchronous reluctance motor, etc. show salient pole characteristics with respect to the driving voltage and current. These motors also show salient pole characteristics even with the application of high-frequency voltage. On the other hand, the surface magnet type permanent magnet synchronous motor and induction motor that do not show salient pole characteristics with respect to the driving voltage and current show salient pole characteristics with the application of high-frequency voltage. The hybrid excitation type synchronous motor has the characteristics of both permanent magnet type and wound rotor type synchronous motors, and can show salient pole characteristics with the application of high-frequency voltage. In particular, the self-excited hybrid excitation synchronous motor has a strong salient pole characteristic.

[0037] The present invention relates to a state estimation device provided in a drive control device for controlling the drive of the AC motor. For example, the present invention relates to a digital rotor phase speed estimation device for estimating the phase (synonymous with position) or speed of the rotor of the AC motor without using a position speed sensor, i.e., in a sensorless manner. The shape of the applied high-frequency voltage may be rectangular, and the frequency of the applied high-frequency voltage may be an extremely high frequency that is the same as (i.e., the same as or a fraction of) the switching operation frequency of the power converter (the carrier frequency when PWM is used).

[0038] In addition, this specification captures the components of the stator voltage and stator current into two parts: a driving fundamental wave component and a high-frequency component. The driving fundamental wave component is a component directly related to the rotation speed of the motor (i.e., a frequency component of the same degree as the speed (electrical speed) of the rotor), and the high-frequency component is a component with a frequency much higher than the driving fundamental wave component (the value of the frequency is known). In particular, in the present invention, the high-frequency component refers to an extremely high frequency component that is the same degree (i.e., the same or a fraction thereof) as the switching operation frequency of the power converter (the carrier frequency when using PWM).

[0039] The rotor phase to be estimated can be determined at any position of the rotor, but generally, either the negative salient pole phase or the positive salient pole phase of the rotor is selected as the rotor phase. As is well known to those skilled in the art, there is only a phase deviation of ±π / 2 (rad) electrically between the negative salient pole phase and the positive salient pole phase. As long as the phase of one of them is determined, the phase of the other is naturally determined. Taking the above into consideration, hereinafter, unless otherwise explicitly stated, the negative salient pole phase of the rotor is set as the rotor phase. In addition, a two-axis orthogonal coordinate system consisting of a d-axis synchronized with the rotor phase without phase deviation and a q-axis orthogonal to the d-axis is referred to as a dq synchronous coordinate system (refer to Figure 1 ).

[0040] From the perspective of the coordinate system to which the high-frequency voltage is applied, the high-frequency voltage application method for applying a rectangular high-frequency voltage is further subdivided. Specifically, it is subdivided into a method of applying a high-frequency voltage on an αβ fixed coordinate system, and a method of applying a high-frequency voltage on a γδ quasi-synchronous coordinate system (a type of rotating coordinate system) that aims to converge to a dq synchronous coordinate system with a phase difference of zero (see Figure 1 ). The high-frequency voltage application method targeted by the present invention is a method for applying a rectangular high-frequency voltage on a γδ quasi-synchronous coordinate system. The high-frequency voltage application method targeted by the present invention can apply a rectangular high-frequency voltage only on the γ axis, that is, high-frequency voltage application on the δ axis can be omitted.

[0041] The technology of the present invention can process the stator current containing high-frequency components in the γδ quasi-synchronous coordinate system.

[0042] Figure 1 Shown to control the coordinate system speed ω specified by the designer γ The γδ quasi-synchronous coordinate system is composed of a γ axis whose phase changes so that the phase deviation with the rotor becomes zero and a δ axis orthogonal to the γ axis. γ It can be one of the estimated values ​​of the rotor speed. The rotation from the main shaft (γ axis) to the secondary shaft (δ axis) is set as the positive direction. Unless otherwise specified, the 2x1 vector signal representing the physical quantity of the AC motor is set as a vector signal defined entirely on this coordinate system.

[0043] Figure 2 1 is a block diagram showing a configuration example of a drive control system according to an embodiment. The drive control system 100 is a system for controlling the driving of an AC motor 1. The drive control system 100 includes the AC motor 1 and a drive control device 200.

[0044] The AC motor 1 is an AC motor in which a rotor exhibits salient pole characteristics in response to application of a high-frequency voltage having a frequency higher than a driving fundamental frequency.

[0045] The drive control device 200 is a digital drive control device that controls the driving of the AC motor 1. The drive control device 200 includes a power converter (inverter) 2, a current detector 3, a 3-phase 2-phase converter 4a, a 2-phase 3-phase converter 4b, vector rotators 5a, 5b, a current controller 6, a command converter 7, a speed controller 8, a fundamental wave current converter 9, a phase speed estimator 10, a multiplier 11, and a cosine-sine signal generator 12.

[0046] The drive control device 200 has, for example, a memory and a processor (e.g., a CPU (Central Processing Unit)), and a part or all of the functions of the drive control device 200 are implemented by the processor operating a program stored in the memory. A part or all of the functions of the drive control device 200 may also be implemented by an FPGA (Field Programmable Gate Array) or an ASIC (Application Specific Integrated Circuit).

[0047] The three-phase stator current i detected by the current detector 3 u 、i v 、i w After being converted into a two-phase current i1 on the αβ fixed coordinate system by the three-phase to two-phase converter 4a, it is converted into a two-phase current i1 on the γδ quasi-synchronous coordinate system with zero phase deviation to the rotor phase (the same phase as the dq synchronous coordinate system) with phase synchronization as the goal by the vector rotator 5a. The fundamental current converter 9 extracts the driving current i1 from the converted two-phase current i1 1f (driving fundamental frequency component of stator current), the extracted driving current i 1f The (driving fundamental frequency component of the stator current) is sent to the current controller 6. The current controller 6 generates a driving two-phase voltage command value v on the γδ quasi-synchronous coordinate system. 1f ＊ This allows the two-phase driving current on the γδ quasi-synchronous coordinate system to follow the current command value of each phase.

[0048] The rectangular single-phase high-frequency voltage command value (γ-axis high-frequency voltage command value v γh ＊ ) is superimposed on the driving 2-phase voltage command value v 1f ＊ The γ-axis element of the superimposed composite 2-phase voltage command value v1 ＊ The δ-axis voltage command value remains unchanged as the driving voltage command value and is sent to the vector rotator 5b. The vector rotator 5b converts the superimposed and synthesized two-phase voltage command value v1 on the γδ quasi-synchronous coordinate system into ＊ Transformed into the 2-phase voltage command value v1 in the αβ fixed coordinate system ＊ , and the transformed 2-phase voltage command value v1 ＊ The 2-phase 3-phase converter 4b converts the 2-phase voltage command value v1 of the αβ fixed coordinate system to ＊ Converted into 3-phase voltage command value v u ＊ 、vv ＊ 、v w ＊ , and the three-phase voltage command value v u ＊ 、v v ＊ 、v w ＊ The final command value is output to the power converter 2. The power converter 2 generates the three-phase voltage command value v u ＊ 、v v ＊ 、v w ＊ The corresponding electric power is applied to the AC motor 1 to drive the AC motor 1 .

[0049] The phase velocity estimator 10 receives as input the stator current (two-phase current i1 in the γδ quasi-synchronous coordinate system) as an output signal of the vector rotator 5a, and outputs a rotor phase estimation value θ α ^、Estimated rotor (electrical) speed ω 2n ^, and the rectangular single-phase high-frequency voltage command value (γ-axis high-frequency voltage command value v γh ＊ ). Rotor phase estimated value θ α After being converted into cosine / sine signals by the cosine-sine signal generator 12, they are transmitted to the vector rotators 5a and 5b that determine the γδ quasi-synchronous coordinate system. α ^ is set to the phase of the γδ quasi-synchronous coordinate system (equivalent to the phase of the γ axis).

[0050] Two-phase current command value i1 in the γδ quasi-synchronous coordinate system ＊ As is well known to those skilled in the art, by changing the torque command value τ ＊ The command converter 7 converts the rotor speed estimated value ω, which is one of the output signals from the phase speed estimator 10. 2n ^ (rotor electrical speed estimation value) is multiplied by the pole pair number N which is a fixed value by a multiplier 11. p The reciprocal of the mechanical speed is converted into the estimated value ω 2m ^, Estimated mechanical speed value after transformation ω 2m ^ is sent to the speed controller 8. Figure 2 An example of configuring a speed control system is shown, so the torque command value τ ＊ As the output of the speed controller 8. As is well known to those skilled in the art, when the control purpose is torque control and a speed control system is not constituted, the speed controller 8 is not required. In this case, the torque command value τ is supplied from the outside.＊ .

[0051] The phase speed estimator 10 is an example of an estimator that estimates the phase or speed of the rotor of the AC motor 1. The phase speed estimator 10 is a digital rotor phase speed estimating device (state estimating device) that estimates the phase or speed of the rotor by applying a high-frequency voltage having a frequency higher than the driving fundamental frequency.

[0052] Next, an example of a state estimation method executed by the phase velocity estimator 10 will be described.

[0053] Figure 3 : is a timing chart for explaining a state estimation method according to an embodiment. Assume that a control period, which is a period for executing a process related to sensorless vector control, is T S , the current detection period is Tc, and the period of the applied high frequency voltage is Th. The subscript k of the current i means the discrete time detection moment when t=kTc, and the subscript k-1 of the voltage v means the voltage applied during the time t=(k-1)Tc~kTc.

[0054] Figure 3 This is an example of the waveform switching time of the rectangular high-frequency voltage and the discrete time detection time (sampling time) of the current. Figure 3 In the figure, (a) means the discrete time detection moment of the period Tc of the stator current, (b) means the rectangular high-frequency voltage of the period Th, and (c) means the high-frequency current corresponding to the applied high-frequency voltage and its discrete time detection value (indicated by the circle). In addition, (d) is an example of the relationship between the PWM carrier and the control period used for the power converter when the high-frequency voltage is applied, and the PWM period is Tc. And, the control period T is the period for executing the processing related to the sensorless vector control. S The relationship between the current detection period Tc and the period Th of the applied high frequency voltage is: S =2×Tc=Th. That is, the phase velocity estimator 10 calculates the phase velocity according to T S =2×Tc periodically generates the rotor phase estimation value and the rotor speed estimation value (refer to Figure 3 In other words, the phase speed estimator 10 applies high frequency voltage and detects current at each cycle Tc, and removes the processing interval for estimating the rotor phase or speed once. In the drive control device of the power converter using PWM carrier, in order to detect the stator current at discrete time with high accuracy, the detection time needs to be set to the peak or valley or both of the PWM carrier. Figure 3 As shown, the discrete time detection instant of the current is also synchronized with the PWM carrier.

[0055] The phase velocity estimator 10 superimposes a high-frequency voltage for phase estimation on the driving voltage of the AC motor 1. When the high-frequency voltage for phase estimation is superimposed on the driving voltage of the AC motor 1, the stator voltage v1, current i1, and linkage flux φ1 can be expressed as synthetic vectors of the basic driving frequency component and the high-frequency component.

[0056] In addition, the frequency ω of the high frequency voltage superimposed and applied for phase estimation is h Set as 『|ω r / ω h The frequency relative ratio is, for example, 0.01 or less. r Indicates the driving frequency of the AC motor 1.

[0057] The phase velocity estimator 10 is r / ω h |<<1' holds true, an amplitude V is applied on the γ axis with a period Th corresponding to a frequency outside the human audible range. h The rectangular high frequency voltage v γh . Through the high frequency voltage v γh The application of high frequency current is generated.

[0058] The human audible range can generally be said to be above 20 Hz and below 20 kHz, but since humans cannot hear high-frequency sounds, it can also be defined as "above 20 Hz and below 19 kHz" or "above 20 Hz and below 16 kHz."

[0059] Rectangular high frequency voltage v γh The upper limit of the period Th is about 1 / 20 of the electrical time constant (L / R) of the AC motor 1. The upper limit of the control period Ts is naturally determined by the upper limit of the period Th. L is the winding inductance of the AC motor 1, and R is the winding resistance value of the AC motor 1.

[0060] Next, the phase velocity estimator 10 performs a rectangular high frequency voltage v in synchronization with the discrete time detection timing (sampling timing) of the stator current. γh That is, the rectangular high-frequency voltage v γh The waveform switching time of the sampling value is consistent with one of the discrete time detection times. In this case, the high-frequency current changes linearly between the discrete time detection times. Therefore, in this case, the differential value of the high-frequency current can be accurately replaced by the time difference of the sampling value.

[0061] The phase velocity estimator 10 detects the stator current including the driving fundamental frequency component and the high frequency component at discrete times of a period Tc. The phase velocity estimator 10 generates the γ-axis element signal c by performing time differentiation on the γδ quasi-synchronous coordinate system at each period Ts between the two most recent stator current detection values. γ And the delta axis element signal s δ For example, the γ-axis component signal c γ And the delta axis element signal s δ It can be expressed as follows.

[0062] c γ =i γ,k -i γ,k－1

[0063] s δ =i δ,k -i δ,k－1

[0064] i γ is the stator current on the γ axis, i δ is the stator current on the δ-axis.

[0065] The phase velocity estimator 10 uses the γ-axis component signal c γ And the delta axis element signal s δ At least one of the above generates a phase deviation θ with respect to the rotor (d-axis) and the γ-axis at each period Ts. γ Positively correlated signal p c For example, the positive correlation signal p c It can be expressed as follows.

[0066] p c =s δ / c γ =(i δ,k -i δ,k－1 ) / (i γ,k -i γ,k－1 ).

[0067] Figure 4 is a positive correlation signal p c A diagram showing an example of the positive correlation characteristics of . Positive correlation signal p c The relative saliency ratios r s The positive correlation characteristic of saliency ratio r s It can be expressed as follows.

[0068] r s =(L q -L d ) / (L q +L d ).

[0069] Ld is the d-axis inductance, L q is the q-axis inductance.

[0070] according to Figure 4 , at a larger saliency ratio r s = 0.5, when the phase deviation θ γ The positive correlation is ensured within the region of 1.1 [rad], and at a small saliency ratio r s = 0.1, when the phase deviation θ γ The positive correlation is ensured within the region of about 0.8 [rad]. As can be understood from the figure, in the phase deviation θ γ In the region within 0.5 [rad], the following relationship holds.

[0071] p c ≒K θ θ γ

[0072] θγ≦0.5

[0073] K θ =(L q -L d ) / L q .

[0074] from Figure 4 The positive correlation characteristic can be understood that the positive correlation signal in the positive correlation area is the effective phase deviation θ γ , should be called the phase deviation equivalent value (refer to Figure 1 ). Of course, the units of the positive correlation signal are rad, the same as the units of the phase deviation.

[0075] Next, the phase velocity estimator 10 makes the positive correlation signal p c The positive correlation signal p c Signal processing is performed to generate an estimated value of the phase of the γδ quasi-synchronous coordinate system. Figure 4 As can be understood from the example, making the positive correlation signal in the positive correlation region approach zero means making the phase deviation θ γ approaches zero, thus making the γδ quasi-synchronous coordinate system converge to the dq synchronous coordinate system (refer to Figure 1 ). The phase of the γδ quasi-synchronous coordinate system (γ axis) and its differential equivalent value are equivalent to the estimated values ​​of the rotor phase and speed in the AC motor. Therefore, based on the phase of the γδ quasi-synchronous coordinate system (phase of the γ axis), the rotor phase estimation value and the rotor speed estimation value can be obtained.

[0076] Figure 5 It is shown that the positive correlation signal p cA block diagram of a configuration example of a phase synchronizer that generates a rotor phase estimation value and a rotor speed estimation value by converging to zero. The phase speed estimator 10 includes a phase synchronizer 20. The phase synchronizer 20 is configured based on a digital generalized integral PLL method. p is the proportionality constant, K I is the integration constant and s is the Laplace operator.

[0077] The phase synchronizer 20 receives the positive correlation signal p as an input signal. c , output the phase of the γδ quasi-synchronous coordinate system (in this example, the phase of the γδ quasi-synchronous coordinate system is equivalent to the rotor phase estimation value)θ α The velocity ω of the quasi-synchronous coordinate system of ^ and γδ γ . Coordinate system velocity ω γ and the estimated rotor (electrical) speed ω 2n ^Essentially the same.

[0078] Therefore, according to the phase velocity estimator 10 in this embodiment, by using the positive correlation signal p c , these estimated values ​​can be obtained with very little computational effort.

[0079] The phase velocity estimator 10 in the present embodiment applies an amplitude V on the γ-axis at a period Th corresponding to a frequency outside the human audible range. h The rectangular high frequency voltage v γh Therefore, due to the high frequency voltage v γh By applying a frequency outside the audible range, it is possible to suppress the high-frequency voltage v γh The phase or speed of the rotor can be estimated by using the noise (sound within the human audible range) generated by the application of force.

[0080] For example, in Patent Document 1, the frequency of the applied high-frequency voltage is set to half of the PWM frequency and the control frequency (the period of the applied high-frequency voltage is set to twice the PWM period and the control period). In this case, the upper limit of the PWM frequency and the control frequency mainly depends on the computing power (processing speed) of the microcomputer because the control frequency becomes a shackle. If low-cost microcomputers that use sensorless control are considered, it is currently around 20kHz. In this way, the frequency of the applied high-frequency voltage is around 10kHz, which is within the human audible range. As a result, high-frequency noise (sound within the human audible range) generated by the application of the high-frequency voltage is generated, which may cause the user to feel uncomfortable.

[0081] In contrast, in this embodiment, due to the high frequency voltage v γh By applying a frequency outside the audible range, it is possible to suppress the high-frequency voltage v γh The noise (sound within the human audible range) generated by the application of Figure 3 In the example, the PWM frequency and current detection frequency are set to 40kHz (PWM period = current detection period Tc = 25μs), and the high-frequency voltage frequency and control frequency are set to half of 20kHz (high-frequency voltage period Th = control period Ts = 50μs). γh The frequency of 20kHz deviates from the human audible range, so it is possible to suppress the high-frequency voltage v γh Noise (sound within the human audible range) generated by the application of

[0082] also, Figure 3 (c) only records the applied high-frequency current, but the stator current is detected by including not only the high-frequency component current but also the driving fundamental component current. The actual detected current can be expressed as follows:

[0083] Current detection D1: I all,k－1 =I c -I h

[0084] Current detection D2: I all,k =I c +I h .

[0085] I c Indicates the current of the driving fundamental component (fundamental current), I all It represents the stator current including the driving fundamental frequency component and the high frequency component. r / ω h |<<1" holds true, I c It can be considered that there is almost no change between the timing of current detection D1 and the timing of current detection D2.

[0086] That is, the fundamental wave current converter 9 (see Figure 2 ) is calculated based on the following relationship: c , the driving current i can be extracted from the converted 2-phase current i1 1f (=I c ).

[0087] (I all,k－1 +I all,k ) / 2=I c

[0088] That is, since the driving current i can be extracted by such a simple process 1f (=I c ), so the load applied to the microcomputer can be suppressed.

[0089] Figure 61 is a block diagram showing a configuration example of a fundamental wave current converter. The converted two-phase current i1 includes a γ-axis current i 1γ and the δ-axis current i 1δ The fundamental current converter 9 converts the γ-axis current i 1γ and the δ-axis current i 1δ By applying the above-mentioned relational expressions respectively, the driving current i1f can be extracted.

[0090] Although the embodiments have been described above, the technology of the present invention is not limited to the above-described embodiments, and various modifications and improvements such as combination with or replacement of part or all of other embodiments are possible.

[0091] For example, the current detector 3 is not limited to a sensor such as a CT (Current Transformer) for detecting the current flowing to the AC motor 1, but may detect a single shunt current flowing on the DC side of the power converter 2. The single shunt current detection method is a method of using a shunt resistor connected to the DC bus of the power converter (inverter) to detect the phase current of each phase for controlling the motor (for example, refer to Patent Documents 2 and 3).

[0092] Figure 7 1 is a timing diagram for explaining a state estimation method according to an embodiment of the present invention when the drive of an AC motor is controlled by a single shunt current detection method. In the case of the single shunt current detection method, the switching timing of the high-frequency voltage is not necessarily consistent with the current detection timing. Therefore, in the single shunt current detection method, the discrete time detection period of the current is also synchronized with the period of the PWM carrier, so the rectangular high-frequency voltage based on the phase velocity estimator is applied at a time synchronized with the discrete time detection period of the stator current. As a result, the rectangular high-frequency voltage v based on the phase velocity estimator 10 is γh The application is performed in synchronization with the discrete-time detection instants of the stator current.

[0093] This international application claims priority based on Japanese Patent Application No. 2022-087083 filed on May 27, 2022, and Japanese Patent Application No. 2023-021523 filed on February 15, 2023, and all contents of both applications are incorporated herein by reference.

[0094] Explanation of symbols

[0095] 1: AC motor

[0096] 2: Power Converter

[0097] 3: Current detector

[0098] 4a: 3-phase 2-phase converter

[0099] 4b: 2-phase to 3-phase converter

[0100] 5a: Vector Rotator

[0101] 5b: Vector Rotator

[0102] 6: Current controller

[0103] 7: Instruction converter

[0104] 8: Speed ​​controller

[0105] 9: Fundamental current converter

[0106] 10: Phase velocity estimator

[0107] 11: Multiplier

[0108] 12: Cosine and sine signal generator

[0109] 20: Phase Synchronizer

[0110] 100: Drive control system

[0111] 200: Drive control device.

Claims

1. A drive control device for controlling the driving of an AC motor whose rotor exhibits salient pole characteristics in response to application of a high-frequency voltage having a frequency higher than a driving fundamental frequency, The drive control device includes an estimator for estimating the phase or speed of the rotor. The estimator, When the control period as the period for executing the processing related to the sensorless vector control is Ts, the current detection period is Tc, and the period of the applied high-frequency voltage is Th, the relationship Ts=2×Tc=Th is established. A high-frequency voltage of a period Th corresponding to a frequency outside the audible range is applied in synchronization with a control period Ts and a current detection period Tc on a γ-axis of a γδ quasi-synchronous coordinate system consisting of a γ-axis whose phase changes in such a way that the deviation from the phase of the rotor becomes zero and a δ-axis orthogonal to the γ-axis, The stator current including the driving fundamental frequency component and the high frequency component is detected at the current detection period Tc, and the time difference of the two most recent stator current detection values ​​is performed at each period Ts on the γδ quasi-synchronous coordinate system to generate the γ-axis element signal c γ And the delta axis element signal s δ At least one of the above is used, and a positive correlation signal p having a positive correlation with respect to the deviation between the rotor phase and the γ-axis phase is generated at each period Ts using the generated axis element signal. c , So that the positive correlation signal p c The positive correlation signal p c Signal processing is performed to generate an estimated value of the phase or speed of the rotor.

2. A drive control system comprising the drive control device according to claim 1 and the AC motor.

3. A state estimation method, wherein an estimator provided in a drive control device for controlling the drive of an AC motor whose rotor exhibits salient pole characteristics by applying a high-frequency voltage having a frequency higher than a driving base frequency estimates the phase or speed of the rotor, wherein: When the control period as the period for executing the processing related to the sensorless vector control is Ts, the current detection period is Tc, and the period of the applied high-frequency voltage is Th, the relationship Ts=2×Tc=Th is established. A high-frequency voltage of a period Th corresponding to a frequency outside the audible range is applied in synchronization with a control period Ts and a current detection period Tc on a γ-axis of a γδ quasi-synchronous coordinate system consisting of a γ-axis whose phase changes in such a way that the deviation from the phase of the rotor becomes zero and a δ-axis orthogonal to the γ-axis, The stator current including the driving fundamental frequency component and the high frequency component is detected at the current detection period Tc, and the time difference of the two most recent stator current detection values ​​is performed at each period Ts on the γδ quasi-synchronous coordinate system to generate the γ-axis element signal c γ and the delta axis element signal s δ At least one of the above, and using the generated axis element signal, each cycle T s Generate a positive correlation signal p that has a positive correlation with the deviation between the rotor phase and the γ-axis phase c , So that the positive correlation signal p c The positive correlation signal p c Signal processing is performed to generate an estimated value of the phase or speed of the rotor.

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