PWM modulation method and device for dual three-phase permanent magnet synchronous motor

Abstract

The invention discloses a dual three-phase permanent magnet synchronous motor PWM modulation method and device, solves the problem of how to suppress equivalent odd switch harmonic waves of a dual three-phase permanent magnet synchronous motor driving system, and belongs to the field of motor driving noise suppression. According to the invention, a six-phase CHB drive circuit is adopted to respectively drive a phase winding of a dual three-phase permanent magnet synchronous motor; each phase of CHB driving circuit is formed by cascading N single bodies, and each single body is a full-bridge circuit powered by an independent bus power supply; unipolar carrier phase-shifting modulation is carried out on the dual three-phase permanent magnet synchronous motor by using a six-phase CHB driving circuit, a full-bridge circuit in each single body is decomposed into two half-bridges by the unipolar carrier phase-shifting modulation, two half-bridge modulation waves are staggered by 180 degrees, each carrier single body is sequentially expanded according to pi / N, and the carrier group phase of a second group of three-phase windings is integrally lagged by theta c2; and # imgabs0 # zeta is the phase belt angle of the dual three-phase permanent magnet synchronous motor. According to the invention, frequency multiplication of 4N times of equivalent switching frequency in the aspect of electromagnetic torque can be realized.

Description

Technical Field

[0001] The invention relates to a dual three-phase permanent magnet synchronous motor PWM modulation method and device, belonging to the field of motor drive noise suppression. Background Art

[0002] For electrical systems such as electric vehicles, all-electric aircraft, and all-electric ships that use energy storage as their main energy source, there is a need for efficient operation within a wide power output range. To ensure full utilization of the energy storage system's energy, this type of drive system is generally a low-voltage, high-power output application. Compared with traditional three-phase motors, dual three-phase motors have the advantages of low back electromotive force, small single-phase current, high output power, and high operating efficiency. They show good application value in the field of low-voltage, high-power, and efficient drive operations.

[0003] The characteristics of the electromagnetic torque noise spectrum are directly related to factors such as the system switching frequency, modulation mode, and topology. How to suppress noise so that the motor drive system meets the low noise index requirements in a wider frequency band is also the main problem faced by the electric drive system. The core of noise suppression is to increase the switching frequency, but a higher switching frequency will increase the switching loss of the switch tube and reduce the system efficiency. Therefore, the upper limit of the driver switching frequency will be limited by the system efficiency requirements. Compared with the traditional two-level circuit, the multi-level circuit has the characteristics of low voltage stress, low switching loss, low harmonic content, low electromagnetic interference, and increased device control dimension. It is suitable for applications with sensitive requirements for torque pulsation and noise suppression. The equivalent switching frequency is the same. The more levels there are, the lower the actual switching frequency is, and the higher the efficiency is generally. Therefore, the multi-level circuit is an effective solution for high-efficiency and low-vibration motor drive. Cascaded H-Bridge (CHB) converter is generally used as a high-voltage and high-power power electronic device. It is mainly used in power applications such as static VAR generators and active power filters in power systems that have harmonic suppression and reactive power compensation requirements. At the same time, CHB is also used in high-voltage and high-power transmission systems because of its high-voltage and high-power output characteristics. CHB is rarely used in low-voltage motor drives, especially low-voltage dual three-phase permanent magnet synchronous motor drives.

[0004] At present, the improvement of the equivalent switching frequency of the CHB driver is mainly achieved by increasing the number of cascades or increasing the switching frequency of the switch tube. The former requires additional circuits, which is not conducive to the high power density integration of the system, and the latter will bring additional component losses and cannot guarantee the system efficiency. Summary of the invention

[0005] In order to solve the problem of how to suppress the equivalent odd-order switching harmonics of a dual three-phase permanent magnet synchronous motor drive system, the present invention provides a dual three-phase permanent magnet synchronous motor PWM modulation method and device.

[0006] A dual three-phase permanent magnet synchronous motor PWM modulation method of the present invention adopts a six-phase CHB drive circuit to drive the dual three-phase permanent magnet synchronous motor, and the six-phase CHB drive circuits respectively drive one phase winding of the dual three-phase permanent magnet synchronous motor; each phase CHB drive circuit is formed by cascading N monomers, and each monomer is a full-bridge circuit powered by an independent bus power supply; the modulation method of the present invention includes:

[0007] S1, generating a six-phase voltage modulation wave of a CHB-driven dual three-phase permanent magnet synchronous motor, staggering the modulation waves of the two half-bridges in each monomer of the CHB drive circuit by 180°, and obtaining a given modulation wave voltage of each half-bridge group of six phases;

[0008] S2. The phase shift angle is obtained according to the cascade number of CHB and the phase angle of the dual three-phase permanent magnet synchronous motor:

[0009]

[0010] In the formula, θ c2 is the phase shift angle; ζ is the phase angle of the dual three-phase permanent magnet synchronous motor;

[0011] S3, each carrier monomer of the two groups of three-phase windings is sequentially expanded according to π / N, and the carrier group phase of the second group of three-phase windings is overall delayed by θ c2 , and then generate two groups of three-phase winding carrier groups;

[0012] S4. Perform three-phase decoupling modulation on the carrier groups of the two generated three-phase windings according to the given modulation wave voltage of each half-bridge of the six phases.

[0013] Preferably, S1 comprises:

[0014] The control loop of the dual three-phase permanent magnet synchronous motor is controlled according to the dual dq vector control. The command voltage generated by the control loop is independent of each other at the three-phase neutral points and the zero-sequence component is 0. The inverse transformation of the coordinate transformation matrix is ​​used to generate a six-phase voltage modulation wave.

[0015] According to the unipolar modulation principle, the modulation waves of the two half-bridges in each monomer of the CHB driving circuit are staggered by 180° to generate a given modulation wave voltage for each group of half-bridges in the six phases.

[0016] The present invention also provides a dual three-phase permanent magnet synchronous motor PWM modulation device, comprising a six-phase CHB drive circuit and a modulation module, wherein the six-phase CHB drive circuits respectively drive one-phase windings of the dual three-phase permanent magnet synchronous motor; each phase CHB drive circuit is formed by cascading N monomers, and each monomer is a full-bridge circuit powered by an independent bus power supply;

[0017] The modulation module is used to perform unipolar carrier phase-shift modulation on a dual three-phase permanent magnet synchronous motor using a six-phase CHB drive circuit. The unipolar carrier phase-shift modulation decomposes the full-bridge circuit in each monomer into two half-bridges. The two half-bridge modulation waves are staggered by 180 degrees, and each carrier monomer is unfolded in sequence according to π / N for modulation.

[0018] Preferably, the modulation module is further configured to lag the overall phase of the carrier group of the second three-phase winding by θ on the basis of sequentially expanding each carrier monomer according to π / N. c2 ;

[0019]

[0020] In the formula, θ c2 is the phase shift angle; ζ is the phase belt angle of the dual three-phase permanent magnet synchronous motor.

[0021] The beneficial effects of the present invention are as follows: the present invention does not require hardware changes to the system, and retains the advantages of simple driver structure and reliable operation; the present invention uses CHB to drive a dual three-phase permanent magnet synchronous motor, and realizes 2N times the equivalent switching frequency multiplication at an actual switching frequency; the present invention combines the CHB output voltage with the radial electromagnetic force suppression characteristics of the dual three-phase permanent magnet synchronous motor to provide a CHB-driven dual three-phase permanent magnet synchronous motor-specific carrier harmonic suppression modulation method, which suppresses the odd switching harmonics of the electromagnetic torque on the basis of the original 2N times the equivalent switching frequency, thereby further multiplying the equivalent switching frequency related to the system vibration. BRIEF DESCRIPTION OF THE DRAWINGS

[0022] Figure 1 The topological structure diagram of the dual three-phase permanent magnet synchronous motor system driven by CHB, in which the neutral points of the two sets of three-phase windings are independent of each other, the full-bridge monomer is composed of two half-bridge monomers, and each phase drive topology is composed of a cascade of full-bridge monomers. The cascade number i ranges from 1 to N. ABC is the first set of three-phase windings, DEF is the second set of three-phase windings, and each full-bridge bus voltage is V dc ;

[0023] Figure 2 It is the full-bridge monomer modulation waveform;

[0024] Figure 3 It is a multi-monomer cascade modulation waveform;

[0025] Figure 4 (a) is the equivalent circuit of the motor synchronous rotating coordinate system. Figure 4 (b) is the equivalent circuit of high-order harmonics in the synchronous rotating coordinate system. Figure 4 (c) is the natural coordinate system high-order harmonic equivalent circuit;

[0026] Figure 5 is a control block diagram of the modulation method of the present invention, wherein Ud1 , U q1 , U d2 , U q2 The modulation wave command voltage generated by the dual three-phase permanent magnet synchronous motor control loop, U a1 , U b1 , U c1 , U a2 , U b2 , U c2 is the phase voltage of each phase winding;

[0027] Figure 6 The phase combination relationship between carrier group 1 and carrier group 2 when the number of circuit cascades is 3 for CHB, where carriers 1, 2, and 3 belong to carrier group 1, and carriers 4, 5, and 6 belong to carrier group 2. The overall phase difference between carrier group 1 and carrier group 2 is θ c2 ;

[0028] FIG7 is a current waveform diagram and a current Fourier decomposition diagram of a dual three-phase permanent magnet synchronous motor driven by a CHB driving circuit with a phase angle of 60°, wherein the horizontal axis is the frequency and the vertical axis is the current magnitude, wherein FIG7(a) is a current waveform and a Fourier decomposition diagram of the first set of three-phase windings when the six-phase CHB driving circuit works independently without mutual coordination, i.e., when the two carrier groups overlap, and FIG7(b) is a current waveform and a Fourier decomposition diagram of the six-phase CHB driving circuit when the carrier group 2 is phase-shifted θ in coordination with each other according to the method of the present invention. c2 After that, the first set of three-phase winding current waveforms and their Fourier decomposition;

[0029] FIG8 is a Fourier decomposition diagram of electromagnetic torque of a dual three-phase permanent magnet synchronous motor with a phase angle of 60° driven by a CHB driving circuit, wherein FIG8(a) is a Fourier decomposition diagram of electromagnetic torque of a dual three-phase permanent magnet synchronous motor when the six-phase CHB driving circuit works independently without mutual coordination, i.e., two groups of carrier groups coincide with each other, and FIG8(b) is a diagram of the six-phase CHB driving circuit working in accordance with the method of the present invention in accordance with mutual coordination of carrier group 2 phase shift θ c2 After that, the electromagnetic torque of the dual three-phase permanent magnet synchronous motor is Fourier decomposed;

[0030] FIG9 is a current waveform diagram and a current Fourier decomposition diagram of a dual three-phase permanent magnet synchronous motor driven by a CHB drive circuit with a phase angle of 30°, wherein FIG9(a) is a current waveform and a Fourier decomposition diagram of the first set of three-phase winding current when the six-phase CHB drive circuit works independently without mutual coordination, that is, when the two carrier groups overlap, and FIG9(b) is a current waveform and a Fourier decomposition diagram of the six-phase CHB drive circuit when the carrier group 2 is phase-shifted θ in coordination with each other according to the method of the present invention. c2 After that, the first set of three-phase winding current waveforms and their Fourier decomposition;

[0031] FIG10 is a Fourier decomposition diagram of electromagnetic torque of a dual three-phase permanent magnet synchronous motor with a phase angle of 30° driven by a CHB driving circuit, wherein FIG10(a) is a Fourier decomposition diagram of electromagnetic torque of a dual three-phase permanent magnet synchronous motor when the six-phase CHB driving circuit works independently without mutual coordination, i.e., two carrier groups coincide with each other, and FIG10(b) is a diagram of the six-phase CHB driving circuit working in accordance with the method of the present invention in cooperation with carrier group 2 to shift phase θ c2 Finally, the electromagnetic torque of the dual three-phase permanent magnet synchronous motor is Fourier decomposed. DETAILED DESCRIPTION

[0032] The following will be combined with the drawings in the embodiments of the present invention to clearly and completely describe the technical solutions in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0033] It should be noted that, in the absence of conflict, the embodiments of the present invention and the features in the embodiments may be combined with each other.

[0034] The present invention will be further described below in conjunction with the accompanying drawings and specific embodiments, but they are not intended to limit the present invention.

[0035] The dual three-phase permanent magnet synchronous motor PWM modulation method of this embodiment adopts a six-phase CHB drive circuit to drive, and the six-phase CHB drive circuit drives one phase winding of the dual three-phase permanent magnet synchronous motor respectively; each phase CHB drive circuit is composed of N monomers cascaded, and each monomer is a full-bridge circuit powered by an independent bus power supply;

[0036] CHB drive circuit drives dual three-phase permanent magnet synchronous motor topology such as Figure 1 As shown, its monomers are composed of full-bridge circuits powered by independent bus power supplies, and each phase is composed of N monomers cascaded. Taking phase A as an example, its output phase voltage is the sum of each cascaded full bridge:

[0037]

[0038] The expression of the output voltage of the full-bridge monomer depends on the modulation algorithm adopted by the CHB driving circuit. This embodiment uses the most common unipolar carrier phase shift modulation of the CHB driving circuit. The modulation algorithm is to decompose the full-bridge monomer into two half-bridges, and the two half-bridge modulation waves are staggered by 180 degrees, and then each full-bridge carrier monomer is modulated in turn according to π / n.

[0039] Vibration analysis needs to clarify the switching subharmonic band of the inverter output voltage. Therefore, based on the modulation method, the double Fourier decomposition expression of the CHB drive circuit output voltage needs to be listed. From formula (1), it can be seen that the cascaded H-bridge phase voltage is the sum of each full-bridge monomer, so the double Fourier decomposition expression of the full-bridge monomer output voltage needs to be solved. The full-bridge monomer output voltage is composed of dual half-bridge modulation with 180° staggered. The double Fourier decomposition expression of the half-bridge output voltage is:

[0040]

[0041] Where V dc is the bus voltage; M is the modulation ratio, the ratio of the modulation wave peak value to the carrier peak value; ω0 is the modulation wave angular frequency; ω c is the carrier angular frequency; θ0 is the initial phase angle of the modulation wave; θ c is the initial phase angle of the carrier; n is the harmonic number of the modulation wave; m is the harmonic number of the carrier; J n (x) is the nth-order Bessel function of the independent variable x.

[0042] Formula (2) is Figure 1 The double Fourier decomposition of the voltages at points a and z in the half-bridge unit shown in the figure shows that the first term represents the fundamental component generated by the modulation, and the second term represents the carrier harmonic (corresponding to n=0) and the sideband harmonics formed by the interaction of the carrier harmonic modulation and the modulated harmonic (corresponding to n≠0).

[0043] Depend on Figure 2 and Figure 3 It can be seen that the full-bridge unit of the CHB driving circuit with carrier phase shift modulation can be equivalent to the difference of the half-bridge units that are staggered by 180°. Therefore, after substituting 0 and π in equation (2) and performing a series of simplifications, the double Fourier decomposition expression of the full-bridge output voltage can be obtained:

[0044]

[0045] From formula (3), it can be seen that this modulation method only retains the odd-order (2n-1) modulation wave sideband terms of the even-order (2m) carrier harmonics, and the rest are eliminated. Substituting formula (3) into formula (1) and simplifying it, the double Fourier decomposition expression of the output voltage of the CHB driving circuit can be obtained:

[0046]

[0047] In formula (4), the harmonics of the output voltage of the CHB driving circuit change from the 2n-1 modulation wave sideband around the 2m carrier harmonic of the full bridge to the 2n-1 sideband around 2Nm after cascading, which is equivalent to achieving an equivalent switching frequency of 2N times at the actual switching frequency, which conforms to the equivalent switching frequency formula of the CHB driving circuit.

[0048] The motor vibration is mainly caused by the radial electromagnetic force brought by the carrier sideband harmonics. According to the Maxwell tensor method, ignoring the axial electromagnetic force, the radial electromagnetic force can be expressed as follows when the motor is completely aligned and has no tilt:

[0049]

[0050] In the formula, B air-gap (θ, t) is the air gap magnetic induction intensity, μ0 is the vacuum magnetic permeability. The air gap magnetic field is generated by the stator magnetic field and the rotor magnetic field. Here we only focus on the vibration generated by time harmonics, so the electrical angle is regarded as a constant, and then equation (9) can be expressed as:

[0051]

[0052] Where f p (t) is the spatial magnetomotive force generated by the rotor permanent magnet, f s (t) is the fundamental wave magnetomotive force of the stator armature reaction, f h (t) is the stator armature reaction harmonic magnetomotive force.

[0053] The vibration focuses on the harmonic magnetomotive force, and the rotor magnetomotive force and the stator armature reaction fundamental magnetomotive force are combined into the main magnetomotive force f M (t) = f p (t)+f s (t), formula (6) can be expressed as:

[0054]

[0055] At the same time, the harmonic magnetomotive force is decomposed by combining Hopkins' law. The magnetomotive force can be expressed as the product of the turns function and the current. Therefore, the harmonic magnetomotive force can be expressed as:

[0056] f h (θ,t)=ΣN h (θ)(i 1h (t)+i 2h (t)) (8)

[0057] Where N h (θ) is the equivalent turns function with the spatial electrical angle as the independent variable, i 1h (t), i 2h (t) represent the sideband harmonics near the carrier of the first and second sets of three-phase windings respectively.

[0058] Substituting equation (8) into equation (7), the radial electromagnetic force can be expressed as the following three parts:

[0059]

[0060] Where i h1 、ih2 They correspond to the common single-phase carrier sideband harmonic currents in the two sets of three-phase windings. From the above formula, it can be seen that when the carrier sideband harmonic currents of the first set of three-phase windings and the second set of three-phase windings are in opposite phases, the p in formula (9) can be 2r 、p 3r The radial electromagnetic force is eliminated, thereby achieving motor noise reduction. Therefore, the core of motor noise reduction is to construct the anti-phase carrier sideband harmonic current.

[0061] According to the synchronous rotating coordinate mathematical model of the motor, the equivalent circuit can be simplified to: Figure 4 (a), where Ls is the synchronous inductance corresponding to L d With L q , U s1 , U s2 Corresponding to the AC and DC axis voltages of the two sets of three-phase windings, i s1 、i s2 They correspond to the AC and DC axis currents of the two sets of three-phase windings respectively, and e1 and e2 correspond to the AC and DC axis back electromotive force of the two sets of three-phase windings respectively.

[0062] Will Figure 4 (a) High frequency processing can be obtained Figure 4 (b) The equivalent circuit of high-order harmonics in synchronous rotating coordinate system, where U sh1 , U sh2 Corresponding to the AC and DC high-frequency harmonic voltages of the two sets of three-phase windings, i sh1 、i sh2 They correspond to the AC and DC axis harmonic currents of the two sets of three-phase windings respectively.

[0063] The synchronously rotating coordinate system is restored to the natural coordinate system through the inverse transformation matrix. Figure 4 In (c), L1 is the stator self-inductance, M is the stator mutual inductance, and U h1 , U h2 They correspond to the common single-phase carrier sideband harmonic voltage in the two sets of three-phase windings respectively.

[0064] right Figure 4 Writing Kirchhoff's voltage equation in the circuit (c) yields:

[0065]

[0066] From formula (10), we can get that when u h1 =-u h2 When h1 +i h2 =0, so when the inverter outputs the sideband harmonic voltages of the two sets of three-phase motor stator windings in anti-phase, the radial electromagnetic force can be suppressed, thereby achieving motor noise reduction.

[0067] The fundamental phase angles of different three-phase windings of the dual three-phase permanent magnet synchronous motor with a phase angle not equal to 60° are different, which is related to the spatial distribution of the motor phase angle. Assuming that the phase angle of the first set of three-phase windings is 0°, the fundamental phase angle of the second set of three-phase windings is the phase shift angle ζ. When the carrier phase angle of the first set of three-phase windings is 0°, the output voltage expression can be written as formula (11):

[0068]

[0069] The second set of three-phase winding voltage expressions can be expressed as follows:

[0070]

[0071] Compared with the first set of three-phase winding harmonic voltage, the harmonic phase angle of the second set of three-phase winding harmonic voltage is 2mNθ more. c2 +(2n-1)ζ, in order to make the harmonic voltage of the first three-phase winding anti-phase with the harmonic voltage of the second three-phase winding, it is necessary to ensure that:

[0072] 2mNθ c2 +(2n-1)ζ=kπ k=1,3,5,7··· (13)

[0073] It can be known that the carrier phase angle of the second three-phase winding must satisfy:

[0074]

[0075] Because k and 2n-1 are both odd numbers, the carrier phase angle offset of the second set of three-phase windings can be further expressed as:

[0076]

[0077] From formula (15), we can see that when the carrier phase angle of the second three-phase winding satisfies When the system vibration is reduced to an equivalent switching frequency of 4Nm, high-frequency noise reduction can be achieved.

[0078] The dual three-phase permanent magnet synchronous motor PWM modulation method of this embodiment includes:

[0079] Step 1: Figure 5 As shown, the modulation voltage command generated by the control loop is combined with the electrical angle of the dual three-phase permanent magnet synchronous motor to perform coordinate transformation to generate a six-phase voltage modulation wave.

[0080] The dual three-phase permanent magnet synchronous motor control loop follows the dual dq vector control, and the command voltage U generated by the control loop is d1 , U q1 , U d2 , U q2On the basis that the three-phase neutral points are independent of each other and the zero-sequence component is 0, the inverse transformation of the coordinate transformation matrix of the dual three-phase permanent magnet synchronous motor is used to convert the dual dq plane given voltage value into U a1 , U b1 , U c1 , U a2 , U b2 , U c2 Phase voltage of each phase winding.

[0081] The modulation waves of the two half-bridges in each monomer of the CHB driving circuit are staggered by 180° to obtain a given modulation wave voltage of each half-bridge group of six phases;

[0082] Step 2: Obtain the phase shift angle according to the cascade number of the CHB drive circuit and the phase angle of the dual three-phase permanent magnet synchronous motor:

[0083]

[0084] In the formula, θ c2 is the phase shift angle; ζ is the phase angle of the dual three-phase permanent magnet synchronous motor;

[0085] Step 3: Expand each carrier monomer of the two three-phase windings in accordance with π / N, and lag the overall phase of the carrier group of the second three-phase winding by θ c2 ,like Figure 6 As shown, two carrier groups of three-phase windings are generated;

[0086] Step 4: Perform three-phase decoupling modulation on the carrier groups of the two generated three-phase windings according to the given modulation wave voltage of each half-bridge of the six phases.

[0087] The phase angle of the dual three-phase permanent magnet synchronous motor is generally 30° or 60°. The following specifically describes this implementation mode by way of examples for these two situations.

[0088] Example 1: This example uses a 16-pole, 120rpm, 60° phase angle dual three-phase permanent magnet synchronous motor for simulation experiments. The carrier center frequency is selected as 4kHz, and the number of CHB drive circuit cascades is 3. Therefore, the actual switching frequency satisfies the equivalent 1 / 2N times relationship, which is 667Hz. In order to enhance the persuasiveness, this experiment compares the two sets of three-phase windings using independent unipolar carrier phase shift modulation with the method proposed in this embodiment.

[0089] In order to suppress the odd-order carrier frequency harmonic components of the electromagnetic torque, this embodiment adopts the following technical solution:

[0090] Step 1: The dual three-phase permanent magnet synchronous motor control loop is controlled according to the dual dq vector control, and the command voltage U generated by the control loop is d1 , U q1 , U d2, U q2 On the basis that the three-phase neutral points are independent of each other and the zero-sequence component is 0, the inverse transformation of the coordinate transformation matrix of the 60° phase angle dual three-phase permanent magnet synchronous motor is used to convert the given voltage value of the double dq plane into U a1 , U b1 , U c1 , U a2 , U b2 , U c2 Phase voltage of each phase winding.

[0091]

[0092] Step 2: Substitute the CHB drive circuit cascade number 3 and the dual three-phase permanent magnet synchronous motor phase angle 60° (equivalent to two sets of three-phase windings in the same phase) into the phase shift angle calculation equation:

[0093]

[0094] Step 3: According to the second step, the phase difference between the two three-phase winding carrier groups is 30°. The monomers of the CHB drive circuit are all expanded in sequence according to the principle of unipolar carrier modulation, and each group of three carriers is π / N=π / 3. Then, the three carrier phases of the second three-phase winding carrier group are delayed by 30° as a whole, thereby generating two three-phase winding carrier groups.

[0095] Step 4: According to U a1 , U b1 , U c1 , U a2 , U b2 , U c2 The phase voltage of each phase winding, in order to further increase the equivalent switching frequency, this embodiment follows the unipolar modulation principle. The modulation waves of the two half-bridges in each monomer of the CHB driving circuit are staggered by 180° to generate a given modulation wave voltage for each half-bridge of the six phases. Then, the three-phase decoupled sinusoidal pulse width modulation is performed with the two carrier groups generated in the third step, and finally the corresponding switch driving signal is generated to drive the switch tube.

[0096] As shown in Figure 8(a) and Figure 8(b). By comparison, it can be seen that the 60° phase angle dual three-phase permanent magnet synchronous motor and the CHB drive circuit work independently without mutual cooperation, that is, the two groups of carrier groups overlap and operate in comparison. After using the modulation scheme of the two sets of three-phase windings of the present invention, the reverse harmonic voltage can fully comply with the law of eliminating the odd-order switching harmonics of the electromagnetic torque and its sideband harmonics to complete the equivalent frequency doubling. As shown in Figure 7(a) and Figure 7(b), when the first set of three-phase winding current has no coupled inductance at the inverter output, the completely reverse harmonic voltage will form a large differential mode voltage, which will eventually make the current harmonic quality slightly worse.

[0097] Example 2: This example uses a 16-pole, 120rpm, 30° phase angle dual three-phase permanent magnet synchronous motor for simulation experiments. The carrier center frequency is selected as 4kHz and the CHB cascade number is 3. Therefore, the actual switching frequency satisfies the equivalent 1 / 2N times relationship, which is 667Hz. In order to enhance the persuasiveness, this experiment compares the two sets of three-phase windings using independent unipolar carrier phase shift modulation with the method proposed in this example.

[0098] In order to suppress the odd-order carrier frequency harmonic components of the electromagnetic torque, this embodiment adopts the following technical solution:

[0099] Step 1: The dual three-phase permanent magnet synchronous motor control loop is controlled according to the dual dq vector control, and the command voltage U generated by the control loop is d1 , U q1 , U d2 , U q2 On the basis that the three-phase neutral points are independent of each other and the zero-sequence component is 0, the inverse transformation of the coordinate transformation matrix of the 30° phase angle dual three-phase permanent magnet synchronous motor is used to convert the given voltage value of the double dq plane into U a1 , U b1 , U c1 , U a2 , U b2 , U c2 Phase voltage of each phase winding.

[0100]

[0101] Step 2: Substitute the CHB drive circuit cascade number 3 and the dual three-phase permanent magnet synchronous motor phase angle 30° (equivalent to two sets of three-phase windings in the same phase) into the phase shift angle calculation equation:

[0102]

[0103] Step 3: According to the second step, the phase difference between the two three-phase winding carrier groups is 25°. The monomers of the CHB drive circuit are all expanded in sequence according to the principle of unipolar carrier modulation, and each group of three carriers is π / N=π / 3. Then, the three carrier phases of the second three-phase winding carrier group are delayed by 25° as a whole, thereby generating two three-phase winding carrier groups.

[0104] Step 4: According to U a1 , U b1 , U c1 , U a2 , U b2 , U c2The phase voltage of each phase winding, in order to further increase the equivalent switching frequency, this embodiment staggers the modulation waves of each half-bridge of the two groups of three-phase windings by 180 degrees according to the unipolar modulation principle to generate a given modulation wave voltage of each half-bridge of the six phases. Then, the three-phase decoupled sinusoidal pulse width modulation is performed with the two groups of carriers generated in the third step, and finally the corresponding switch drive signal is generated to drive the switch tube.

[0105] As shown in Figure 10 (a) and Figure 10 (b). By comparison, it can be seen that the 30° phase angle dual three-phase permanent magnet synchronous motor and the CHB drive circuit work independently without mutual cooperation, that is, the two carrier groups overlap and operate in comparison. After using the modulation scheme of the two sets of three-phase windings in this embodiment, the reverse harmonic voltage can complete the elimination of the odd switching harmonics of the electromagnetic torque and the suppression of the odd switching sideband harmonics. As shown in Figure 9 (a) and Figure 9 (b), because the 30° phase angle dual three-phase permanent magnet synchronous motor does not completely eliminate the odd switching sideband harmonics, the differential mode voltage formed by it will not be as severe as the 60° phase angle dual three-phase permanent magnet synchronous motor. Finally, the quality of the current switching harmonics of the first set of three-phase windings is improved.

[0106] This implementation is also applicable to applications where dual three-phase motors have different phase angles (such as 15°) and other natural numbers of CHB drive circuits are used as cascade numbers.

[0107] The modulation method of this embodiment can also be implemented by a modulation device, including a six-phase CHB drive circuit and a modulation module, the six-phase CHB drive circuits respectively drive one-phase windings of the dual three-phase permanent magnet synchronous motor; each phase CHB drive circuit is formed by cascading N monomers, and each monomer is a full-bridge circuit powered by an independent bus power supply;

[0108] The modulation module is used to perform unipolar carrier phase shift modulation on the dual three-phase permanent magnet synchronous motor using a six-phase CHB drive circuit. The unipolar carrier phase shift modulation decomposes the full-bridge circuit in each monomer into two half-bridges. The two half-bridge modulation waves are staggered by 180 degrees. Each carrier monomer is sequentially expanded according to π / N for modulation. On the basis of each carrier monomer being sequentially expanded according to π / N, the phase of the carrier group of the second group of three-phase windings is overall delayed by θ c2 ;

[0109]

[0110] In the formula, θ c2 is the phase shift angle; ζ is the phase angle of the dual three-phase permanent magnet synchronous motor. The principle is the same as the modulation method.

[0111] Although the present invention is described herein with reference to specific embodiments, it should be understood that these embodiments are merely examples of the principles and applications of the present invention. It should therefore be understood that many modifications may be made to the exemplary embodiments and that other arrangements may be devised without departing from the spirit and scope of the present invention as defined by the appended claims. It should be understood that the various dependent claims and features described herein may be combined in a manner different from that described in the original claims. It should also be understood that features described in conjunction with individual embodiments may be used in other described embodiments.

Claims

1. A PWM modulation method for a dual three-phase permanent magnet synchronous motor, characterized in that: A six-phase CHB drive circuit is used to drive a dual three-phase permanent magnet synchronous motor. Each of the six-phase CHB drive circuits drives one phase winding of the dual three-phase permanent magnet synchronous motor. Each phase CHB drive circuit is composed of N cascaded monomers, each of which is a full-bridge circuit powered by an independent bus power supply. The modulation method comprises: S1. Generate a six-phase voltage modulation wave for a dual three-phase permanent magnet synchronous motor driven by a CHB. Stagger the modulation waves of the two half-bridges in each unit of the CHB drive circuit by 180° to obtain a given modulation wave voltage for each half-bridge group of the six phases. S2. The phase shift angle is obtained according to the cascade number of CHBs and the phase angle of the dual three-phase permanent magnet synchronous motor: Where θ c2 is the phase shift angle;ζ is the phase angle of the dual three-phase permanent magnet synchronous motor; S3, each carrier monomer of the two three-phase windings is sequentially expanded according to π / N, and the carrier group phase of the second three-phase winding is delayed by θ c2 , and then generate two groups of three-phase winding carrier groups; S4. Perform three-phase decoupling modulation on the carrier groups of the two generated three-phase windings according to the given modulation wave voltage of each half-bridge of the six phases.

2. The PWM modulation method of a dual three-phase permanent magnet synchronous motor according to claim 1, characterized in that: Said S1 comprises: The control loop of the dual three-phase permanent magnet synchronous motor follows dual dq vector control. The command voltage generated by the control loop is independent of each other at the three-phase neutral points and the zero-sequence component is 0. The inverse transformation of the coordinate transformation matrix is ​​used to generate a six-phase voltage modulation wave. According to the unipolar modulation principle, the modulation waves of the two half-bridges in each monomer of the CHB drive circuit are staggered by 180° to generate a given modulation wave voltage for each group of half-bridges in the six phases.

3. The PWM modulation method of a dual three-phase permanent magnet synchronous motor according to claim 2, characterized in that: The phase angle of the dual three-phase permanent magnet synchronous motor is 30°, and the coordinate transformation matrix in S1 is: Wherein, θ represents the electrical angle of the dual three-phase permanent magnet synchronous motor.

4. The PWM modulation method of a dual three-phase permanent magnet synchronous motor according to claim 2, characterized in that: The phase angle of the dual three-phase permanent magnet synchronous motor is 60°, and the coordinate transformation matrix in S1 is: Wherein, θ represents the electrical angle of the dual three-phase permanent magnet synchronous motor.

5. A computer-readable storage device storing a computer program, characterized in that: When the computer program is executed by a processor, the steps of the dual three-phase permanent magnet synchronous motor PWM modulation method according to any one of claims 1 to 4 are implemented.

6. A mass moment attitude control device for a disk satellite, comprising a storage device, a processor, and a computer program stored in the storage device and executable on the processor, characterized in that: The processor executes the computer program to implement the steps of the dual three-phase permanent magnet synchronous motor PWM modulation method according to any one of claims 1 to 4.

7. A computer program product comprising a computer program, characterized in that When the computer program is executed by a processor, the steps of the dual three-phase permanent magnet synchronous motor PWM modulation method according to any one of claims 1 to 4 are implemented.

8. A dual three-phase permanent magnet synchronous motor PWM modulation device, characterized in that: It includes a six-phase CHB drive circuit and a modulation module. The six-phase CHB drive circuit drives one phase winding of the dual three-phase permanent magnet synchronous motor respectively. Each phase CHB drive circuit is composed of N cascaded monomers, and each monomer is a full-bridge circuit powered by an independent bus power supply. The modulation module is used to perform unipolar carrier phase-shift modulation on a dual three-phase permanent magnet synchronous motor using a six-phase CHB drive circuit. The unipolar carrier phase-shift modulation decomposes the full-bridge circuit in each monomer into two half-bridges. The two half-bridge modulation waves are staggered by 180 degrees, and each carrier monomer is modulated in sequence according to π / N.

9. The dual three-phase permanent magnet synchronous motor PWM modulation device according to claim 8, characterized in that: The modulation module is further configured to lag the overall phase of the carrier group of the second three-phase winding by θ on the basis of sequentially expanding each carrier monomer according to π / N. c2 ; Where θ c2 is the phase shift angle; ζ is the phase angle of the dual three-phase permanent magnet synchronous motor.

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