DC-DC electric energy converter for LED driving power supply and regulation and control method thereof

Abstract

The invention discloses a regulation and control method of a DC-DC electric energy converter for an LED driving power supply. A semi-active resonant converter of which the primary side is a controllable full bridge and the secondary side is a half-controlled half bridge is used; based on the topological structure, an unbalanced variable pulse width modulation method is applied, an inner phase shift angle alpha is introduced to a primary side, and an outer phase shift angle phi is introduced to a secondary side; the optimal working mode operation condition is obtained through mode analysis; obtaining an equivalent power model through a fundamental wave approximation method; and applying a path optimization control strategy on the basis, and outputting an optimal path track of the converter to obtain an optimal control scheme. According to the invention, the converter can realize full-soft switching operation in a full-load range, the switching loss is reduced, the converter operates in an optimal path track, the circulating power is reduced, the stability of output electric energy is improved, and the efficient operation of the system is ensured.

Description

Technical Field

[0001] The present invention relates to the technical field of converters, and in particular to a DC-DC power converter for an LED driving power supply and a control method thereof. Background Art

[0002] In recent years, the rapid development of semiconductor technology has spawned a new type of environmentally friendly lighting: light-emitting diodes (LEDs). LEDs have gained widespread attention in the global lighting field for their high safety, environmental protection, and energy saving. They have the characteristics of low voltage and low current, and are not prone to electric shock risks during use, which improves safety. Compared with traditional lighting methods, LEDs reduce environmental pollution during manufacturing and use, which is in line with the concept of sustainable development. In addition, LEDs have higher energy efficiency and lower power consumption. LED lighting has been widely used in homes, commercial places, and public spaces, and has become an important means to reduce energy waste and protect the environment.

[0003] LED driver power supply is an important part of LED lighting system. Although LED has many advantages, key issues such as reliability, performance and life of its driver power supply have become factors affecting the promotion and application of LED. Once the driver power supply is aged or damaged or reaches the end of its life, the stability and life of the LED lighting system will be seriously affected. At present, the common optimization direction of LED driver power supply is to improve efficiency, reduce size, improve stability and reliability. Therefore, it is of great theoretical significance and practical value to study an optimization control method of DC-DC power converter suitable for LED driver power supply to improve the output power stability and service life of LED driver power supply. Summary of the invention

[0004] The purpose of the present invention is to provide a DC-DC power converter for LED driving power supply and a control method thereof, using a semi-active resonant converter with a controllable full bridge on the primary side and a half-controlled half bridge on the secondary side; based on this topological structure, an unbalanced variable pulse width modulation method is applied to introduce an inner phase shift angle α on the primary side and an outer phase shift angle α on the secondary side. It can reduce the circulating power while achieving full-cycle zero-voltage turn-on of all switching tubes and operate at the optimal path trajectory, effectively reducing the conduction loss.

[0005] The technical solution to achieve the purpose of the present invention is:

[0006] A DC-DC power converter for LED driving power supply, comprising a primary side controllable full bridge, a resonant tank, a transformer and a secondary side half-controlled half bridge connected in sequence, wherein the primary side full bridge comprises a switch tube S 1 ~S 4 , the switch tube S 1 ~S4 The primary side active full bridge circuit is formed; the resonant tank includes a resonant capacitor C connected in sequence between the primary side and the transformer s and resonant inductor L s ; The turns ratio of the transformer is 1:n; The secondary side half-controlled half-bridge includes a switch tube Q 1 ~Q 2 and diode D 1 ~D 2 , the switch tube Q 1 ~Q 2 Construct an active half-bridge circuit on the secondary side; introduce an internal phase shift angle α on the primary side and an external phase shift angle on the secondary side The converter is controlled to operate in an optimal path trajectory.

[0007] The present invention also discloses a control method for a DC-DC power converter for an LED driving power supply, which is applied to the above-mentioned DC-DC power converter for an LED driving power supply. The control method comprises the following steps:

[0008] S01: Controls the switch tube S on the primary side 1 The pulse width is 0 to π, the switch tube S 2 The pulse width is π to 2π, and the switch tube S 3 The pulse width is α to 2π, the switch tube S 4 The pulse width is 0 to α, which generates an AC voltage v ab Contains +V in , 0, V out Three voltage levels, V in is the input voltage, V out is the output voltage;

[0009] S02: Controls the switch tube Q on the secondary side 1 The pulse width is arrive Switching tube Q 2 The pulse width is arrive And the switch tube Q 1 Hysteresis switch S 2 Angle This generates an AC voltage v cd ;

[0010] S03: The resonant current i L The zero crossing point of is adjusted to β, so that

[0011] In the preferred technical solution, after step S03, the method further includes: performing steady-state analysis on the semi-active resonant converter using a fundamental wave approximation method to obtain a phasor domain equivalent circuit model, normalizing the parameters, and obtaining an equivalent power model of the converter.

[0012] In a preferred technical solution, the method for obtaining the equivalent power model of the converter includes:

[0013] The base values ​​of voltage, impedance and power are expressed as follows:

[0014] V B =V in ,

[0015] Where n is the number of transformer turns; P rate Indicates rated power;

[0016] According to the equivalent phasor domain circuit model, the equivalent normalized phasors of the primary side voltage and the secondary side voltage are obtained respectively:

[0017]

[0018] in, Respectively represent the equivalent normalized phasors of the primary and secondary voltages; β represents the zero-crossing phase of the equivalent current; M represents the voltage gain of the converter,

[0019] The normalized phasor expression of the resonant current peak is further obtained:

[0020]

[0021] Phase of resonant current:

[0022]

[0023] The resonant current is expressed as:

[0024] i L,pu (ω s t)=I L,pu sin(ω s t+∠i L,pu )

[0025] Where Q represents the quality factor; F represents the normalized switching frequency; ω s represents the switching angular frequency;

[0026] Then the normalized expression of output power is obtained:

[0027]

[0028] In the preferred technical solution, the obtained equivalent power model is optimized and controlled to obtain the optimal path trajectory.

[0029] In the preferred technical solution, the method for optimizing and controlling the obtained equivalent power model includes:

[0030] The normalized expression of the effective value of the resonant current is obtained:

[0031]

[0032] Where I L,pu is the normalized resonant current;

[0033] The optimization control algorithm is:

[0034] L=I rms,pu +λ(P o,pu -C)

[0035] Among them, λ represents the optimal path constraint coefficient, P o,pu is the equivalent power model, and C is a constant.

[0036] In the preferred technical solution, the optimal path trajectory is:

[0037]

[0038] In the preferred technical solution, based on the phase shift angle condition of the optimal working mode, according to the energy transmission of the power converter under the condition of voltage gain M=0.5, the optimal path trajectory condition is further optimized as follows:

[0039]

[0040] The normalized expression of the optimized output power is:

[0041]

[0042] By controlling v ab The level pulse width α can control the converter's optimal path output power P o size.

[0043] The present invention further discloses a control system for a DC-DC power converter for an LED driving power supply, which is applied to the above-mentioned DC-DC power converter for an LED driving power supply. The control system comprises:

[0044] The primary side waveform adjustment module controls the primary side switch tube S 1 The pulse width is 0 to π, the switch tube S 2 The pulse width is π to 2π, and the switch tube S 3 The pulse width is α to 2π, the switch tube S4 The pulse width is 0 to α, which generates an AC voltage v ab Contains +V in , 0, V out Three voltage levels, V in is the input voltage, V out is the output voltage;

[0045] Secondary side waveform adjustment module, controls the secondary side switch tube Q 1 The pulse width is arrive Switching tube Q 2 The pulse width is arrive And the switch tube Q 1 Hysteresis switch S 2 Angle This generates an AC voltage v cd ;

[0046] The control module adjusts the resonant current i L The zero crossing point of is adjusted to β, so that

[0047] In the preferred technical solution, the control module also includes using the fundamental wave approximation method to perform steady-state analysis on the semi-active resonant converter to obtain a phasor domain equivalent circuit model, normalizing the parameters to obtain an equivalent power model of the converter, and optimizing the obtained equivalent power model to obtain the optimal path trajectory.

[0048] Compared with the prior art, the present invention has the following significant advantages:

[0049] (1) The semi-active resonant converter of the present invention can achieve ZVS operation in the full load range, reducing the switching loss of the converter and improving the operating efficiency of the converter.

[0050] (2) The present invention uses a non-balanced variable pulse width modulation method, which can reduce the circulating power while achieving full-cycle zero voltage turn-on of all switch tubes to operate with the optimal path trajectory, effectively reducing the conduction loss. BRIEF DESCRIPTION OF THE DRAWINGS

[0051] Figure 1 It is the schematic diagram of the DC-DC power converter used for LED drive power supply;

[0052] Figure 2 It is the topological diagram of the power converter;

[0053] Figure 3 It is the optimal working mode waveform of the electric energy converter;

[0054] Figure 4It is the equivalent phasor domain circuit of the power converter;

[0055] Figure 5 Yes V in =200V, V out =90V, M=0.5, P rate =400W, simulation waveforms of various circuit parameters;

[0056] Figure 6 Yes V in =200V, V out =90V, M=0.5, P rate =300W, simulation waveforms of various circuit parameters;

[0057] Figure 7 Yes V in =200V, V out =90V, M=0.5, P rate =200W, simulation waveforms of various circuit parameters;

[0058] Figure 8 Yes V in =200V, V out =90V, M=0.5, P rate =100W, the simulation waveforms of various circuit parameters. DETAILED DESCRIPTION

[0059] The principle of the present invention is: using a semi-active resonant converter with a controllable full-bridge on the primary side and a half-controlled half-bridge on the secondary side; applying a non-balanced variable pulse width modulation method based on this topological structure, introducing an inner phase shift angle α on the primary side and an outer phase shift angle φ on the secondary side, which can reduce the circulating power while achieving full-cycle zero voltage turn-on of all switch tubes to operate with the optimal path trajectory, effectively reducing the conduction loss.

[0060] Embodiment 1:

[0061] The topology of the semi-active resonant converter is as follows: Figure 2 The semi-active resonant converter includes a primary-side controllable full bridge, a resonant tank, a transformer, and a secondary-side half-controlled half bridge connected in sequence. The primary-side full bridge includes a switch tube S 1 ~S 4 , switch tube S 1 ~S 4 The primary side active full bridge circuit is formed; the resonant tank includes a resonant capacitor C connected in sequence between the primary side and the transformer s and resonant inductor L s ; The turns ratio of the transformer is 1:n; The secondary side half-controlled half-bridge includes the switch tube Q 1 ~Q 2 and diode D1 ~D 2 , switch tube Q 1 ~Q 2 A secondary side active half-bridge circuit is formed.

[0062] like Figure 1 As shown, an unbalanced variable pulse width modulation method is applied to a semi-active resonant converter to output a three-level voltage waveform; the optimal operating mode operating conditions are obtained through mode analysis; the equivalent power model is obtained through the fundamental wave approximation method; on this basis, a path optimization control strategy is applied to output the optimal path trajectory of the converter and obtain the optimal control scheme.

[0063] The non-balanced variable pulse width modulation method comprises the following steps:

[0064] S01: Controls the switch tube S on the primary side of the semi-active resonant converter 1 The pulse width is 0 to π, the switch tube S 2 The pulse width is π to 2π, and the switch tube S 3 The pulse width is α to 2π, the switch tube S 4 The pulse width is 0 to α, which generates an AC voltage v ab Contains +V in , 0, -V out Three voltage levels, such as Figure 3 As shown;

[0065] S02: Control the switch Q on the secondary side of the semi-active resonant converter 1 The pulse width is arrive Switching tube Q 2 The pulse width is arrive And the switch tube Q 1 Hysteresis switch S 2 Angle This generates an AC voltage v cd ;

[0066] S03: The resonant current i of the semi-active resonant converter L The zero-crossing point of is adjusted to β, which reduces the circulating power of the converter.

[0067] According to the pattern analysis, Under these conditions, the semi-active resonant converter is in the optimal working mode. In this mode, all switches are turned on at zero voltage throughout the entire cycle.

[0068] It should be noted that pattern analysis is a prior art and will not be described in detail here.

[0069] Based on the optimal working mode, the fundamental wave approximation method is used to perform steady-state analysis on the semi-active resonant converter, and the phasor domain equivalent circuit model is obtained. Figure 4 The parameters are normalized and the base values ​​of the main parameters are expressed as follows:

[0070] V B =V in ,

[0071] Where n is the number of transformer turns; P rate Indicates rated power;

[0072] According to the equivalent phasor domain circuit model, the equivalent normalized phasors of the primary side voltage and the secondary side voltage can be obtained respectively:

[0073]

[0074] in, Respectively represent the equivalent normalized phasors of the primary and secondary voltages; β represents the zero-crossing phase of the equivalent current; M represents the voltage gain of the converter, which conforms to the following relationship:

[0075]

[0076] The normalized phasor expression of the resonant current peak is further obtained:

[0077]

[0078] Phase of resonant current:

[0079]

[0080] The resonant current can be expressed as:

[0081] i L,pu (ω s t)=I L,pu sin(ω s t+∠i L,pu )

[0082] Where Q represents the quality factor; F represents the normalized switching frequency; ω s Represents the switching angular frequency; ω r Represents the resonant angular frequency. They can be used to normalize the capacitance and inductance values, and the expressions are as follows:

[0083]

[0084] Then the normalized expression of output power is obtained:

[0085]

[0086] According to the above analysis, the present invention establishes an equivalent power model of the electric energy converter.

[0087] The obtained equivalent output power model is optimized using the path optimization control strategy.

[0088] Specifically, firstly, the normalized expression of the effective value of the resonant current is obtained:

[0089]

[0090] Where I rms,pu is the resonant current i L Normalized effective value, I L,pu is the normalized resonant current. The optimization control algorithm is:

[0091] L=I rms,pu +λ(P o,pu -C)

[0092] Where λ represents the optimal path constraint coefficient; C is a constant.

[0093] Applying the optimization control algorithm to the equivalent power model, the optimal path trajectory expression is obtained:

[0094]

[0095] Based on the phase shift angle condition of the optimal working mode, according to the energy transmission of the power converter under the condition of voltage gain M = 0.5, the optimal path trajectory condition can be further optimized as follows:

[0096]

[0097] The normalized expression of the optimized output power is:

[0098]

[0099] Therefore, by controlling v ab The level pulse width α can control the optimal path output power P of the semi-active resonant converter o size.

[0100] In another embodiment, a control system for a DC-DC power converter for an LED driving power supply is applied to the above-mentioned DC-DC power converter for an LED driving power supply, and the control system includes:

[0101] The primary side waveform adjustment module controls the primary side switch tube S 1 The pulse width is 0 to π, the switch tube S 2 The pulse width is π to 2π, and the switch tube S 3The pulse width is α to 2π, the switch tube S 4 The pulse width is 0 to α, which generates an AC voltage v ab Contains +V in , 0, V out Three voltage levels, V in is the input voltage, V out is the output voltage;

[0102] Secondary side waveform adjustment module, controls the secondary side switch tube Q 1 The pulse width is arrive Switching tube Q 2 The pulse width is arrive And the switch tube Q 1 Hysteresis switch S 2 Angle This generates an AC voltage v cd ;

[0103] The control module adjusts the resonant current i L The zero crossing point of is adjusted to β, so that

[0104] The control module also includes using the fundamental wave approximation method to perform steady-state analysis on the semi-active resonant converter to obtain a phasor domain equivalent circuit model, normalizing the parameters to obtain an equivalent power model of the converter, and optimizing the obtained equivalent power model to obtain the optimal path trajectory.

[0105] The specific implementation is the same as above and will not be repeated here.

[0106] In order for the converter to operate efficiently, appropriate parameter design is required:

[0107] Design: Input voltage V in =200V; output voltage V out =90V; voltage gain M = 0.5; transformer turns ratio is 20:18; normalized switching frequency F = 1.5; quality factor Q = 1.45937; resonant angular frequency ω s =200πkrad / s; resonant network inductance value L s =87.1μH; resonant network capacitance value C s =65.4346nF; full load rated power P rate =400W.

[0108] In order to verify the correctness of the theory, the above parameters were used to simulate full load (400W), 75% load (300W), 50% load (200W), and 25% load (100W) in PSIM software.

[0109] V in =200V, V out =90V, M=0.5, P rate =400W, the simulation waveforms of various circuit parameters are as follows Figure 5 shown.

[0110] V in =200V, V out =90V, M=0.5, P rate =300W, the simulation waveforms of various circuit parameters are as follows Figure 6 shown.

[0111] V in =200V, V out =90V, M=0.5, P rate = 200W, the simulation waveforms of various circuit parameters are as follows Figure 7 shown.

[0112] V in =200V, V out =90V, M=0.5, P rate = 100W, the simulation waveforms of various circuit parameters are as follows Figure 8 shown.

[0113] After verification combined with simulation waveforms, it was found that the theory was consistent with practice, proving that the present invention is feasible.

[0114] The above embodiments are preferred implementation modes of the present invention, but the implementation modes of the present invention are not limited to the above embodiments. Any other changes, modifications, substitutions, combinations, and simplifications made without departing from the spirit and principles of the present invention shall be equivalent replacement modes and shall be included in the protection scope of the present invention.

Claims

1. A DC-DC power converter for LED driving power supply, characterized in that: The invention comprises a primary-side controllable full bridge, a resonant tank, a transformer and a secondary-side half-controlled half bridge connected in sequence, wherein the primary-side full bridge comprises switch tubes S1 to S4, and the switch tubes S1 to S4 constitute a primary-side active full bridge circuit; the resonant tank comprises a resonant capacitor C connected in sequence between the primary side and the transformer s and resonant inductor L s ; The turns ratio of the transformer is 1:n; The secondary side half-controlled half-bridge includes switch tubes Q1~Q2 and diodes D1~D2, and the switch tubes Q1~Q2 constitute a secondary side active half-bridge circuit; an internal phase shift angle α is introduced on the primary side, and an external phase shift angle is introduced on the secondary side The converter is controlled to operate in an optimal path trajectory.

2. A control method for a DC-DC power converter for an LED driving power supply, characterized in that: The DC-DC power converter for LED driving power supply according to claim 1, wherein the control method comprises the following steps: S01: The pulse width of the switch S1 on the primary side is controlled to be 0 to π, the pulse width of the switch S2 is π to 2π, the pulse width of the switch S3 is α to 2π, and the pulse width of the switch S4 is 0 to α, thereby generating an AC voltage v ab Contains +V in , 0, V out Three voltage levels, V in is the input voltage, V out is the output voltage; S02: The pulse width of the switch Q1 on the secondary side is controlled to be arrive The pulse width of the switch tube Q2 is arrive And the switch tube Q1 lags behind the switch tube S2 by an angle of This generates an AC voltage v cd ; S03: The resonant current i L The zero crossing point of is adjusted to β, so that 3. The control method of the DC-DC power converter for LED driving power supply according to claim 2, characterized in that: After step S03, the method further includes: performing steady-state analysis on the semi-active resonant converter by using a fundamental wave approximation method to obtain a phasor domain equivalent circuit model, normalizing the parameters, and obtaining an equivalent power model of the converter.

4. The control method of the DC-DC power converter for LED driving power supply according to claim 3, characterized in that: Methods for obtaining the equivalent power model of the converter include: The base values ​​of voltage, impedance and power are expressed as follows: V B =V in , Where n is the number of transformer turns; P rate Indicates rated power; According to the equivalent phasor domain circuit model, the equivalent normalized phasors of the primary side voltage and the secondary side voltage are obtained respectively: in, Respectively represent the equivalent normalized phasors of the primary and secondary voltages; β represents the zero-crossing phase of the equivalent current; M represents the voltage gain of the converter, The normalized phasor expression of the resonant current peak is further obtained: Phase of resonant current: The resonant current is expressed as: i L,pu (ω s t)=I L,pu sin(ω s t+∠i L,pu ) Where Q represents the quality factor; F represents the normalized switching frequency; ω s represents the switching angular frequency; Then the normalized expression of output power is obtained:

5. The control method of the DC-DC power converter for LED driving power supply according to claim 3, characterized in that: The obtained equivalent power model is optimized and controlled to obtain the optimal path trajectory.

6. The control method of the DC-DC power converter for LED driving power supply according to claim 5, characterized in that: The method for optimizing the equivalent power model includes: The normalized expression of the effective value of the resonant current is obtained: Where I L,pu is the normalized resonant current; The optimization control algorithm is: L=I rms,pu +λ(P o,pu -C) Among them, λ represents the optimal path constraint coefficient, P o,pu is the equivalent power model, and C is a constant.

7. The control method of the DC-DC power converter for LED driving power supply according to claim 5, characterized in that: The optimal path trajectory is:

8. The control method of the DC-DC power converter for LED driving power supply according to claim 5, characterized in that: Based on the phase shift angle condition of the optimal working mode, according to the energy transmission of the power converter under the condition of voltage gain M = 0.5, the optimal path trajectory condition is further optimized as follows: The normalized expression of the optimized output power is: By controlling v ab The level pulse width α can control the converter's optimal path output power P o size.

9. A control system for a DC-DC power converter of an LED driving power supply, characterized in that: The DC-DC power converter for LED driving power supply according to claim 1, wherein the control system comprises: The primary side waveform adjustment module controls the pulse width of the primary side switch S1 to be 0 to π, the pulse width of the switch S2 to be π to 2π, the pulse width of the switch S3 to be α to 2π, and the pulse width of the switch S4 to be 0 to α, thereby generating an AC voltage v ab Contains +V in , 0, V out Three voltage levels, V in is the input voltage, V out is the output voltage; The secondary side waveform adjustment module controls the pulse width of the secondary side switch tube Q1 to arrive The pulse width of the switch tube Q2 is arrive And the switch tube Q1 lags behind the switch tube S2 by an angle of This generates an AC voltage v cd ; The control module adjusts the resonant current i L The zero crossing point of is adjusted to β, so that 10. The control system of the DC-DC power converter for LED driving power supply according to claim 9, characterized in that: The control module also includes using the fundamental wave approximation method to perform steady-state analysis on the semi-active resonant converter to obtain a phasor domain equivalent circuit model, normalizing the parameters to obtain an equivalent power model of the converter, and optimizing the obtained equivalent power model to obtain an optimal path trajectory.

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