Inverter with AC forward bridge and improved DC / DC topology

Abstract

The present invention relates to a DC to AC power converter having a main DC input (1) and a main single-phase AC output (4), the DC to AC power converter comprising: a single DC to DC converter (5); and firstly, according to a direct path, a bidirectional voltage type DC to AC converter (6) cascaded with the DC to DC converter (5), the bidirectional voltage type DC to AC converter (6) having a DC input-output (11) connected in parallel to the DC output (10) and an AC output-input (12) connected to the main AC output (4); and secondly, according to a bypass path, and connected to the bidirectional voltage type DC to AC converter (6) and the low A current-type low-frequency full-switch H-bridge (7) having a DC input and an AC output, hereinafter referred to as an AC forward bridge, is connected in parallel with a low-frequency diode (2), the DC input is connected to the DC output (10) of the single DC to DC converter (5), and the AC output is connected in parallel with the main AC output (4), the operating frequency of the AC forward bridge is less than 1 kHz, so that when the instantaneous voltage between the terminals of the main AC output (4) reaches a predetermined level, the low-frequency forward AC bridge (7) is turned on, the low-frequency diode (2) is reverse biased and non-conductive, and the DC to DC converter (5) directly provides constant power to the load.

Description

Technical Field

[0001] The present invention relates to a DC / AC power converter with improved efficiency, which comprises a main path consisting of a DC / DC converter followed by a bidirectional voltage-type DC / AC converter, and a bypass path consisting of a current-type DC / AC converter, wherein power peaks are delivered by the bypass converter to minimize losses.

[0002] In particular, the present invention is an improvement of the DC / AC power converter described in document WO 2016 / 083143 A1. Background Art

[0003] Figure 1 The classic inverter topology is shown in Figure 1. Whether (potentially) isolated or not, the DC / DC input stage transforms the input DC voltage into a different DC voltage that is well suited for powering the output non-isolated DC / AC inverter bridge, which typically operates in the frequency range of a few kHz to a few MHz and produces a 50Hz or 60Hz sine wave output by typically using high frequency PWM modulation.

[0004] In order to increase the efficiency, document US Pat. No. 7,710,752 B2 shows a parallel configuration in which one path is built around a boost bridge converter and the other direct path is built around a simple bridge converter.

[0005] The problem with this topology is that the most efficient direct path can only be activated when the instantaneous voltage at the output is low enough. It cannot be activated when the instantaneous power demand is greatest. The efficiency improvement is not optimal in terms of the number of components added.

[0006] Document EP 2 270 624 A1 proposes another solution that optimizes efficiency by connecting three current generators (with different characteristics) alternately to three output phases based on which connection is best to maximize efficiency. This topology can only be applied to three-phase output systems and does not solve the insulation problem.

[0007] Document WO 2016 / 083143 A1 proposes an isolated DC to AC converter that provides efficiency improvement by adding a direct path from DC input to AC output. This approach improves efficiency, but its disadvantage is that an additional switching converter or an additional winding is required in the isolated DC / DC converter transformer.

[0008] Document US 7,778,046 B1 proposes an interesting solution to improve the efficiency of a DC / DC converter that can be used in an inverter to perform the required insulation. Figure 35a is especially representative of it. Despite being a very interesting operating mode, this invention has several disadvantages:

[0009] -The converter cannot fully operate in soft-switching operation, which prevents the use of high operating frequencies;

[0010] - Switch S has no overvoltage protection (no clamping effect);

[0011] -Diode CR2 has no overvoltage protection. When switch S is closed (or turned on), capacitor C2 continues to be charged by the current flowing in the leakage inductance of the transformer, and diode CR2 is blocked due to the overvoltage that appears at the terminals of diode CR2 (after its reverse recovery phase), which may be several times the diode blocking voltage and may damage diode CR2. In addition, the possible clamping effect of diode CR1 is offset by the series inductance Lr, because an overvoltage is generated across diode CR1 when switch S is closed. This problem makes it difficult to increase the power as required.

[0012] Document EP 1 852 964 A1 discloses a maximum single-phase inverter 3B-INV, which uses a DC voltage V3B boosted from a solar photovoltaic voltage by a boost chopper circuit arranged at the center as its DC power supply, and single-phase inverters 2B-INV and 1B-INV are arranged on both sides of the maximum single-phase inverter 3B-INV, and these single-phase inverters use DC power supplies V1B and V2B supplied from the maximum DC power supply V3B as their inputs. The AC sides of the corresponding single-phase inverters are connected in series. Therefore, the power conditioner is configured to provide an output voltage by using the sum of the voltages generated by the corresponding single-phase inverters. The chopper circuit is connected between the maximum DC power supply V3B and the DC power supplies V1B and V2B, and power is provided from the maximum DC power supply V3B to the DC power supplies V1B and V2B via a switching device in the single-phase inverter.

[0013] Document US 2017 / 025962 A1 discloses two versions of an isolated single-stage converter AC / DC power factor correction (PFC) converter topology. One version is with a full-bridge rectifier at its input, while the other version is a true bridgeless version. The two topology versions are characterized by their new configurations and circuits including a simplified damper circuit and a clamping capacitor flip circuit, and their control methods allow them to implement an improved single-stage isolated power factor converter suitable for high-power operation, with zero voltage switching characteristics to maximize conversion efficiency and minimize electromagnetic interference generation, no additional circuit is required to limit surge current, achieves fairly low input current total harmonic distortion (THD), and is easy to control. The second version provides a true bridgeless single-stage isolated power factor converter with even higher efficiency and lower input current THD.

[0014] Document US 2013 / 223106 A1 discloses a switching circuit for a power converter, the switching circuit comprising a first active switch coupled between a first terminal of an input terminal of the power converter and a first terminal of a primary winding of a transformer. A second active switch is coupled between a second terminal of the input terminal and a second terminal of the primary winding. An output capacitance of the first active switch is greater than an output capacitance of the second active switch. A first passive switch is coupled between a second terminal of the primary winding and a first terminal of the input terminal. A second passive switch is coupled between a second terminal of the input terminal and a first terminal of the primary winding. A reverse recovery time of the first passive switch is greater than a reverse recovery time of the second passive switch.

[0015] Purpose of the Invention

[0016] The present invention aims to propose an inverter which converts a wide range of DC input voltages into an AC output voltage with the highest possible efficiency while maintaining strict requirements on the input DC current shape.

[0017] In particular, one of the requirements is to maintain a constant DC input current, and therefore a constant input power, even if the AC output power fluctuates during the 50 Hz / 60 Hz AC cycle.

[0018] Another object of the invention is to make the inverter as compact as possible. Summary of the invention

[0019] A first aspect of the invention relates to a DC to AC power converter having a main DC input and a main single-phase AC output, capable of transforming and adapting a DC voltage at the main DC input to a fundamental frequency f at the main AC output. 0 The DC to AC power converter can provide a sinusoidal AC voltage and can deliver the rated power at the main AC output to the load, the DC to AC power converter comprising:

[0020] - a single DC to DC converter having said main DC input as input and having a DC output and a tank capacitor connected to said DC output, two low frequency diodes being biased so as to be able to pass current from the DC output to the tank capacitor and, respectively, from the tank capacitor to the DC output;

[0021] a bidirectional voltage-type DC-to-AC converter cascaded with the DC-to-DC converter according to a direct path, the bidirectional voltage-type DC-to-AC converter having a DC input-output connected to the DC output and an AC output-input connected to the main AC output;

[0022] - a current-source low-frequency full-switch H-bridge with a DC input and an AC output, hereinafter referred to as an AC forward bridge, according to a bypass path and in parallel with the bidirectional voltage-source DC to AC converter and the low-frequency diode, the DC input being connected to the DC output of the single DC to DC converter and the AC output being connected in parallel with the main AC output, the operating frequency of the AC forward bridge being less than 1 kHz and preferably 400 Hz or 50 / 60 Hz;

[0023] - control means for controlling the bidirectional voltage type DC to AC converter to deliver a sinusoidal AC voltage at the first AC output-input, and for controlling the AC forward bridge to deliver a quasi-square AC forward current in phase with the sinusoidal AC voltage, the control means being capable of controlling the bidirectional voltage type DC to AC converter and the AC forward bridge so that the latter can operate simultaneously;

[0024] Thus, when the instantaneous voltage between the terminals of the main AC output reaches a predetermined level, the low frequency forward AC bridge is turned on, the low frequency diode is reverse biased and non-conducting, and the DC to DC converter provides constant power directly to the load.

[0025] According to a preferred embodiment, the DC to AC power converter of the present invention is such that:

[0026] - Selecting two closed switches (TPH, TNL; TPL, TNH) of the low-frequency AC forward bridge according to the polarity of the output AC voltage;

[0027] - The DC-to-DC converter is designed to support variable output voltage while delivering nearly constant power;

[0028] - the DC to DC converter is isolated and includes an active clamp on the primary side of the transformer, the active clamp consisting of a main MOSFET (MP), a resonant capacitor (CRP), a capacitor (CAUX) and a second MOSFET (MAUX), the main MOSFET (MP) connecting the primary winding (TFO-P) of the transformer to the primary power supply (VIN+, VIN-) providing the main DC input, the resonant capacitor (CRP) is connected in parallel with the primary MOSFET (MP) to allow the primary MOSFET (MP) to operate under ZVT, the capacitor (CAUX) and the second MOSFET (MAUX) are arranged to provide a voltage clamp on the main MOSFET (MP) to protect the main MOSFET (MP) from overvoltage;

[0029] - the DC to DC converter is of an isolated type and comprises, on the primary side of the transformer: a dual transistor forward converter primary stage consisting of two MOSFETs (MP1, MP2), a respective end of each MOSFET being directly connected to one end of the primary winding (TFO-P) of the transformer and a respective primary power supply terminal (VIN-, VIN+) providing the main DC input; a resonant capacitor (CRP1, CRP2), respectively connected in parallel with each MOSFET (MP1, MP2) to allow the MOSFET (MP1, MP2) to operate under ZVT; and a diode (DAUX1, DAUX2), arranged to respectively connect each MOSFET (MP1, MP2) to a primary power supply terminal (VIN+, VIN-), which primary power supply terminal (VIN+, VIN-) is not directly connected to the primary power supply terminal (VIN-, VIN+) of the corresponding MOSFET (MP1, MP2);

[0030] - the DC to DC converter further comprises, on the secondary side of the transformer: at least a first capacitor (CS1, CS2) establishing an AC connection to the secondary winding (TFO-S) of the transformer; a resonant inductor (LR) capable of being reduced to the leakage inductance of the transformer; a rectifier diode (DR) and a freewheeling diode (DLR) for rectifying the voltage generated at the secondary of the transformer; and a secondary resonant capacitor (CRS) connected in parallel with the rectifier diode (DR); and a decoupling capacitor (CS3) connected in parallel with the terminals of the output DC voltage;

[0031] - the first capacitor (CS1), the resonant inductor (LR), the rectifier diode (DR) and the flyback diode (DLR) are arranged so that during the magnetization phase of the transformer, the input voltage VIN of the converter, which is reflected to the secondary of the transformer as NVIN, charges the first capacitor (CS1) through the rectifier diode (DR) and generates resonance between the first capacitor (CS1) and the resonant inductor (LR), wherein N is the transformer turns ratio, the flyback diode (DRL) is not conducting, and there is no overvoltage at the node between the rectifier diode (DR) and the flyback diode (DRL);

[0032] - the first capacitor (CS1), the resonant inductor (LR), the rectifier diode (DR) and the freewheeling diode (DLR) are arranged so that during the demagnetization phase of the transformer, a current flows from the charged first capacitor (CS1) through the resonant inductor (LR) and the freewheeling diode (DRL) to the load, said current not only transferring the magnetization energy of the transformer but also transferring the energy stored in the first capacitor (CS1) during the magnetization phase;

[0033] -The output power of the DC / DC converter (P O ) is combined with the output power (P FB )Related:

[0034] P O =P FB M, where

[0035] Among them, V IN and V O are the input voltage and output voltage of the DC / DC converter, respectively, and M is a multiplication factor greater than 1, preferably greater than 2;

[0036] - The DC to AC power converter is bidirectional;

[0037] -These low-frequency diodes are replaced by controlled switches;

[0038] - These controlled switches are MOSFETs, IGBTs or relays;

[0039] - The DC to DC converter is non-isolated. BRIEF DESCRIPTION OF THE DRAWINGS

[0040] Figure 1 An inverter topology according to the prior art is schematically represented.

[0041] Figure 2 The principle of a DC / AC power converter with improved efficiency according to the present invention is schematically represented.

[0042] Figure 3 Shown in Figure 2 Typical waveforms in the case of a power converter.

[0043] Figure 4 A circuit topology is shown for a specific embodiment according to the invention.

[0044] Figure 5 is a variable that allows for operational analysis Figure 4 Circuit topology.

[0045] Figure 6 Shows Figure 5 Characteristic voltage and current waveforms in three-phase operation of the circuit depicted in FIG.

[0046] Figure 7 FIG. 4 shows a topology according to another embodiment of the present invention, wherein: Figure 5 The input stage of the topology has been replaced by the input stage of a two-transistor forward converter. DETAILED DESCRIPTION

[0047] Figure 2 The solution proposed by the present invention is presented in FIG.

[0048] An isolated DC / DC converter 5 transfers power from a DC input (typically 48Vdc) to a tank voltage (typically 400Vdc) in constant power mode. A low frequency diode 2 allows current to flow to the tank capacitor 3. A DC / AC converter 6 (typically a full bridge converter) converts the DC voltage into an AC sine wave, typically a 50Hz or 60Hz sine wave.

[0049] During the time evolution of the output sinusoidal wave voltage, as long as the instantaneous output voltage between the node LOUT and the node NOUT is high enough (i.e. typically 200V), the low-frequency "AC forward bridge" 7 can be turned on, and the D1 and D2 diodes 2 are automatically reverse biased (non-conducting), and then the constant power provided by the DC / DC converter 5 is directly provided to the load.

[0050] The DC / AC converter only has to provide the remaining / supplementary part of the power and therefore operates at a lower power level and with lower losses.

[0051] It should be noted that the two switches (TPH / TNL or TNH / TPL) that are turned on in the low frequency "AC forward bridge" 7 are selected according to the polarity of the output voltage.

[0052] The advantage of this topology is that the efficiency of the DC / DC+AC forward bridge is much higher than that of the DC / DC+DC / AC path because the AC forward bridge 7 operates at low frequency and does not require an output inductor. Since the switching frequency is low (typically 100Hz), the commutation losses are low, and the bridge efficiency is limited only by the conduction losses in the switches, which can be increased by placing the switches in parallel. Since the effective resistance is the resistance of one switch divided by the number of switches in parallel, increasing the number of switches in parallel can actually reduce the conduction losses.

[0053] Working Principle of AC Forward Bridge

[0054] Figure 3 Shows Figure 2 Typical voltage and current waveforms of the converter are presented in .

[0055] The output voltage vout is set by the DC / AC converter 6 by methods well known to those skilled in the art of power electronics and inverters.

[0056] From t0 to t1, the AC forward bridge is disconnected and converters 5 and 6 operate as in the prior art. Diodes D1 and D2 (diode 2) are conducting.

[0057] At t1, the output voltage vout, which is equal to V(LOUT)-V(NOUT), is high enough to allow the AC forward bridge 7 to transfer power directly to the output; therefore, the AC forward bridge 7 is turned on. When vout is positive, transistors TPH and TNL are turned on during the interval t1 to t2.

[0058] During the interval t1 to t2, some relations exist:

[0059] -vout is lower than Vtank, and TPH and TNL are turned on, so diodes D1 and D2 are turned off and idc is equal to 0;

[0060] -iforward is directly generated by the isolated DC / DC converter which must maintain constant power (requirement) at its input. Therefore, iforward*vout represents constant power. This law determines the shape of iforward;

[0061] -The output current must be supplied to the load, therefore, iac = iout - iforward. This law determines the shape of iac.

[0062] like Figure 3 As can be seen above, during this time interval, iac is much lower than iout, which explains why the total loss is reduced.

[0063] At t2, transistors TPH and TNL turn off and the converter returns to "normal operation", ie its operation is similar to that during t0 and t1. During this interval, diodes D1 and D2 (diode 2) are conducting again.

[0064] In T 0 / 2 later, the next half cycle begins and vout is negative, the operating sequence is similar to that described above except that transistors TPL and TNH are turned on instead of transistors TPH and TNL.

[0065] DC / DC Converter

[0066] describe

[0067] As explained above, the efficiency of the AC forward bridge 7 is limited only by its conduction losses and is therefore very high. Therefore, the overall inverter efficiency is mainly affected by the efficiency of the DC / DC converter. Figure 2 The proposed structure requires us to realize an efficient DC / DC converter that can support variable output voltage while transmitting almost constant power. The present invention proposes an efficient converter that meets the above requirements. US 7,778,046B1 describes an example of such a converter, but has some disadvantages as explained above. An object of the present invention is to solve these disadvantages in a simple manner.

[0068] One embodiment of the present invention is as follows Figure 4 The circuit topology is shown.

[0069] TFO-P and TFO-S are the primary winding and secondary winding of the transformer respectively.

[0070] MP is the main primary MOSFET that connects the transformer to the primary source. CRP is the resonant capacitor that is placed in parallel with MP and allows the primary MOSFET MP to operate in ZVT (zero voltage transition). CAUX and MAUX provide voltage clamping and protect MP from overvoltage respectively. This part of the circuit is known in the art as "active clamping". Finally, CP is the decoupling capacitor on the primary side.

[0071] In this application, the term ZVT for "zero voltage transition" will be used instead of ZVS for "zero voltage switching" because the transitions are made over the entire cycle (in order to reverse the magnetizing current) and not in short time intervals.

[0072] On the secondary side, capacitors CS1 and CS2 establish the AC connection to the transformer. The secondary transformer winding is TFO-S. LR is the resonant inductance which usually represents the leakage inductance of the transformer. The voltage generated at the secondary of the transformer is rectified by DR (rectifier diode) and DRL (freewheeling diode). CRS is the secondary resonant capacitor connected directly across diode DR. CS3 is the output decoupling capacitor.

[0073] It should be noted that the arrangement of CS1, CS2 and CS3 may be modified without modifying the principle of the proposed circuit.

[0074] The DC / AC converter of the invention has only one DC / DC converter, which is not the case in the prior art (the transformer of the DC / DC converter has, for example, several secondaries). Therefore, the invention has the advantage of reducing the number of electronic components.

[0075] How it works

[0076] Figure 5 A circuit is shown, the operation of which will be further analyzed. LM is the inductor which simulates the magnetizing inductance of the transformer as seen from the primary side. Figure 6 The corresponding waveforms are shown.

[0077] Note that the values ​​of CP (typically 10uF), CAUX (typically 2uF), and CS3 (typically 1uF) are very high, and the corresponding voltages on these capacitors (i.e., VIN, VOUT / N, and VOUT) are approximately constant. It should also be considered that each MOSFET (MP, MAUX) has an intrinsic reverse biased diode (not shown). When MP is off, the potential at D will increase, but is limited by the voltage VOUT / N on CAUX. Above this value, the intrinsic diode of MAUX will turn on.

[0078] The circuit operation is basically divided into three stages (see Figure 6 ).

[0079] - Phase 1:

[0080] In phase 1, the primary MOSFET voltage V(d) is zero. At the beginning of this phase, V(d) naturally and relatively slowly reaches zero, and thus the primary MOSFET MP can turn on at zero voltage. Note that for clarity, the gate voltages Vg and Vga of the MOSFET are Figure 6is arbitrarily vertically displaced in . During this phase, the magnetizing current I(Lm) increases linearly. It should be noted that the supply voltage VIN is reflected to the secondary of the transformer. This voltage is applied across CS1 and resonates with LR, which ensures that the voltage across CS1 is fixed equal to VIN*N at CMID, where N is the transformer turns ratio. For the flyback converter, the energy of the transformer is also stored during this phase, but charging capacitor CS1 allows some additional energy transfer and the available power density is increased. Figure 6 It is shown that the voltage V(cmid) on the CS1 capacitor is almost constant during phase 1, with a small positive slope ripple. The charging current of CS1 is exactly I(Dr). The current in diode DR is part of the resonant sine wave and returns to zero before the end of phase 1 (i.e., before the main primary MOSFET MP is turned off). Therefore, diode DR is cut off at zero current. Diode DRL is not conducting in phase 1.

[0081] - Phase 2

[0082] The main primary MOSFET MP is turned off and the voltage V(d) increases quickly according to the required power level, the internal diode of the auxiliary MOSFET MAUX clamps the voltage V(d), which decreases a little during phase 2. During this phase, the magnetizing current I(Lm) decreases linearly to 0. It should be noted that the capacitor CRP across MP limits the slew rate dV / dt of the voltage V(d) and helps MP to turn off gently. Since the capacitor CAUX clamps V(d), the current in the diode DRL does not follow the magnetizing current at the beginning of phase 2. It takes time until the current in the resonant inductor LR increases. This corresponds to the time required for the current to flow completely to the secondary. Finally, I(DRL) decreases with the magnetizing current. During phase 2, the auxiliary MOSFET MAUX can be turned on to ensure the charge balance of the capacitor CAUX. It should be remembered that the gate voltage V(ga) must be negative to turn MAUX on, because it is a P-type MOSFET (according to Figure 5 ). Therefore, both MOSFET drivers can be commanded with respect to the same potential (ground). At the end of this phase, the magnetizing current I(Lm) reaches 0. At the same time, the current in DRL also reaches 0, resulting in zero current switching of the DRL.

[0083] - Phase 3

[0084] The magnetizing current reverses direction and resonance occurs between the magnetizing inductance LM and the equivalent resonant capacitor. The equivalent capacitor is a combination of CRP and CRS connected in parallel via the transformer. During this resonance, the voltage V(d) decreases gently and reaches 0. This allows the next phase 1 to be started by switching the primary MOSFET MP at zero voltage.

[0085] according to Figure 7 An alternative embodiment of the invention shown in Figure 5 The input stage of the topology is replaced by the input stage of a two-transistor forward (or two-transistor flyback) converter, which is known in the prior art. The main primary MOSFET MP is replaced by two primary MOSFETs MP1, MP2, and the active clamping component consisting of the clamping capacitor CAUX and the auxiliary MOSFET MAUX is replaced by two corresponding diodes DAUX1, DAUX2. The resonant capacitor CRP placed in parallel with MP is replaced by resonant capacitors CRP1, CRP2 placed in parallel with the primary MOSFETs MP1 and MP2 respectively. The operation of the above topology is similar to the above topology ( Figure 5 ) is very similar to the operation.

[0086] Two-transistor forward converters are often used to handle higher input voltages.

[0087] Advantages of the present invention

[0088] Soft switching and compactness

[0089] It is recognized by those skilled in the art that the use of high frequencies when operating DC / DC converters is a key factor in achieving compactness. However, the use of higher frequencies also means an increase in switching losses. From the description of the above stages, the circuit appears to be optimal in terms of switching losses, because both MOSFETs are switched at zero voltage and both diodes are cut off (or reverse polarized) at zero current. Therefore, the proposed circuit is very suitable for building very compact converters.

[0090] Efficient power control and transmission

[0091] like Figure 2 As schematically shown in FIG, the DC / DC converter 5 must maintain a constant current, which is the initial requirement, but its output switches between different voltages. Flyback converters, well known to those skilled in the art, can operate in discontinuous conduction mode and are therefore ideal circuits for this application because the output current is naturally controlled. However, the problem with flyback converters is that they are limited to operating at a few hundred watts because the transformer must store the total transferred energy during the first part of the switching cycle to restore it to the output on the second part of the switching cycle.

[0092] In the present invention, during phase 2, the magnetizing current reflected to the secondary side of the transformer (which flows through the diode DRL) causes the stored transformer magnetizing energy to be transferred to the output. This behavior is very similar to that of a flyback converter. However, the difference of the proposed converter compared to the flyback converter is that the output winding of the transformer is not connected in parallel to the output via DRL, but here is connected in series with CS1. This means that both the energy stored in the magnetizing inductance of the transformer and the energy stored in CS1 are transferred to the output simultaneously. It can be seen that the output power (P O ) is combined with the output power (P FB )Related:

[0093] P O =P FB M, where

[0094] Among them, V IN and V O are the input voltage and output voltage of the DC / DC converter, respectively. The multiplication factor M is greater than 1. Typical values ​​of M are even greater than 2. Therefore, for this application, the proposed converter has the same advantages as the flyback converter, but is able to transfer at least twice the power under the same conditions.

[0095] Constant voltage limiting for semiconductors

[0096] The proposed circuit has very special and interesting properties regarding the semiconductor maximum stress voltage:

[0097] -The operating peak voltage of DRL and DR is V O .

[0098] -MP and MAUX operating peak voltage is V O / N.

[0099] Therefore, the operating peak voltage of all semiconductors is related to V IN Unrelated For large input voltage ranges this is the ideal situation since the switch is used optimally independent of the input voltage.

[0100] This is a very rare property for a DC / DC converter.

[0101] Bidirectional operation

[0102] It should be noted that DR and DRL can be replaced by controlled switches such as MOSFETs, IGBTs, relays, etc. that are controlled simultaneously with MP and MAUX, respectively. In this case, the converter can operate in bidirectional mode and can transfer power from right to left.

Claims

1. A DC to AC power converter having a main DC input (1) and a main single-phase AC output (4), capable of converting and adapting a DC voltage at the main DC input (1) to a sinusoidal AC voltage of a fundamental frequency f0 at the main single-phase AC output (4), and capable of delivering a rated power at the main single-phase AC output (4) to a load, the DC to AC power converter comprising: - a single DC to DC converter (5) having said main DC input (1) as input and having a DC output (10) and a tank capacitor (3) connected to said DC output (10), two low frequency diodes (2) being biased so as to be able to pass current from said DC output (10) to said tank capacitor (3) and, respectively, from said tank capacitor (3) to said DC output (10); - a bidirectional voltage-type DC to AC converter (6) cascaded with the DC to DC converter (5) according to a direct path, the bidirectional voltage-type DC to AC converter (6) having a DC input-output (11) connected to the DC output (10) and a first AC output-input (12) connected to the main single-phase AC output (4); - a current-source low-frequency full-switch H-bridge (7) having a DC input and an AC output, hereinafter referred to as an AC forward bridge, according to a bypass path and in parallel with the bidirectional voltage-source DC to AC converter (6) and the low-frequency diode (2), the DC input being connected to the DC output (10) of the single DC to DC converter (5) and the AC output being connected in parallel with the main single-phase AC output (4), the operating frequency of the AC forward bridge being less than 1 kHz; - control means for controlling the bidirectional voltage type DC to AC converter (6) to deliver a sinusoidal AC voltage at the first AC output-input (12), and for controlling the AC forward bridge to deliver a quasi-square AC forward current in phase with the sinusoidal AC voltage, the control means being capable of controlling the bidirectional voltage type DC to AC converter (6) and the AC forward bridge so that the latter can operate continuously; Therefore, when the instantaneous voltage between the terminals of the main single-phase AC output (4) reaches a predetermined level, the current-mode low-frequency full-switch H-bridge (7) is turned on, the low-frequency diode (2) is reverse biased and non-conductive, and the DC to DC converter (5) supplies constant power directly to the load.

2. The DC to AC power converter according to claim 1, wherein: The two closed switches (TPH, TNL; TPL, TNH) of the current-type low-frequency full-switch H bridge (7) are selected according to the polarity of the output AC voltage.

3. The DC to AC power converter according to claim 1, wherein: The DC to DC converter (5) is designed to support a variable output voltage while delivering constant power.

4. The DC to AC power converter according to claim 1, wherein: The DC to DC converter (5) is of isolated type and comprises an active clamp on the primary side of the transformer, the active clamp being composed of a main MOSFET (MP), a resonant capacitor (CRP), a capacitor (CAUX) and a second MOSFET (MAUX), the main MOSFET (MP) connecting the primary winding (TFO-P) of the transformer to the primary power supply (VIN+, VIN-) providing the main DC input (1), the resonant capacitor (CRP) being connected in parallel with the main MOSFET (MP) to allow the main MOSFET (MP) to operate under ZVT, the capacitor (CAUX) and the second MOSFET (MAUX) being arranged to provide a voltage clamp on the main MOSFET (MP) to protect the main MOSFET (MP) from overvoltage.

5. The DC to AC power converter according to claim 1, wherein: The DC to DC converter (5) is of an isolated type and comprises, on the primary side of the transformer: a dual transistor forward converter primary stage consisting of two MOSFETs (MP1, MP2), a respective end of each MOSFET being directly connected to one end of the primary winding (TFO-P) of the transformer and a respective primary power supply terminal (VIN-, VIN+) providing the main DC input (1); a resonant capacitor (CRP1, CRP2), respectively connected in parallel with each MOSFET (MP1, MP2) to allow the MOSFET (MP1, MP2) to operate under ZVT; and a diode (DAUX1, DAUX2), respectively arranged to connect each MOSFET (MP1, MP2) to a primary power supply terminal (VIN+, VIN-), which is not directly connected to the primary power supply terminal (VIN-, VIN+) of the corresponding MOSFET (MP1, MP2).

6. A DC to AC power converter according to claim 4 or 5, wherein: The DC to DC converter (5) further comprises, on the secondary side of the transformer: at least a first capacitor (CS1) establishing an AC connection to the secondary winding (TFO-S) of the transformer; a resonant inductor (LR) capable of being reduced to the leakage inductance of the transformer; a rectifier diode (DR) and a freewheeling diode (DRL) for rectifying the voltage generated at the secondary of the transformer; and a secondary resonant capacitor (CRS) connected in parallel with the rectifier diode (DR); and a decoupling capacitor (CS3) connected in parallel with the terminals of the DC output (10).

7. The DC to AC power converter according to claim 6, wherein: The first capacitor (CS1), the resonant inductor (LR), the rectifier diode (DR) and the freewheeling diode (DRL) are arranged so that during the magnetization phase of the transformer, the input voltage VIN of the converter, which is reflected to the secondary of the transformer as NVIN, charges the first capacitor (CS1) through the rectifier diode (DR) and generates resonance between the first capacitor (CS1) and the resonant inductor (LR), wherein N is the transformer turns ratio, the freewheeling diode (DRL) is not conducting, and there is no overvoltage at the node between the rectifier diode (DR) and the freewheeling diode (DRL).

8. The DC to AC power converter according to claim 6, wherein: The first capacitor (CS1), the resonant inductor (LR), the rectifier diode (DR) and the freewheeling diode (DRL) are arranged so that during the demagnetization phase of the transformer, a current flows from the charged first capacitor (CS1) through the resonant inductor (LR) and the freewheeling diode (DRL) to the load, and the current not only transmits the magnetization energy of the transformer but also simultaneously transmits the energy stored in the first capacitor (CS1) during the magnetization phase.

9. The DC to AC power converter according to claim 8, wherein: The output power P of the DC / DC converter O The output power P of the equivalent flyback converter is calculated by the following equation FB Related: P O =P FB M, where Among them, V IN and V O are the input voltage and output voltage of the DC / DC converter respectively, and M is a multiplication factor greater than 1.

10. The DC to AC power converter according to claim 1, wherein: These low frequency diodes (2) are replaced by controlled switches.

11. The DC to AC power converter according to claim 10, wherein: These controlled switches are MOSFETs, IGBTs or relays.

12. The DC to AC power converter of claim 1, wherein: The DC to DC converter is non-isolated.

13. The DC to AC power converter of claim 1, wherein: The operating frequency of the AC forward bridge is 400 Hz or 50 / 60 Hz.

14. The DC to AC power converter according to claim 9, wherein: M is a multiplication factor greater than 2.

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